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The measurement of the pulse duration is possibly affected by
the propagation effect along the amplifier, therefore it
becomes necessary to measure the chromatic dispersion of the
amplifying fibre to determine, through computer simulation,
the pulse shape and duration at the output of the ring.
The mechanism that provides the build up of the bunch of
ultrashort pulses is not clear yet; nevertheless it may concern
the nonlinear polarisation evolution inside the cavity. Simultaneously to the pulse autocorrelation, a number of measurements were made to verify that the output state of
polarisation effectively depends on the pump power. This
observation suggests that the dynamical evolution of polarisation inside the ring depends on the power level of the circulating light. If the cavity presents a small anisotropic loss, and
this is possible, the mechanism of selfadjustment of the light
polarisation state induces a reduction of the laser threshold
and pulse reshaping occurs [3, 41;then, in combination with
the nonlinear propagation in the anomalous dispersion region
of the fibre, the generated pulse may really shorten to
hundreds of femtoseconds because of the soliton compression
effect [7].
Conclusion: Passive and active modelocking in an erbium
doped fibre single ring laser is reported. Whenever active
modulation is used, the picosecond regime switches to the
femtosecond regime by simply adjusting the polarisation state
of the light circulating in the ring. This condition is independent of the modulation frequency as long as this is tuned to a
harmonic of the cavity fundamental frequency.
Bordoni and the Italian PT administration. N. Manfredini
wishes to thank Soc. Cavi Pirelli for the study grant.
5th May 1992
F. Fontana and G. Grasso (Pirelli C a d s.p.a., Viale Sarca 202, 20146
Milano, Italy)
N. Manfredini (Dip. Elettronica, Uniuersitd di Pavia, Via Abbiategrasso 209,27100 Pavia, Italy)
M. Romagnoli and B. Daino (Fondazione Ugo Bordoni, Via B. Castiglione 59,00142, Rome, Italy)
'Subpicosecond all-fibre erbium laser', Electron.
DULING, I. N. 111:
Lett., 1991.27, pp. 544-545
MATSAS, v. I.: '3201s generation with passively mode-locked erbium
fibre laser', Electron. Lett., 1991, 27, pp. 730-732
and MENYUK, c . R.: 'Soliton fiber ring
laser', Opt. Lett., 1992.17, pp. 417-419
HOFER, M., FERMAN, M. E., HABERL, F., OBER, M. H., and SCHMIDT, A. 1.:
'Mode-locking with cross-phase and self-phase modulation', Opt.
Lett., 1991,16, pp. 502-504
DAVEY, R. P., SMITH, K., and MCGUIRB A.: 'High speed, mode-locked,
tunable, integrated erbium fibre laser', Electron. Lett., 1992, 28, pp.
CHEN, c . J., WAI, P. K. A.,
and PAY% D. N.: 'Energy quantisation in figure eight fibre laser', Electron. Lett., 1992, 28, pp.
and WOOD, 0.: 'Trapping ofenergy into
solitary waves in amplified nonlinear dispersive systems', Opt.
Lett., 1987,12, pp. 1011-1013
BLOW, K. I., GORAN, N. I.,
Acknowledgment: This work has been carried out in the
framework of the agreement between Fondazione Ugo
X.Nie, D. Raghuramireddy and R. Unbehauen
Indexing terms: Digitalfilters, Filters
A functional theoretical framework is reported for obtaining
efficient 2-D digital filter realisations. A given 2-D transfer
function having a separable denominator polynomial is
developed into certain expansions which may be realised as
two-stage structures. In such structures, only suitable 1-D
substructures need to be searched for to obtain a complete
realisation by connecting 1-D substructures using the corresponding weighting matrix.
(HXz), H j z ) ) = 1/2Rj $I,I= H,(z)F:(l/z) dz/z where the coeffcients of Hf are the complex conjugates of those of H j x ) . The
set of functions
{l,-,- 2 - a l
z - a z ,..,, z - a n
builds a basis of the Hilbert space 2".
By representing the
basis functions as a vector g(z), H(z) can be expressed as
H ( 4 = Cro,P1,....p.Mz)
If another basis of the Hilbert space is given as Yz),there is a
nonsingular constant matrix T such that
= TYZ)
The corresponding expansion of H ( z ) is
Introduction: In 2-D signal processing, it is known that 2-D
recursive digital filters having a denominator-separable transfer function (DSFT) possess a number of favourable properties
with respect to the design and implementations [l-31. Circularly symmetric and quadrantal symmetric fan filters which
are widely used in practice have denominator-separable
transfer functions [SI. The purpose of this Letter is to explain
a functional theoretical approach for this class of transfer
function and point out a very general approach for generating
a class of eficient realisations for DSTFs.
First, we consider the transfer function of a 1-D stable
digital filter H(z) having no poles in the region I z I 2 1. If the
poles of H(z) are assumed to be simple for brevity and denoted
by ai ( i = 1, 2, . .., n), we have
H(z) = p o +
Let Yz) be an orthonormal basis of 2"as given in Reference
4; we then have the Parseval equality
where llH(z)1Iz denotes the 1, denotes the /,-norm of H(z), i.e.
J(H(z), H W .
2-0 denominator-separable tranger functions : A denominatorseparable, stable, quarter-plane 2-D digital filter is described
by its transfer function [l, 51
z - ai
where n is the degree of the transfer function and p i ( i = 0,
1 , . .., n) are constants in the partial fraction expansion of
H(z). NOW,we denote the set of all transfer functions whose
poles are fixed at ai (i = 1, 2 , . .., n ) by R. It can be shown that
the set R represents a Hilbert space under the inner product
ELECTRONICS LETTERS 2nd July 1992 Vol. 28 No. 14
The order of the filter is (m, n) if D,(zl) and D,(z,) are polynomials of degree m and n, respectively. We assume that H(zl,
z2) is analytic in J z l I z 1 and Jz21> 1 and continuous in
Izl 12 1 and 1z212 1. Let the zeros of Dl(zl) and D2(z2) be
simple and denoted by ai(i = 1, 2, ..., m) and /?,U = 1, 2,. ..,
n), respectively. With respect to each complex variable zI and
z2, we can easily construct a set of basis functions. The set of
basis functions formed by the variable z1 and poles ai is
denoted by G I . Similarly, 9, is formed by the basis functions
with the variable zz and the poles p p It can be shown that
6 , €3 6 , is exactly a basis for all possible functions H(zl, zz)
with the same denominator, where 6 , Q !PZis defined as the
tensor product of the two sets [6] 0 , and eZ.Given el as
gl(z,) and GZas dz(z2),H(zl, zz) can be expressed by
W I ,
zz) = d1(z,)Cg,(z,)
where C is an ( m 1) x (n 1) matrix of constants and the
superscript ‘P indicates transpose. Eqn. 7 forms a general
frame for various realisations of 2-D DSTFs with different
1-D basic building blocks. We next give a general realisation
of the 2-D DSTF, and mention some important realisations in
Implication t o implementation: The general expression of eqn.
7 can be implemented as a two-stage realisation as follows. A
one-input-@ 1)-output network is synthesised which realises the vector gl(zl) as the fmt stage. In the second stage,
g2(zz) should be realised as an (n 1)-input-one-output
network. The complete realisation of H(zl, zz) will be completed by connecting the two stages through a weighting
matrix C, as shown in Fig. 1.
compared to the first-order 1-D structure used in the partial
fraction expansion. It is predicted that it is equally true for the
2-D DSTF realisation based on eqn. 9. If other finite wordlength effects need to be accounted for, more general twostage structures have to he considered. Another interesting
expansion is the orthonormal expansion [9]. In this expansion
the vectors g,(z,) and g2(zz)are set to
Fs.1 General two-stage structure for realisationof 2-D DSTF
The two-stage structure has first been explicitly introduced
in Reference 5 for the partial fraction expansion of H(zl, z2)
,..., zl -a,
z1 - a1 z1 - az
Because the basis functions form an orthonormal set [9], we
have the Parseval equality
where gij is the (i, j)th component of matrix C of eqn. 7. This
structure has been shown to exhibit very low roundoff noise
[lo]. It has been shown [lo, 111 that the digital lattice structure proposed by Gray and Markel [lZ 131 can easily be
applied to realise another set of orthonormal basis functions
and we will obtain a two-stage 2-D lattice structure. It is
shown [lo, 111 that a two-stage 2-D lattice structure has
slightly larger roundoff noise than the two-stage orthonormal
structure based on eqn. 10. Furthermore, a two-stage structure
can also be realised using wave digital filers as substructures
for the realisation of gl(z,) and g&). In this way, we can
obtain a class of efficient implementations for the 2-D DSTF
in the general frame of the two-stage structure which can
possess low roundoff nose and/or low sensitivity and be free
from overflow oscillations. Finally, it may be mentioned that a
complete comparison taking all aspacts of the implementation
of the two-stage realisation of a 2-D DSTF using different
substructures is yet to be made. At present, work on roundoff
noise and sensitivity considerations is in progress.
9th April 1992
X. Nie, D. Raghuramiddy and R. Unbehauen (Lehrstuhlfiir Allgemine und Theoretische Elektrotechnik, Cauerstrasse 7, D-8520
Erlangen, Germany)
In References 5 and 7, the aspect of implementation expense
has been discussed successfully. Another expansion of H(zl,
az), which fits into the two-stage realisation, uses allpass firstorder sections [8]. The corresponding vectors gl(zl) andgz(zz)
are set to
1 - afz, 1 - a:zl
,..., ~ ~ - a , , ,
It may be noted that both realisations based on eqns. 8 and 9
yield parallel form structures for the vectors I & , ) and g2(zz)
with first-order building blocks. It is established. that the
first-order I-D allpass structure possesses a low sensitivity
xre, x., RAGHUMMUEDDY, D., and UNBE~~AIJEN,R.: ‘A general
approach for the realization of any IIR digital filters using first-order
all-pas sections’, submitted to IEEE Trans on Circuits and Systems
Refer1 TZAFFSTAS, s G. (Ed.): ‘Multidimensionalsystems, techniques and
applications’(Marcel Dekker,Inc., New York, 1986)
KAWAMATA, M., and HIGUCHI, T.: ‘Synthesis of 2-D separable
denominator digital filters with minimum round-offn o k and no
overflow oscillations’, IEEE Trans., April 1986, CAS-33,pp. 365372
3 KARIVARATHARAJAN, e., and SWAMY, M. N. s.: ‘Quadrantal symmetric associated with two-dimensional digital transfer functions’,
IEEE Trans.,June 1978, CAS-25,pp. 340-343
4 WALSW, I. L: ‘Interpolation and approximation by rational functions in complex domain’ (American Mathematical Society Colloq.
Publications, Vol. XX, 1960)
5 DABBAGH, M. Y., and ALEXANDER, w. E.: ’Multiprocessorimplementation of 2-D denominator separable digital filters for real-time
processing’, IEEE Trans., June 1989,ASP-37, pp. 872-881
6 RICE, Y. R.: The approximation of functions II’ (Addison-Wiley
Publishing Company, Reading, 1969)
7 RAGHUMMUEDDY, o., and UNBMAUW, R.: ‘Highly modular systolic structures for denominator-separable 2-D recursive filters’,
IEEE Trans., Dcamber 1991, ASSP-39, pp. 2725-2728
8 RAGHURAMIREDDY, D., NI+ x., and UNBEHAUM, R.: ‘Minimaland
low-msitivity implementations for a dass of two-dimensional
digital filters’. To be presented at IEEE ISCAS-92, San Diego,
ELECTRONICS LETTERS 2nd July 1992 Vol. 28 No. 14
D., and UNBMAUEN, R.: 'Orthonormal
minimal structure for n-dimensional denominator-separable recursive filters for multiprocessor implementation', Electron. Lett.,
November 1991,27, pp. 2206-2207
10 N E , x. : 'Beitrage zur Synthese zweidimensionaler Digitalfilter'.
Doctorate Thesis to be submitted to Technische Fakultat der Universitat Erlangen-Niirnberg
R.: 'Untersuchung einer Klasse von Strukturen zur
Realisierung eines 2-D Digitalfilters'. Diplom-Engg.Thesis, Lehrstuhl fur Allgemeine und Theoretische Elektrotechnik Universitat
Erlangen-Number& 1992
12 GRAY, A. H., JUN., and MARKEL,I. 0.:'Digital lattice and ladder filter
synthesis', IEEE Trans. Audio Electroacoust., December 1973,
AU-21, pp. 491-500
13 GRAY, A. H., JUN., and MARKEL, I. D.: 'A normalized digital filter
structure', IEEE Trans., June 1975, ASP-23, pp. 268-277
assumed to be driven by a signal source having an equivalent
output resistance of at least 50Q.
H.-M. Rein, E. Bertagnolli, A. Felder and L. Schmidt
Indexing terms: Integrated circults, Driver circuits, Lasers,
ODtical communicafion
A silicon bipolar laser and line driver IC with an outstanding
symmetry of the (single-ended)output pulse shape is presented. The pulse shape can be optimally adjusted via only two
external potentiometers, independent of operating speed and
output current range. The circuit, fabricated in a 0.8pm selfaligned double-polysilicon technology, operates up to
In high-speed digital communication systems and corresponding measurement equipment the shape of the generated pulses
should be as symmetrical as possible with respect to the
switching level. This can be easily achieved by operating the
circuits in the differential mode (see, for example, Reference I).
However, there are circuit applications which do not allow
differential operation at the input and output terminals. For
example, the laser driver at the transmitting end of an opticalfibre link must drive the laser diode by a single-ended output.
Moreover, in measuring equipment the users often prefer
single-ended signals at the input and output terminals to simplify the measuring setup. However, as a consequence of
single-ended operation the demand on pulse symmetry has
often not been sufficiently met by bipolar circuits (unless
expensive measures using hybrid components were taken or a
strong degradation of the pulse steepness could be tolerated).
This is confirmed by the eye diagrams generated with the
fastest bipolar laser driver ICs reported to date 12-43 with
maximum data rates up to 11Gbit/s [2]. In this Letter a
high-speed silicon bipolar driver IC, operating up to I2Gbit/s,
is presented which does not have this disadvantage. The
circuit principle used allows symmetrical single-ended pulses
to be adjusted without degradation of the edge steepness.*
Fig. 1 shows the diagram of the driver circuit which is
mainly provided for driving laser diodes, e.g. via a matched
25R transmission line [l, 51. The circuit configuration is completely symmetrical (differential operation). It is built from two
different stages, each consisting of three emitter follower pairs
(E) followed by a limiting differential emitter-coupled stage
(DS). The currents of both the DS (and thus the voltage
swings) and the currents through the emitter follower pair E6
are off-chip adjustable. This also holds for the input offset
voltages Vos, and Vos2 of both differential stages. The reasons
for these options are given below. The circuit inputs, which
are matched by on-chip 50R resistors (note R , % s o n ) , are
* For a more detailed description of the theoretical background, see
Reference 5
Vol. 28 No. 14
Fig. 1 Circuit diagram ojdriver IC
The only off-chipcomponents are the potentiometers and external
load ( R L )
For application as a laser driver, an external loading of
R , = 25 R is assumed [I, 51 and the on-chip output resistor is
chosen to be RQ = 75R to halve double reflections. Thus, a
source current through the output stage DS2 of I, = 40mA
results in an output current swing of AIQ = 30mA. By using
the external voltage V,, I, and the AIQ can easily be controlled as required for laser-driver applications. A corresponding deterioration of the eye diagram can be avoided if
simultaneously with I, the current through the driving
emitter-follower stage (E6) is changed (see Fig. 1). As confirmed by measurements, no speed degradation occurs within
the intended control range of 20mA II, I53mA (15mA 5
AI2 5 40mA).
The circuit was fabricated with the bipolar part of a selfaligned double-polysilicon BiCMOS technology characterised
by 0.8pm lithography which results in a minimum effective
emitter width of 0.4pm and a transit frequency of the transistors of -25GHz (VcE= IV) [SI. However, it should be
noted that the minimum emitter width is not necessarily the
optimum; e.g. for output transistors TQ and T,.a n d for the
transistor current sources, the effective emitter width was
doubled (i.e. 0.8 pm). The total power consumption at a single
supply voltage of 5 V is 1.2 W (including the external load).
For completeness, some measurement results are presented
before the measures used for achieving symmetrical pulse
shapes are discussed. In all cases laser driver application was
assumed ( R L = 25R, R, = 750) with I, = 40mA. The chip
was mounted unpacked on a low-dielectric substrate and
driven by a pulse pattern generator.
Fig. 2 shows three output eye diagrams of the circuit in Fig.
1 measured at IOGbit/s. In Fig. 2a the differential output
voltage up,, is given. The corresponding single-ended output
voltage up is shown in Fig. 2b which demonstrates the asymmetry of the pulse shape. Using simple adjustment of two
external potentiometers as discussed below, a symmetrical
pulse shape of up is obtained without degradation of the edge
steepeness, as demonstrated by Fig. 2c.
It should be noted that in Fig. 2a and b the bases of the
transistors Tn and T Q of the output stage are driven by symmetrical differential signals. Therefore, the asymmetry in Fig.
2b is mainly caused by the transistor junction capacitances
loading the common emitter node. As a consequence, the sum
of collector currents i, and i;. can differ from the constant
source current I,.
Two main reasons for the asymmetrical pulse shape can be
observed (see Fig. 26) [SI:
(1) At the end of the turn-on transient of TQ the parasitic
capacitance at the common emitter node allows i, to exceed
I, resulting in an undershoot of the falling edge of U,. In
contrast, no overshoot in the rising edge of U, is observed,
because the collector current cannot change its sign.
(2) The turn-on delay of Tp (falling edge of up) is longer than
its turn-off delay. This effect, which is well known from tran1295
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