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It can be seen that good performance is achieved over a
wide bandwidth (2-8GHz), in good agreement with classical
l o @ H*G
REF 0 . 0 d0
5.0 d W
The performance may be summarised as :
operating frequency:
coupling :
phase difference:
isolation :
return loss:
5.0 f 3.0 GHz (120% bandwidth)
3.6 f 0.6 dB
90" f 1.5"
20 dB
20 dB
Conclusion: These results should enable us to develop wideband circuits such as mixers or phase shifters using this technology.
The ultimate limitations on performance will be the accuracy of the models used and the minimum dimensions that
can be achieved, with adequate repeatability.
25th August 1992
D. Bourreau, B. Della, E. Daniel, C. Person and S. Toutain (ENST de
Bretagne, BP 832,29285 Erest Cedex, France)
'Multi-layer thick-film technology for microwave integrated circuits'. M.I.O.P. '90 Microwaves and Optronics, 24-26 April 1990,
Stuttgart, Germany
2 PRESSER,A.: 'Interdigitated microstrip coupler design' IEEE Trans.,
1978, MlT-26, (lo),pp. 801-805
3 HP/MDS application software
10.845000000 Ml
Fig. Sa Return loss
S. N. Dunaev and Y. K. Fetisov
Indexina terms: Maqnetostatic waves, YIG-film
The interaction of two magnetostatic waves (MSW)in a garnet film placed in a harmonically-modulated external magnetising field is experimentally investigated. An MSW
frequency shift of up to MOMHz and interaction efficiency of
up to a few percent were obtained.
Introduction: Magnetostatic wave (MSW) excitation and proe.045ew0.0
Fig. 5 ) Isolation
l o g MAG
REF 0 . 0 dB
Fig. k Coupling and direct ports
pagation in planar structures with nonstationary parameters
based on yttrium-iron-garnet (YIG) films are accompanied by
various dynamic effects, such as MSW amplitude and phase
modulation [l-23, MSW packet compression [SI, and MSW
phase synchronisation [4]. These interactions have already
been applied to microwave signal processing. We report the
first observation of the parametric interaction of two MSWs
in a garnet film placed in an external harmonically modulated
magnetising field.
The experimental setup is shown in Fig. 1. The MSW transmission line contained a YIG film of thickness d = 20ym,
Fig. 1 M S W transmission line
Vol. 28 No. 21
saturation magnetisation 4nM = 1750Gs, uniform resonance
linewidth AH 0.5Oe, prepared by LPE technology on a
gadolinium-gallium-garnet (GGG) substrate of 3 x 25 mmz
dimensions. Two wideband microstrip transducers of 50pm
width deposited on the film surface at a distance y = 0.4cm
apart were used to excite and receive MSWs. The structure
was placed in an external uniform DC magnetising field of
H , = 1.95 kOe applied at an angle of 8,= 17' in the plane
containing the normal to the film and the wave propagation
direction. Field modulation with a frequency F = 20300 MHz and magnitude of up to h = lOOe was applied using
an electromagnetic coil wound on the substrate MSWs were
excited by a continuous electromagnetic (EM) signal of frequency f = 3-4GHz and power P = 1 mW. The spectra of
reflected P , ( f ) and transmitted P 2 ( f )signals were investigated
by using a directional coupler and a narrowband electronically tuned filter.
as in the transmitted signals [4]. For F 2 100-300MHr.
when f belonged to the MSBVW (or MSFVW) branch and
frequencyf+ F in the other branch, the harmonic of the frequencyf+ F ( o r f - F ) appeared only in the reflected signal,
as shown in Fig. 3. The power of the harmonic increased
nonlinearly with an increase in the magnitude of the modulation field.
Experimentul results and discussion: There are two kinds of
MSW in the structure considered: magnetostatic forward
volume wave (MSFVW) and magnetostatic backward volume
wave (MSBVW). Both branches of the dispersion curve are
well described by the simple equation
1 psin2p + cos2D1
Fig. 3 Spectrum of reflected microwuce signal
Horizontal scale: 60 MHz division
\I( - P )
P = 1 + ( f H f M M f ;-f?, fil = Y H ,
f M = YM,
y = 2.8 MHz/Oe, N = 0, 1, 2 . . . is the volume mode number,
and H and 11 are the strength and angle of the internal magnetic field in the film, which differ from those of the external
magnetising field because of demagnetising effects and may be
easily calculated. MSFVW and MSBVW of the parallel wave
vectors have oppositely directed group velocities.
Under stationary conditions, the measured amplitude
response L(f)and dispersion k ( f ) for MSVWs in the transmission line are presented in Fig. 2u and b. The shape of the
amplitude response is determined by the EM MSW conversion efficiency and MSW propagation losses at each frequency. The theoretical dispersion curve (dashed line) was
calculated using eqn. 2 for the main mode ( N = 1) and parameter values H = 0.8 kOe, fi = 45". Propagation of the waves
with maximal wave numbers up to k = 300~x1-' were
observed. Minimal insertion loss was equal to -15 dB for
both MSFVWs and MSBVWs of wave numbers
k 1125"'.
The generation of a lateral harmonic is explained by the
parametric interaction of two MSVWs in a space-uniform
nonstationary medium when the following phase-matching
conditions are satisfied:
f i + F = f 2
where ( R , , f , J and ( R 2 , . f 2are
) the wave vectors and the frequencies of the interacting waves, respectively, and F is the
frequency of the magnetic field modulation. The nonsymmetry
of reflected and transmitted signal spectra may be easily
understood by taking into account the contradirectionality of
the group velocities of the initial and generated waves. The
input signal of frequency f excites a wave in the ferrite film of
wave vector R I , whose power flow is directed to the output
transducer. In the domain between the transducers a wave of
frequencyf+ F ( o r f - F ) and wave vector R, = R I is generated, whose power flow is directed in the opposite direction.
The last wave is registered by the input transducer.
We describe the MSW interaction efficiency by a coefficient
where P , w s , ( f ) and P , M s w ( f TF ) are the powers of the initial
and generated waves near the input transducer, respectively. q
may be approximately expressed in terms of the measured
amplitude response of the transmission line and powers of
microwave signals P(f)and P i ( f k F ) :
-10-30 -50
loss L , dB
m ~ v e n u m b e rk,c"
Fig. 2 Amplitude response and dispersion churacreristlc o / M S W trum-
mission line
a Amplitude response
h Dispersion curves
After modulation, lateral harmonics appeared in both the
reflected and transmitted signal spectra. The power of the
harmonics and their number depended on thefposition in the
dispersion curve and value of F . For F 5 50MHz, whenfwas
placed in the middle part of the MSFVW or MSBVW frequency band, the previously observed harmonics of the frequencies/+ nF (n an integer) appeared in the reflected as well
ELECTRONICS LETTERS 8th October 1992 Vol. 28
No. 21
where L(f)and L(f? F ) should be substituted in relative
units. The dashed line in Fig. 3. shows the dependence of
interaction efficiency q on field modulation frequency F for
k 1 125cm
calculated using eqn. 4 and experimentally
obtained data. Maximal q. as expected, was obtained under
exact synchronisation of the interacting waves. The interaction bandwidth at -3dB level was equal to 6 F = 100MHz;
it decreased to 6 F 5 60MHz with an increase of k. Under
exact synchronisation conditions, q increased with k increasing, that is, with an increase of interaction length y to wavelength i ratio, maximal value q = 3.2% was reached for
y i l = 8 and then decreased for k 2 125cm . The last was
due to the violation of the phase-matching conditions because
of nonuniformity in the internal magnetising field in the film
and a decrease of the modulation field magnitude at higher
Conclusion: We believe that the efficiency of parametric MSW
interaction may be increased up to several tens of a percent by
increasing the modulating field magnitude and by using more
uniform ferrite structures with lower losses. The phenomenon
described is promising for applications in the design of microwave mixers and spectrum transformation devices.
where a = {ak} is a sequence of independent, equiprobable,
M-ary symbols from the set { 4 m ) = 2m - 1 - M: m = 1.2,. . .
M}. The specular component is
10th August 1992
S. N.Dunaev and Y . K.Fetisov (Moscow Institute of Radio Engineering & Electronics and Automation, Vemadskogo 78, 117454 Moscow,
where P. is the average power,
4 k =
+ ak n/M
p(t) is a unit energy pulse and
diffuse component is
D. c., and MOORP. R.: ‘YIG magnetostatic mode serrodyne’,
Proc. IEEE, 1966.54, pp. 685-686
2 REZENDE, s. M., and MORGENTHALW, P. R.: ‘Magnetostaticwaves in
time-varying magnetic fields. I1 Experiment’, J. Appl. Phys., 1969,
40,pp. 537-545
3 PREQBRAZHENSKY, v. L., RYBAKOV, v. P., and m v , Y. IL: ‘Space4
time focusing of magnetostatic wave packet in a nonstationary
medium’, JETP Lett. (Sou.), 1987,46, pp. 114-117
DUNAEV, s. N., and FETISOV, Y. K.: ‘Multimode oscillation and mode
locking of magnetostatic wave delay line oscillator’,Electron. Lett.,
1992,28, pp. 789-791
denotes convolution. The
d(t, a) = -.j(Pd)S(t - td, a)e(t)exp
where P , is the average power, fD is the Doppler frequency
shift, @ t ) is a zero-mean, complex, Gaussian process with
autocorrelation RAT) which for an omnidirectional antenna
used often in land mobile radio is
Here Jo( ) is the zero order Bessel function of first kind, the
bar denotes the average value and the subscript * denotes the
complex conjugate. The noise in eqn. 1 is zero mean, Gaussian. The autocorrelation of &,U ) is
a) = 0.5&
a)p*(t - T , a)
= RAt, T, a)
Indexing t e r m : Phase-shift keying, Mobile radio system,
Satellite relay system
use of pulse shaping to combat multipath propagation
effect in Rician channels is proposed. Because of the time
delay td between the specular and diffuse signal components,
pulse shaping can reduce their correlation. Analysis and
numerical results are given.
Introduction: The received signal of the satellite mobile
channel (SMC) is composed of three parts: a specular component, a diffuse component and noise [I]. It has a Rician
envelope. The diffuse component, caused by multipath scattering, makes the design of a satellite mobile communication
system very challenging. The SMC can be characterised by a
parameter K which is the ratio of powers of the specular and
diffuse components. In a typical situation, the diffuse component makes up 10% of the received signal (K = 10) and
has a delay of up to Sops [2].
In most communication systems, the pulse in the receiver is
matched (both in time and shape) to the transmitted pulse
to achieve a minimum probability of error. This probability is
independent of the kind of pulse used in the system. In the
SMC, although the diffuse signal has the same pulse as the
specular signal, because of time delay, it is only partially correlated with the pulse in the receiver. Different pulses have
different values of this correlation, and therefore we can select
pulse shaping to reduce the effect of diffuse signals.
+ R,(T)
RAT)= N o
x S*(t - td - T, a) exp (2JnfDT)
I p(f)1’ exp Ci2nfT) df
The signal ratio is related to the energy to noise ratio per bit
where E, is the bit energy. We can use the formulas [3,4] to
compute the bit error probability (BEP).
Analysis and numerical results: The BEP is largely affected by
two factors: total value of RAt, 0, a) and intersymbol interference. When both of them are decreased, the performance of
the system in Rician channel is close to that in Gaussian
channel (K = 00). The total value of RAt, 0, a) is determined
by three terms: P,, RdO) and I S(tk- t,, a).1’ P,and RdO) are
fixed for given E,/N,, K and the type of antenna, but we can
reduce the value of I S(tk - t,, U ) 1’ by choosing a proper pulse
shape. Assuming a unit energy pulse, time limited to (-T/2,
T/2),for k = 0, t o = T and 0 It , 5 T , we have
s = Is(t0 - td, a)l’
lexp ( - ; & O ) d - t d )
+ exp ( - J & - l ) d T - td)12
where the first term of eqn. 10 depends on a, and the second
term depends on a- (thus is intersymbol interference) which
can be evaluated from G = g 2 ( T - t,). For binary symbols, we
Fig. 1 Model of receiver
Signal and system: Fig. 1 shows a baseband equivalent model
of the receiver. Assuming the system contains an automatic
carrier tracking circuit and fD, (the maximum Doppler
frequency) is much smaller than the signal bandwidth, the
output of the receiver filter is
= +n/2
s = g’(-t,) + g2(T - t,)
The value of RAC, 0, a) is directly affected by pulse shaping
and independent of the value of symbols. Two pulses used in
the numerical analysis are
pl(t) = J ( I / T ) u A ~ )PA^) = JWT) sin ( n t / T ) U d t )
where U,@) is a rectangular pulse time limited to (- T/2, T/2).
We show BEP, S and G as functions of t d / T for M = 2,
ELECTRONICS LElTERS 8th October 1992
RAt, T, a) = P, R ~ T ) S-( t,,
~ a)
I. Korn and L. Wei
r(t, a) = s(t, a) + p(t, U ) = s(t, a) + d(t, a) + n(t)
A t ) = p(-t)*p(t)
Vol. 28 No. 21
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