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between two different transmission lines. Both transitions
described in this Letter were fabricated at X-band for the proofof-concept demonstrations. A 153IEC-C104 circular waveguide
with an inner diameter of 21mm was chosen as a housing. The
TE,, mode cutoff frequency of the circular waveguide is 8.38GHz.
The aperture size is 5 x 1.5mm2 and the back-short position is L ,
= 12.6mm. A microstrip circuit as shown in Fig. 2 was built on a
O.38mm-thick RT/Duroid 5880 (E, = 2.2) substrate with the following dimensions: the electric probe length is L, = 5.5mm, the
probe width is W, = ISmm, the microstrip line width and length
are 1.13mm ( Z , = 50R) and 18.0mm, respectively. Both type-A
and type-B transitions have the same dimensions. The only difference is the way they are inserted in the circular waveguide.
frequency .GH z
Fig. 4 Measured frequency responses of return loss and insertion loss f b r
back-to-back type-B transition.r wifh Y5mm long circular woveguide,
tn.o lXmm lonx microsfrip lines, and two S M A connectors
Conclusions: Two types of circular waveguide-to-microstrip line
transition were designed using a small and simple structure of a
waveguide-microstrip cross junction. The transitions have more
than 20 dB return loss with a low insertion loss in the design
band. With the advantages of compact configuration, ease of
design, and good performance, these circular wdveguide-to-microstrip transitions will find many applications in microwave and millimetre wave circuits.
0 IEE 1995
5 Drcrmber 1994
Electronics Letters Online No: 19950171
Fig. 2 Substrate dimensions f o r circular waveguide-to-microstrip line
L . Fan, M.-Y. Li and K. Chang (Department nf Electricul En@wering,
Texus A&M Universiry, College station, T X 77843.3128, U S A )
trunsifions in Fig. I
All dimensions in millimetres
SCHNEIDER. M v , GLANCE. B., and BOUIHMANN, W.F: ‘Microwave
and millimeter wave hybrid integrated circuits for radio systems’,
Bell Syst. Techno/. J . , 1969, 48, pp. 1703-1725
VAN HEUVEN, J.H.C : ‘A new integrated waveguide-microstrip
transition’, IEEE Trans., 1976, MTT-24, pp. 144 147
‘Design of waveguide-to-microstrip transition
specially suited to millimeter-wave applications’, Electron. Lett.,
1977, 13, pp. 604605
KNERR. K.H : ‘A new type of waveguide-to-stripline transition’,
IEEE Truns., 1968, MTT-16, pp. 192-194
SHIH. Y c., TON, T N , and
HUI. L.Q.: ‘Waveguide-to-microstrip
transitions for millimeter-wave applications’. 1988 IEEE MTT-S
Int. Microwave Symp. Dig., pp. 473475
frequency , G H z
Fig. 3 Measured frequency responses of return loss and insertion loss
for back-fo-back type-A transifions with Y5mm long circular wavexuide,
two 18mm long microstrip lines, and two S M A connectors
Resulfs und discussiun: To experimentally evaluate the performance of the transition, two transitions of the same type were connected back-to-back. The distance between the two transitions was
95mm. The measurements were made using two standard SMA
connectors and an HP-8510 network analyser. Fig. 3 shows the
measured frequency responses of return loss and insertion loss for
a pair of type-A transitions in the back-to-back configuration. The
return loss is greater than 20dB over a 20% bandwidth from 8.72
to 10.66GHz. The insertion loss is less than 0.5dB in the same frequency range which includes losses in the 95 mm long circular
waveguide, the two 18mm long 50R microstrip lines, the two
SMA coaxial-to-microstrip transitions, and the two transition circuits. The waveguide and microstrip line losses calculated at I O
GHz are 0.014 and 0.19dB, respectively, and the loss estimated for
the two SMA connector transitions is 0.2dB. Therefore the insertion loss for each transition is -0.05dB. The type-B transition has
similar performance as type-A as shown in Fig. 4. The return loss
is greater than 20dB from 9.37 to 11.03GHz. The insertion loss
for a pair of type-B transitions in the back-to-back configuration
is -0.5dB over a bandwidth of 2GHz centred at 10GHz. Comparing the performance of the two types of transitions, the return loss
of type-B is slightly worse than that of type-A. This could be due
to the misalignment of the microstrip probe inserted in the centre
of the longitudinal plane of the circular waveguide.
76th February 1995
Microstrip realisation of loosely coupled
conductors with enhanced directivity
S.R. Mercer
Indexing terms: Strip line components, Direcfional couplers
A planar coupler design to achieve enhanced directivity with
coupling factors < -20dB is presented. The structure is both
simple to design and fabricate. Coupler directivity is improved by
using microstrip gap capacitors to equalise the odd and even
mode phase velocities in the coupled structure. A directivity in
excess of 22dB was achieved over a bandwidth of 18% at 7.7
There are many measurement applications that require the use of
a directional coupler. A coupler directivity of 2 20dB is usually
required to differentiate between transmitted and reflected signals
on a transmission line. It is a simple matter to fabricate coupled
lines in the microstrip medium [ I ] but directivity degrades with
weak coupling. This degradation of directivity in a microstrip coupler can be ascribed to the different phase velocities that arise for
the odd and even modes in the quasi-TEM structure.
Coupler directivity can be improved by equalising the phase
velocities between the odd and even modes. In the past the differ-
Vol. 31
No. 4
ence between the odd mode and even mode phase velocities has
been reduced by using dielectric overlays or by capacitive compensation in a sawtooth coupling structure [2-41. The sawtooth coupling structure is, however, difficult to model due to its complex
geometry. This may result in multiple design iterations before the
desired performance is obtained. Directivity enhancement using
dielectric overlays is mechanically complex and is therefore not
well suited to high volume production.
It has been shown [5] that it is possible to reduce the odd mode
phase velocity to within a few percent of the even mode phase
velocity by connecting a capacitor of value C, across each end of
the coupled line section. The capacitance C, is given by the
CL =
cot ~ , / f4zoo
where f is the coupled line centre frequency, Z,, the odd mode
impedance and
the odd mode phase angle. The capacitance values (C,) required to equalise the odd and even mode phase velocities on microstrip are typically in the sub-picofarad region.
Although it is possible to obtain these small capacitances using
thick-film deposition techniques [6], the use of lumped capacitors
in this capacitance range is impractical. When working in the
microstrip medium it is, however, possible to realise small values
of capacitance with a series gap in a microstrip conductor (gap
capacitor) [7].
A novel planar solution that can be designed using simple
microstrip models is presented here to achieve high directivity with
loosely coupled lines (coupling factor < -20dB).
tee -pnctlon
open CBCUlt
port 3
Fig. 1 Model for enhanced directivity microstrip coupler
The structure shown in Fig. 1 can be simulated using models for
edge coupled parallel lines, tee junctions, gap capacitances and
transmission lines. The discontinuity on the through line from
port 1 to port 2 must be minimised to prevent undue signal reflections on this line. For this reason, the open circuit stubs on the
through line are very short. The stubs on the through line were
included to facilitate the modelling of the gap capacitances.
A coupler was designed using the model shown in Fig. 1 and
was fabricated on 0.635 mm thick alumina (E, = 9.9) with
0.004mm conductor thickness. A layout of this coupler is presented in Fig. 2. The alumina tile was soldered to a gold plated
Kovar carrier plate that was fixed into a test jig equipped with
SMA launchers. A Hewlett Packard 8510C network analyser was
used to measure the coupler’s parameters.
.-. - -.
The curves for measured and predicted coupler directivity are
shown in Fig. 3. A simple parallel line coupler constructed with
50Q lines and without any form of directivity compensation will
have a directivity of -3dB with a coupling factor of -21 dB. It can
be seen from Fig. 3 that a directivity in excess of 22dB was
obtained from 7.0 - 8.4GHz.
Although the predicted frequency of 8.0GHz for maximum
directivity was higher than the measured value, first time design
success was possible with this model. Experimentation with the
models for the gap capacitances has indicated that the predicted
performance is sensitive to small changes in the gap capacitor
model. Work is ongoing to refine the model for the gap capacitors. A manufacturable high directivity microstrip coupler with
adequate bandwidth for many measurement applications was,
however, produced with a single design iteration.
Acknowledgment: The author would like to thank M. Zeeman for
preparing the diagrams for this Letter.
0 IEE 1995
14 December 1994
Electronics Letters Online No: 19950191
S.R. Mercer (Plessey Tellumar South Africa Ltd., 64-74 White Road,
Retreat 7945, South Africa)
and JOHNS, P B.: ‘The design of
coupled microstrip lines’, IEEE Trans., 1975, MTT-23, (6), pp.
SHELEG. B., and SPIELMAN, B.E.: ‘Broad-band directional couplers
using microstrip with dielectric overlays’, IEEE Trans.. 1974,
MTT-22, (12) pp. 121G1220
PODEL. A.F.:‘A high directivity microstrip coupler technique’. IEEE
GMTT Int. Microwave Symp. Dig., 1970, pp. 30-33
DE RONDE, F.c.: ‘Wide-band high directivity in MIC proximity
couplers by planar means’. IEEE M m - S Int. Microwave Symp.
Dig., 1980, pp. 48W82
T.c.: ‘Foundations for microstrip circuit design’ (Wiley,
Chichester, 1981), pp. 151-152
D.c.: ‘Thick-film MIC components in the range 10-18
GHz’. European Microwave Conf., Dig. Papers, Rome,1976, pp.
Fig. 2
GUPTA, c.:
‘Capacitance parameters of
discontinuities in microstriplines’, IEEE Trans., 1978, MTT-26, pp.
16th February 1995
Vol. 31
No. 4
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