This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE Transactions on Industrial Electronics IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS 1 Analysis of a New Single-Stage Soft-Switching Power-Factor-Correction LED Driver with Low DC-Bus Voltage Hosein Khalilian, Hosein Farzanehfard, Member, IEEE, Ehsan Adib, Member, IEEE, Morteza Esteki, Student Member, IEEE Abstract—A new isolated single-stage soft-switching powerfactor-correction (S4 PFC) driver for supplying light-emittingdiodes (LEDs) is introduced in this paper. In the proposed LED driver, the switches voltage stress and also the DC-bus voltage is limited to the peak of the line voltage. Hence, low voltage rated MOSFETs and diodes can be used. The efficiency is improved because the switches are turned on under ZCS condition and turned off under ZVZCS condition. Also, no current feedback is required since the converter provides an output current independent of output voltage. In this paper, operating principle of the proposed LED driver is presented and design considerations are discussed. To verify the theoretical analysis a laboratory prototype of the proposed converter for supplying a 50 W/70 V LED module from 220 Vrms/50 Hz ac mains is implemented, and the experimental results are presented. Since, current feedback is not used, dimming property and operation for universal voltage range is not achievable. Index Terms—light-emitting diodes (LEDs) driver, powerfactor correction (PFC), single stage, soft switching I. INTRODUCTION R ECENTLY light-emitting diodes (LEDs) have increasingly become popular as solid-state lighting sources [1]-[3]. LEDs are well suited for indoor and outdoor lighting applications due to their long life and low maintenance costs. General lighting, architectural lighting, traffic lighting, background lighting of displays, street lighting, automotive and motorcycle lighting, decorative lighting, are some of the LED applications [4]-[8]. In comparison to the traditional high-pressure sodium lamps and high-pressure mercury lamps, LEDs can provide better lighting efficacy, save more energy and also can offer a long lifetime without adding pollution to the environment [8]-[9]. Usually a string of LEDs is used and a driver is required to supply the LED string. This driver is composed of an AC-DC This Manuscript received April 25, 2017; revised June 19, 2017, August 29, 2017; accepted September 21, 2017. Copyright © 2017 IEEE. Personal use of this material is permitted. However, permission to use this material for any other purposes must be obtained from the IEEE by sending a request to pubs-permissions@ieee.org The authors are with the Department of Electrical and Computer Engineering, Isfahan University of Technology, Isfahan 84156-83111, Iran (email: h.khalilian@ec.iut.ac.ir; hosein@cc.iut.ac.ir, e.adib@cc.iut.ac.ir; morteza.esteki@ieee.org). and a DC-DC converter. For the AC-DC converters, there are some regulations for limiting the input current harmonics (such as IEC 61000-3-2) which impose power factor (PF) requirements [10]. In order to meet these requirements, powerfactor correction (PFC) techniques must be used. In the previously presented structures for LED driver, an AC-DC converter is used to form input current for powerfactor correction [11]-[13]. However, since the output voltage of this AC-DC converter has a low frequency ripple (twice of input voltage frequency), a DC-DC stage must be added as the second stage to regulate the voltage. In two stage PFC converters, the power is processed twice and also two different converters are used for this purpose. Therefore, high power density is not achievable using these converters. One method to overcome these problems is to integrate the two stages as a single stage AC-DC converter [14]-[28]. As a result, usually a boost type converter which operates in discontinuous conduction mode (DCM) is integrated with an isolated converter by sharing some of the components especially the switch or switches and provides a driver with improved power factor. However, in these types of drivers in order to attain an input current with low THD, the voltage of the bus capacitor must be higher than the peak of line voltage. Besides, usually the switch current stress in these converters is also high. Therefore, high voltage and current semiconductor components must be used which limits the power level. Besides, high voltage MOSFETS have poor characteristics especially in the case of switch on-resistance which increase the conduction losses. Moreover, since high voltage rated capacitor with large volume is required for the DC bus, the driver power density is low, too. In [14], a new single-switch, single-stage power factor correction (PFC) converter based on discontinuous capacitor voltage mode (DCVM) is presented. The topology proposed in [14] is derived from a DCVM buck converter integrated with a flyback converter and in the converter unlike the conventional single-stage PFC converters, the voltage across the bulk capacitor is low and the input current is continuous. Furthermore, this topology does not have high voltage stress across the bulk capacitor at light loads. However, the converter operates under hard switching condition. Also, in DCVM converters, the input capacitor is charged to at least twice the instantaneous input voltage and the switch voltage 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE Transactions on Industrial Electronics IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS 2 stress is high. In [15], a LED driver consisting of a buck-boost converter and a buck converter is introduced. The buck-boost converter operates in discontinuous-conduction mode (DCM) to perform the function of power-factor correction to ensure almost unity power factor at the input line and the buck converter steps down the output voltage of the buck-boost converter to drive LEDs. In this converter both active switches can operate at zero-voltage switching (ZVS) at turn on by freewheeling the inductor current of the converters to flow through the intrinsic diodes of the MOSFETs. Although, the converter uses many components including active switches, it does not provide isolation. Therefore, providing low output voltages for a string consisting of few LED lamps is difficult which limits the converter application. A single-stage singleswitch soft-switching converter based on boost-flyback power factor correction (PFC) scheme is presented in [16]. Although soft-switching condition is achieved by using passive components, the semiconductor components voltage stresses are much higher than line peak voltage. In [17] a three-level (TL) ac-dc single-stage converter operating with standard phase-shift PWM is proposed. However since the converter uses four active switches, it is not suitable for low power LED driver. In [18], a modified bridgeless PFC AC-DC converter is integrated with a half-bridge-type LLC DC-DC resonant converter to make a single-stage conversion circuit topology for street-lighting applications. The AC-DC resonant driver provides input current shaping, and it offers attributes of lowered switching losses to the soft-switching functions obtained on two power switches and two output-rectifier diodes. However, since the input current shaping PFC is a boost type converter, the voltage stress of the switches are much higher than the peak of line voltage and makes it hard to select proper low cost switches at 220 Vac input voltage applications. An integrated buck–flyback converter is used to provide PFC from a universal ac source for street light LED application in [19]. Although the circuit is simple, the converter suffers from zero crossing problem and high voltage stress. Besides, due to the energy stored in the leakage inductance of the flyback transformer, an extra passive clamp circuit is needed and since the converter is hard switched, the efficiency is poor. A new transformerless single-stage single-switch (S4) converter which integrates a buck-type power factor correction (PFC) cell with a buck-type dc-dc output cell is presented in [20]. The voltage stress across the dc-link capacitor is low and high step-down input-to-output voltage is achieved. However, since the converter is non-isolated, its application may be limited. Besides, it suffers from zero crossing problem. In [21], an integrated double buck–boost PFC converter is proposed as an offline power supply for LED lamps. In [22], which is the isolated version of [21], a buck-boost converter is integrated with a flyback converter. The power factor of the converter is near unity however since both converters operate in DCM, high current stress is imposed on the converter switch especially in higher output applications. A flyback-based parallel PFC is proposed in [23]. A portion of the input power is transferred to the output through a flyback converter and an extra isolated auxiliary circuit based on forward topology is added to the converter to serve as a buffer and a ripple suppression circuit. Although the converter uses a single transformer to lower the size and cost of the converter, it uses two switches operating under hard switching condition and its control circuit is too complicated. The significant advantage of this circuit is that the power is not processed twice. A Single Stage PFC converter with coupled inductors for LED driver as street lighting is presented in [24]. This LED driver integrates a dual buck-boost PFC AC-DC converter with coupled inductors and a half bridge LLC DC-DC resonant converter into a single-stage-conversion circuit topology. Due to DCM operation of the coupled inductors inside the dual buck-boost converter sub-circuit, high power factor is obtained. Moreover, soft switching condition is achieved for two power switches and output rectifier diodes. However, in this converter, the Bus capacitor voltage is much higher than the peak of input line voltage and semiconductor elements voltage stress is very high. A Single-Stage PFC LED Driver Based on SEPIC and LLC circuit is introduced in [25]. A SEPIC converter is integrated with a LLC converter to improve the power factor and regulate the output voltage. One of the LLC converter switches play the role of SEPIC converter switch as well. Although the soft switching condition is achieved and the switching loses are reduced, but the conduction losses are high and the voltage stress of one of the switches is much higher than the input line voltage peak. In [26] and [27], using SEPIC and flyback converters, a single stage LED driver is presented to improve the performance of the system. Although by DCM operation of the SEPIC circuit natural PFC is realized, extra current stress is imposed on the converter switch and also, since the voltage stress of the switch is high, the switch cost is relatively high. Besides, the converter uses too many passive component and is hard switched. A boost-integrated with dual-switch forward ac-dc LED driver with a high power factor and ripple-free output inductor current is analyzed in [28]. In this driver, by adopting a two-switch forward structure, the voltage stress of switches is clamped to the DC bus voltage. However the driver uses two active switches which are hard switched. Also, the converter structure is complex. In [29], design and experimental results of an ac-dc solid state lamp driver based on the asymmetrical half bridge (AHB) flyback converter is presented. The converters operates at variable switching frequency in the range 300-450 kHz, supplying 6W to the load. Although the converter operates under soft-switching condition and have significant volume reduction and consequent power density improvement, it does not have power factor correction capability. To obtain a single stage PFC in a LED driver, the alternative structure, which combines a DC-DC converter with an inductor and a coupled inductor as a current shaper can be considered. By properly selecting a DC-DC converter, many advantages, such as high power factor and minimum DC Bus voltage can be achieved. 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE Transactions on Industrial Electronics IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS To achieve this goal, a half-bridge series resonant converter is selected as a DC-DC converter to use in the LED driver as shown in Fig. 1. The series resonant converter in DCM mode (fres>2.fsw) acts as a constant current source without using any current sensor [30]. In the suggested LED driver, the converter switches are turned on under ZCS condition and turned off under ZVZCS condition. The voltage of the bus capacitor and the switches voltage stress is approximately equal to the peak of input voltage which is much lower as compared to the converters proposed in [24] and [25]. Also, the resonant converter provides an output current independent of output voltage in this mode and therefore, current feedback is not required. 3 Db1 Db3 Vin Lf + S1 VC1 - na2 C1 na1 * Lin Dr1 * Ta Dr3 T Lr Co + Vo - Co + Vo - Cr Cf n1:n2 Db4 Db2 + VC2 - C2 S2 Dr4 Dr2 (a) Db1 S1 Db3 S3 na1 Vin Lf Lin Dr1 na2 * * Ta Cf C Db4 Db2 Lr + VC - S4 Dr3 T Cr n1:n2 S2 Dr4 Dr2 (b) II. PROPOSED LED DRIVER OPERATING PRINCIPLES Fig.2 (a) shows the proposed LED Driver circuit. In the driver, C1 and C2 are the DC bus and also the half bridge converter input capacitors. Cr, Lr, T, Co, Dr1-Dr4 and S1 and S2 are series resonant converter resonant capacitor, resonant inductor, transformer, output capacitor, output rectifier diodes and switches respectively. The input and output voltages are illustrated as Vin and Vo respectively. The inductor Lin and transformer Ta are power factor correction elements. In the proposed driver, a near-unity power factor is achieved by DCM operation of input inductor Lin. Since the input current is discontinuous, a LC filter is added to remove the highfrequency harmonics. For converter operation analysis, illustrate the filtered Vin is shown by Vac in Fig.1. Also, transformer Ta is modeled by an ideal transformer, na1 and na2. The magnetizing inductance Lm and also the leakage inductance of this transformer is Lin. To simplify the analysis, it is assumed that all semiconductor elements are ideal. Also, the value of capacitors C0, C1 and C2 are large enough so that their voltages can be considered constant. Magnetizing inductance of Transformer T is very large so it can be neglected. The switching frequency fs is much larger than the line frequency fl so that the input voltage can be considered fixed in a switching cycle. Also, (fsw/fr)<1/2 and thus, the series resonant half bridge converter operates in DCM mode. The ratio of transformers are considered (n1/n2) = n and (na1/na2) = na. By considering the mentioned assumptions, the proposed converter has 7 distinct modes in a switching cycle at steady state condition. The theoretical waveforms of the converter are shown in Fig. 2 and the converter equivalent circuit in each mode is shown in Fig. 3. It is assumed that the circuit is at steady state and both switches S1 and S2, and diodes Db1~Db4, Dr3 and Dr4 are off. The current iLr equals to na.iLm and na.iLm current is charging the output capacitor. Since Lm is large, its current iLm is considered constant and relatively small. Interval I [t0 – t1]: At t0, the switch S1 is turned on and the voltage VC is applied to na2 and also to the resonant circuit. As a result, S1 current gradually increases from zero and thus, S1 is turned on under ZCS condition. Since, VC1 is applied to na2, (vac+na.VC1–(VC1+VC2)) is placed across Lin and its current increases linearly. At the same time, a resonance starts between Lr and Cr through C1-S1-Lr-Cr-n1-C2. Fig.2 shows the Fig. 1. The proposed LED driver. (a) Half bridge, (b) Full bridge. VGS1 t VGS2 t VS1 IS1 VC1+VC2 t VS2 IS2 VC1+VC2 t ILin Peak ILin t IDb1, IDb2 ILin Peak t ILr t IDr t t0 t4 t5 t6 t7 t1 t2 t3 Interval Interval Interval Interval Interval Interval Interval I II III IV V VI VII Fig. 2. The key waveforms of the proposed half bridge LED deriver during a switching cycle. converter equivalent circuit. When S1 current becomes zero this interval ends. Interval II [t1 – t2]: This mode starts when iLr direction changes. At the beginning of this interval, the rectifying diodes Dr1 and Dr2 are reverse biased and Dr3 and Dr4 are on. 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE Transactions on Industrial Electronics IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS IDr Interval I Db1 Db3 na1 Vac C1 * ILin Lin ILm + S1 VC1 - na2 Dr1 + VCr - * ILr Lm Lr Dr3 T Cr Co + Vo - n1:n2 Db4 Db2 Db1 Db3 C2 + VC2 - C1 + S1 VC1 - na2 na1 * ILin Lin ILm Dr4 Dr2 Dr1 Dr3 IDr Interval II Vac S2 * + VCr ILr Lr Lm T Cr Co + Vo - n1:n2 Db4 Db2 Db1 Db3 C2 + VC2 - C1 + S1 VC1 - na2 Interval III na1 Vac * ILin Lin ILm S2 * Dr2 Dr1 Dr3 IDr + VCr ILr Lr Lm Dr4 T Cr Co + Vo - n1:n2 Db4 Db2 Db1 Db3 C2 + VC2 - C1 + S1 VC1 - na2 na1 * ILin Lin ILm Dr4 Dr2 Dr1 Dr3 * + VCr ILr Lr Lm T Cr Co III. ANALYSIS + Vo - n1:n2 Db4 Db2 Db1 Db3 C2 + VC2 - C1 + S1 VC1 - na2 na1 * ILin Lin ILm S2 Dr4 Dr2 Dr1 Dr3 IDr Interval V Vac + VCr - * ILr Lm Lr T Cr Co + Vo - n1:n2 Db4 Db2 Db1 Db3 C2 + VC2 - C1 + S1 VC1 - na2 S2 Dr4 Dr2 Dr1 Dr3 IDr Interval VI na1 Vac * Lin ILm + VCr - * ILr Lm Lr T Cr Co + Vo - n1:n2 Interval VII Db4 Db2 Db1 Db3 na1 Vac C2 + VC2 - C1 + S1 VC1 - na2 * Lin ILm condition. In this interval, the voltage –VC2 is applied across the resonant circuit, and another resonance between Lr and Cr begins. Also, the voltage (vac–na.VC2 – (VC1+VC2)) is applied across Lin and reduces its current. This interval continues until the direction of Lr current changes. Interval V [t4 – t5]: This mode starts when the direction of Lr current changes and S2 can be turned off under ZVZCS condition. Dr3 and Dr4 are off and Dr1 and Dr2 are on. The current of Lin decreases to zero at the end of this interval. Interval VI [t5 – t6]: During this interval, S2 body diode current decreases and at the end of this interval reaches zero. Interval VII [t6 - t7]: When S2 body diode turns off, this mode begins. In this mode, Lm current flows via na1 and na2 and so, iLr equals to na.iLm. Also, iLr charges Co through the transformer T. Since Lm is large enough, during this interval iLr is approximately constant. In this interval both switches are off and the diodes Db1~Db4, Dr3 and Dr4 are reverse biased. After this interval, one switching period is completed. IDr Interval IV Vac S2 4 S2 Dr2 Dr1 Dr3 IDr + VCr - * ILr Lm Dr4 Lr T Cr Co + Vo - n1:n2 Db4 Db2 C2 + VC2 - S2 Dr4 Dr2 Fig. 3. Equivalent circuit of the proposed half bridge LED driver during a switching cycle. During this interval, S1 current is negative and this switch can be turned off under ZCZVS condition. The current of Lin increases similar to the previous interval. The resonance between Lr and Cr is also similar to the previous interval except that iLr is negative. This interval ends when the S1 body diode current reaches zero. Interval III [t2 – t3]: In this mode, both switches are off. The converter is in this situation until S2 switch is turned on. Interval IV [t3 – t4]: At the beginning of this interval, S2 is turned on. Consequently, the switch current increase from zero in a sinusoidal form and thus, S2 is turned on under ZCS In this section a detailed analysis of the proposed converter is presented and design procedure for its elements are discussed. It is assumed that the line voltage has a sinusoidal waveform as Vin (t) =Vm.sin(ωl.t) where ωl=2.π.fl. According to the proposed driver structure, capacitors C1 and C2 are connected in series and placed after the diode bridge and input inductor, hence VC1+VC2 is equal to the maximum value of the line voltage Vm. It should be noted that na1 voltage is a square wave voltage and its average is zero and when the sum of na1 voltage and input voltage is higher than DC bus voltage, a current pulse is injected to DC bus. This pulse changes the voltage of C1 and C2 negligibly. By assuming VC1=VC2=VC, VC is equal to Vm/2. Depending on the input voltage, two different states are possible for Lin current waveform as shown in Fig. 1. If the input voltage become greater than [Vm+Vx -2.(Vm+Vx).Tr/Ts], the current of Lin become as Fig. 4(a) and if the input voltage is smaller than the mentioned value the current can be depicted as Fig. 4(b). In the figure, Vx is as follows (1) Vx 2.(n.Vo VCr1 ) In Fig. 4(a) the current rate in A1, B1 and C1 areas are as follows Vm dI Lm A1 Vin (na 2). 2 (2) dt Lin dI Lm B1 dt dI Lm C1 Vin Vm Vx Lin Vin (2 na ). (3) Vm 2 . (4) dt Lin In order to have a proper PFC operation, when Vin(t) is zero, ILin should be equal to zero, hence equation (2) can be set equal to zero, as a result, na is obtained as 2. By selecting 2 for na, the current rate equations become as follows 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE Transactions on Industrial Electronics IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS 5 Vin .Tr Vin Vm Vx V 2.Vm T3.( in ) I2 Lin T2 . ILin I1 I2 A1 B1 C1 T3 Tr (½).Ts-Tr ∆T3 ILin I1 A2 B2 ∆T2 (b) T [Vm Vx 2. r .(Vm Vx )] Ts T (b) when Vin< [V V 2. r .(V V )] m x m x Ts (a) when Vin> ILin I1 2 Tr B1 (½).Ts-Tr A1 Tr Fig. 5. The current waveform of Lin at the peak of line voltage. A1 dI Lm B1 dt Vin Lin (5) Vin Vm Vx Lin (6) Vin 2.Vm . (7) dt Lin The current rate of Lin in A2 and B2 areas are similar to A1 and B1, respectively. From (5), (6) and (7), I1 and I2 can be obtained as follows V (8) I in .T dI Lm C1 1 Lin I 2 I1 r Vin Vm Vx Ts .( Tr ) Lin 2 (Vm Vx ).Tr (Vin Vm Vx ). (9) Ts 2 Lin Also, form Fig. 1, ∆T2 and ∆T3 are T2 . Ts 2 . 1 1 Vin (18) .T2 .I1 . .Tr2 . 2 2 (Vm Vx Vin ).Lin In series resonant converter VCr1 is as follows (19) VCr1 2.n.Vo By selecting 1/2 for n, from (2), Vx would be equal to zero. (20) Vx 0 According to (20), equations of Lin current rate in the peak of Vin are as follows: dI Lm A1 Vm (21) dt Lin dI Lm B1 (22) 0 dt dI Lm C1 Vm . (23) dt Lin And the current waveform of ILin at the peak of Vin is shown in Fig. 5. According to Fig. 5, the area of A1, B1 and C1 at peak of Vin are as (24) and (25). V 1 (24) A1 C1 .Tr2 . in 2 Lin V T (25) B1 in .Tr .( S Tr ) . Lin 2 In order to simplify the calculations and present a practical relation for designing Lin it can be assumed that for π/4<θ<3.π/4 the line voltage is near its peak voltage and (24) and (25) can be used to calculate the average of ILin. Also, equation (17) and (18) are used to obtain Lin average current in 0<θ<π/4 and B2 A1 dt (Vm Vx ).Tr (Vin Vm Vx ). T [(Vm Vx ).Tr (Vin Vm Vx ). s ]2 1 1 2 . (16) C1 .T3.I 2 . 2 2 Lin .(2.Vm Vin ) The area under the current waveform in Fig. 4(b) is equal to the sum of A2 and B2, where A2, B2 are as follows: 1 V (17) A2 .Tr2 . in 2 Lin Fig. 4. The current waveform of Lin. dI Lm (12) (13) Vin 2.Vm The area under the current waveform in Fig. 4(a) is equal to the sum of A1, B1 and C1 where A1, B1 and C1 are as follows: 1 V (14) A1 .Tr2 . in 2 Lin 1 T B1 .( s Tr ).( I1 I 2 ) 2 2 (15) Ts (Vin Vx Vm ).Tr (Vin Vm Vx ). 1 T 2] .( s Tr ).[ 2 2 Lin (a) Tr (11) Vin Vm Vx I1 Lin (10) 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE Transactions on Industrial Electronics IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS 3.π/4 <θ<π. Hence, Lin average current during the half period of line voltage is 1 Tr2 Vm T 3 T 2.L . T 1 sin( .t ) . sin( L .t ) 0 t l , .Tl t l 8 8 2 L in s I in avg (t ) Vm .Tr Tl 3.Tl t . sin( L .t ) 2.Lin 8 8 Tl (27) by substituting (26) in (27) the input power average can be obtained as follows 1 Pin Vm2 . .[Vm2 . 2 4 r 1 T . Lin Ts sin 2 . 0 1 .d 1 sin Tr . sin 2 .d ] 2.Lin 4 4 1 2 0 sin . 1 sin .d 3. 2 sin 2 .d ] (29) 4 2 m V .Tr T .[2. r .k1 k 2 ] . 2. .Lin Ts where k1 and k2 are Pin k1 3. 2 1 2 4 k 2 0.5 (30) (31) . (32) 4 Neglecting the converter losses and assuming that the input power average is equal to the output power, Lin can be calculated as V 2 .T T (33) Lin m r .[2. r .k1 k 2 ] 2. .Po Ts Also the average output power is 2 T V (34) I o avg . r . m TS Z o where Lr Cr (35) The capacitor C1 can be selected from the following equation C1 2.Po l .Vm .VC1 (36) The capacitor C2 can be selected as C1. The output capacitor can be chosen like the output capacitor of half bridge series resonant converter as T I o .( s Tr ) 2 (37) Co VCo I Dr avg (28) 3. 2 V 2 .T T Pin m r .[2. r 2. .Lin Ts Zo (26) The input power average is 2 2 Pin . Vin (t ).I in avg (t ).dt TL 0 6 Tr Vm . (38) .Ts Z r IV. EXPERIMENTAL RESULTS In order to verify the theoretical analysis, a laboratory prototype is designed and implemented to supply 50W/70V LED module from 220Vrms/50Hz ac mains. The LED module is composed of series connection of 21×3.3V white LEDs. The nominal switching frequency of 200 kHz is selected. Lin is obtained by using (34) as 570 µH. From (36), capacitors C1 and C2 are obtained as 100 µF and Co is selected 100 µF using (37). The key parameters of the implemented prototype are expressed in Table I. As seen from Fig. 6(a), the input current and voltage waveforms are sinusoidal and in-phase. The voltage waveform of VCr and the current waveform of ILr are shown in Fig. 6(b). As observed from these waveforms, the converter operates in DCM (fsw/fr<1/2). The voltage waveform of the S1 (S2) and the current waveform of Lr are shown in Fig. 6 (c). According to this figure, switches are turned on under ZCS and turned off under ZVZCS condition. The voltage stress on the switches is limited to VC1+VC2 which is approximately equal to the peak of the line voltage. The switching frequency range is changed from 100 to 230 kHz for half to full load and 185 to 265 Vrms input voltage range to test the converter. However, the converter can operate without any feedback with constant switching frequency for limited range of input voltage changes and for also for constant output power load such as LED string. The harmonics of input current under three different line voltages (185Vrms, 220Vrms, and 265Vrms) are measured and depicted in Fig. 7. Also, the IEC 61000-3-2 class C standard is illustrated ZVZCS Turn off ZCS Turn on (a) (b) (c) Fig. 6. The experimental waveforms of the implemented LED driver. (a) Input voltage (top) and current (bottom) waveforms. (Vertical scale is 200 V/div or 0.5 A/div and time scale is 5ms/div), (b) Voltage waveform of Cr (top) and current waveform of Lr (bottom) (Vertical scale is 200 V/div or 5 A/div and time scale is 0.5µs/div), (c) Voltage waveform of S1 (top) and resonant current waveform (bottom) (Vertical scale is 200 V/div or 5 A/div and time scale is 0.5µs/div) 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE Transactions on Industrial Electronics IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS 7 Fig. 8. Implemented prototype converter. TABLE I PARAMETERS OF THE IMPLEMENTED PROTOTYPE Fig. 7. Measured input current harmonics of the implemented prototype under three different line voltages compared with the IEC 61000-3-2 class C standard. in this figure which shows the compliance of the proposed LED driver with this standard. Based on the measured harmonics, at 220Vrms line voltage, the total harmonic distortion (THD) and power factor (PF) of the implemented prototype are 16.43% and 98.0%, respectively. Fig. 8 shows the shows the implemented prototype converter. Fig. 9(a) and (b) show the DC bus voltages for various input voltages and loads, respectively. The efficiency of the implemented prototype converter is reported in Fig 9 (c). The measured efficiency of the implemented prototype at full load is 89.4%. The above conditions are tested in order to verify the converter operation in a closed loop manner. However, the main goal of the paper is to design a converter without any feedback for limited range of input voltage and also fix output power. The proposed converter is not suitable for universal applications due to lack of current feedback in final implementation. Parameter Value Output power (Po) Output voltage (Vo) Output current (Io) Input voltage (Vin) Nominal Switching frequency (fsw) Switches (S1 and S2) Diodes Dr1~Dr4 Diodes Db1~Db4 n na Inductor Lin Magnetizing inductance Lm Inductor Lf Inductor Lr Capacitor Cf Capacitors C1 and C2 Capacitor Cr Capacitor Co 50 W 70 V 700 mA 220 Vrms 200 kHz SKP10N60A BYV28-200 MUR460 0.5 2 570 H 4 mH 1 mH 5 H 220 nF / 400V 100 F / 250 V 6 nF / 800V 100 F / 100 V REFERENCES T. Komine and M. Nakagawa, “Fundamental analysis for visible light communication system using LED lights,” IEEE Trans. Consum. Electron., vol. 50, no. 1, pp. 100–107, Feb. 2004. [2] J. Y. Tsao, “Solid-state lighting: Lamps, chips, and materials for tomorrow,” IEEE Circuits Devices Mag., vol. 20, no. 3, pp. 28–37, May/Jun. 2004. [3] E. F. Schubert, Light-Emitting Diodes. Cambridge, U.K.: Cambridge Univ. 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Hence, lower voltage capacitors with lower volume and also low voltage rating MOSFETs and diodes with low on resistance can be employed. However, due to the single stage structure, large electrolyte capacitors are required to reduce the twice the line frequency bus voltage ripple. Also, since current feedback is not used, dimming property and operation for universal voltage range is not achievable. Hence, the proposed converter is promising for LED driver applications with simple structure and high efficiency 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. 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Poorali and E. Adib, "Analysis of the Integrated SEPIC-Flyback Converter as a Single-Stage Single-Switch Power-Factor-Correction LED Driver," in IEEE Transactions on Industrial Electronics, vol. 63, no. 6, pp. 3562-3570, June 2016. [27] Y. Wang, S. Zhang, J. M. Alonso, X. Liu and D. Xu, “A Single-Stage LED Driver with High Performance Primary-Side-Regulated Characteristic,” IEEE Trans. Circuits and Systems II: Express Briefs , vol.PP, no.99, pp.1-1 [28] S. W. Lee and H. L. Do, “Boost-Integrated Two-Switch Forward AC– DC LED Driver With High Power Factor and Ripple-Free Output Inductor Current,” IEEE Trans. Ind. Electron., vol. 64, no. 7, pp. 57895796, July 2017. [29] S. Buso, G. Spiazzi and F. Sichirollo, “Study of the Asymmetrical HalfBridge Flyback Converter as an Effective Line-Fed Solid-State Lamp Driver,” IEEE Trans. Ind. Electron., vol. 61, no. 12, pp. 6730-6738, Dec. 2014. [30] H. van der Broeck, G. Sauerlander and M. Wendt, “Power driver topologies and control schemes for LEDs,” APEC 07 - Twenty-Second Annual IEEE Applied Power Electronics Conference and Exposition, Anaheim, CA, USA, 2007, pp. 1319-1325. Hosein Khalilian received the B.S. degree in electrical engineering from University of Tabriz, Tabriz, Iran in 2004 and the M.S. degree in electrooptical engineering from Malek Ashtar University of Technology, Shahin Shahr, Iran, in 2008. He is currently working toward the Ph.D. degree at the Department of Electrical and Computer Engineering, Isfahan University of Technology, Isfahan, Iran. His current research interests include power converter analysis, PFC design, LED power supplies and digital control of switching power supplies. Hosein Farzanehfard was born in Isfahan, Iran, in 1961. He received the B.S. and M.S. degrees in Electrical Engineering from the University of Missouri, Columbia, in 1983 and 1985, respectively, and the Ph.D. degree from Virginia Polytechnic Institute and State University, Blacksburg, in 1992. Since 1993, he has been a faculty member in the Department of Electrical and Computer Engineering, Isfahan University of Technology, Isfahan, Iran. 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE Transactions on Industrial Electronics IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS 9 Prof. Farzanehfard is the author or coauthor of more than 150 technical papers published in journals and conference proceedings. His current research interests include highfrequency soft-switching converters, power factor correction, bidirectional converter, active power filters, high-frequency electronic ballasts and pulse power applications. Morteza Esteki (S’16) was born in Isfahan, Iran, in 1989. He received the B.S. degree in Electrical Engineering from the University of Bonab, East Azerbaijan, Iran in 2012 and the M.S. degree in Electrical Engineering for Isfahan University of Technology, Isfahan, Iran in 2015. Since 2015, he has been a research assistant and laboratory engineer in the Department of Electrical and Computer Engineering, Isfahan University of Technology. His research interests are dc-dc converters and power-factor correction ac-dc converters. He received the best master thesis award from IEEE Iran section, 2016. Ehsan Adib was born in Isfahan, Iran, in 1982. He received the B.S., M.S., and Ph.D. degrees in electrical engineering from the Isfahan University of Technology, Isfahan, Iran, in 2003, 2006, and 2009, respectively. He is currently a Faculty Member in the Department of Electrical and Computer Engineering, Isfahan University of Technology. He is the author of more than 100 papers in journals and conference proceedings. His research interests include dc–dc converters and their applications and soft-switching techniques. Dr. Adib received best PhD dissertation award from IEEE Iran section, 2010. 0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

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