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This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
1
Analysis of a New Single-Stage Soft-Switching
Power-Factor-Correction LED Driver with Low
DC-Bus Voltage
Hosein Khalilian, Hosein Farzanehfard, Member, IEEE, Ehsan Adib, Member, IEEE, Morteza Esteki,
Student Member, IEEE

Abstract—A new isolated single-stage soft-switching powerfactor-correction (S4 PFC) driver for supplying light-emittingdiodes (LEDs) is introduced in this paper. In the proposed LED
driver, the switches voltage stress and also the DC-bus voltage is
limited to the peak of the line voltage. Hence, low voltage rated
MOSFETs and diodes can be used. The efficiency is improved
because the switches are turned on under ZCS condition and
turned off under ZVZCS condition. Also, no current feedback is
required since the converter provides an output current
independent of output voltage. In this paper, operating principle
of the proposed LED driver is presented and design
considerations are discussed. To verify the theoretical analysis a
laboratory prototype of the proposed converter for supplying a
50 W/70 V LED module from 220 Vrms/50 Hz ac mains is
implemented, and the experimental results are presented. Since,
current feedback is not used, dimming property and operation
for universal voltage range is not achievable.
Index Terms—light-emitting diodes (LEDs) driver, powerfactor correction (PFC), single stage, soft switching
I. INTRODUCTION
R
ECENTLY light-emitting diodes (LEDs) have
increasingly become popular as solid-state lighting
sources [1]-[3]. LEDs are well suited for indoor and outdoor
lighting applications due to their long life and low
maintenance costs. General lighting, architectural lighting,
traffic lighting, background lighting of displays, street
lighting, automotive and motorcycle lighting, decorative
lighting, are some of the LED applications [4]-[8]. In
comparison to the traditional high-pressure sodium lamps and
high-pressure mercury lamps, LEDs can provide better
lighting efficacy, save more energy and also can offer a long
lifetime without adding pollution to the environment [8]-[9].
Usually a string of LEDs is used and a driver is required to
supply the LED string. This driver is composed of an AC-DC
This Manuscript received April 25, 2017; revised June 19, 2017, August
29, 2017; accepted September 21, 2017.
Copyright © 2017 IEEE. Personal use of this material is permitted.
However, permission to use this material for any other purposes must be
obtained from the IEEE by sending a request to pubs-permissions@ieee.org
The authors are with the Department of Electrical and Computer
Engineering, Isfahan University of Technology, Isfahan 84156-83111, Iran (email: h.khalilian@ec.iut.ac.ir; hosein@cc.iut.ac.ir, e.adib@cc.iut.ac.ir;
morteza.esteki@ieee.org).
and a DC-DC converter. For the AC-DC converters, there are
some regulations for limiting the input current harmonics
(such as IEC 61000-3-2) which impose power factor (PF)
requirements [10]. In order to meet these requirements, powerfactor correction (PFC) techniques must be used.
In the previously presented structures for LED driver, an
AC-DC converter is used to form input current for powerfactor correction [11]-[13]. However, since the output voltage
of this AC-DC converter has a low frequency ripple (twice of
input voltage frequency), a DC-DC stage must be added as the
second stage to regulate the voltage. In two stage PFC
converters, the power is processed twice and also two different
converters are used for this purpose. Therefore, high power
density is not achievable using these converters.
One method to overcome these problems is to integrate the
two stages as a single stage AC-DC converter [14]-[28]. As a
result, usually a boost type converter which operates in
discontinuous conduction mode (DCM) is integrated with an
isolated converter by sharing some of the components
especially the switch or switches and provides a driver with
improved power factor. However, in these types of drivers in
order to attain an input current with low THD, the voltage of
the bus capacitor must be higher than the peak of line voltage.
Besides, usually the switch current stress in these converters is
also high. Therefore, high voltage and current semiconductor
components must be used which limits the power level.
Besides, high voltage MOSFETS have poor characteristics
especially in the case of switch on-resistance which increase
the conduction losses. Moreover, since high voltage rated
capacitor with large volume is required for the DC bus, the
driver power density is low, too.
In [14], a new single-switch, single-stage power factor
correction (PFC) converter based on discontinuous capacitor
voltage mode (DCVM) is presented. The topology proposed in
[14] is derived from a DCVM buck converter integrated with a
flyback converter and in the converter unlike the conventional
single-stage PFC converters, the voltage across the bulk
capacitor is low and the input current is continuous.
Furthermore, this topology does not have high voltage stress
across the bulk capacitor at light loads. However, the
converter operates under hard switching condition. Also, in
DCVM converters, the input capacitor is charged to at least
twice the instantaneous input voltage and the switch voltage
0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
2
stress is high. In [15], a LED driver consisting of a buck-boost
converter and a buck converter is introduced. The buck-boost
converter operates in discontinuous-conduction mode (DCM)
to perform the function of power-factor correction to ensure
almost unity power factor at the input line and the buck
converter steps down the output voltage of the buck-boost
converter to drive LEDs. In this converter both active switches
can operate at zero-voltage switching (ZVS) at turn on by
freewheeling the inductor current of the converters to flow
through the intrinsic diodes of the MOSFETs. Although, the
converter uses many components including active switches, it
does not provide isolation. Therefore, providing low output
voltages for a string consisting of few LED lamps is difficult
which limits the converter application. A single-stage singleswitch soft-switching converter based on boost-flyback power
factor correction (PFC) scheme is presented in [16]. Although
soft-switching condition is achieved by using passive
components, the semiconductor components voltage stresses
are much higher than line peak voltage. In [17] a three-level
(TL) ac-dc single-stage converter operating with standard
phase-shift PWM is proposed. However since the converter
uses four active switches, it is not suitable for low power LED
driver. In [18], a modified bridgeless PFC AC-DC converter is
integrated with a half-bridge-type LLC DC-DC resonant
converter to make a single-stage conversion circuit topology
for street-lighting applications. The AC-DC resonant driver
provides input current shaping, and it offers attributes of
lowered switching losses to the soft-switching functions
obtained on two power switches and two output-rectifier
diodes. However, since the input current shaping PFC is a
boost type converter, the voltage stress of the switches are
much higher than the peak of line voltage and makes it hard to
select proper low cost switches at 220 Vac input voltage
applications.
An integrated buck–flyback converter is used to provide
PFC from a universal ac source for street light LED
application in [19]. Although the circuit is simple, the
converter suffers from zero crossing problem and high voltage
stress. Besides, due to the energy stored in the leakage
inductance of the flyback transformer, an extra passive clamp
circuit is needed and since the converter is hard switched, the
efficiency is poor.
A new transformerless single-stage single-switch (S4)
converter which integrates a buck-type power factor correction
(PFC) cell with a buck-type dc-dc output cell is presented in
[20]. The voltage stress across the dc-link capacitor is low and
high step-down input-to-output voltage is achieved. However,
since the converter is non-isolated, its application may be
limited. Besides, it suffers from zero crossing problem.
In [21], an integrated double buck–boost PFC converter is
proposed as an offline power supply for LED lamps. In [22],
which is the isolated version of [21], a buck-boost converter is
integrated with a flyback converter. The power factor of the
converter is near unity however since both converters operate
in DCM, high current stress is imposed on the converter
switch especially in higher output applications.
A flyback-based parallel PFC is proposed in [23]. A
portion of the input power is transferred to the output through
a flyback converter and an extra isolated auxiliary circuit
based on forward topology is added to the converter to serve
as a buffer and a ripple suppression circuit. Although the
converter uses a single transformer to lower the size and cost
of the converter, it uses two switches operating under hard
switching condition and its control circuit is too complicated.
The significant advantage of this circuit is that the power is
not processed twice.
A Single Stage PFC converter with coupled inductors for
LED driver as street lighting is presented in [24]. This LED
driver integrates a dual buck-boost PFC AC-DC converter
with coupled inductors and a half bridge LLC DC-DC
resonant converter into a single-stage-conversion circuit
topology. Due to DCM operation of the coupled inductors
inside the dual buck-boost converter sub-circuit, high power
factor is obtained. Moreover, soft switching condition is
achieved for two power switches and output rectifier diodes.
However, in this converter, the Bus capacitor voltage is much
higher than the peak of input line voltage and semiconductor
elements voltage stress is very high. A Single-Stage PFC LED
Driver Based on SEPIC and LLC circuit is introduced in [25].
A SEPIC converter is integrated with a LLC converter to
improve the power factor and regulate the output voltage. One
of the LLC converter switches play the role of SEPIC
converter switch as well. Although the soft switching
condition is achieved and the switching loses are reduced, but
the conduction losses are high and the voltage stress of one of
the switches is much higher than the input line voltage peak.
In [26] and [27], using SEPIC and flyback converters, a
single stage LED driver is presented to improve the
performance of the system. Although by DCM operation of
the SEPIC circuit natural PFC is realized, extra current stress
is imposed on the converter switch and also, since the voltage
stress of the switch is high, the switch cost is relatively high.
Besides, the converter uses too many passive component and
is hard switched. A boost-integrated with dual-switch forward
ac-dc LED driver with a high power factor and ripple-free
output inductor current is analyzed in [28]. In this driver, by
adopting a two-switch forward structure, the voltage stress of
switches is clamped to the DC bus voltage. However the
driver uses two active switches which are hard switched. Also,
the converter structure is complex.
In [29], design and experimental results of an ac-dc solid
state lamp driver based on the asymmetrical half bridge
(AHB) flyback converter is presented. The converters operates
at variable switching frequency in the range 300-450 kHz,
supplying 6W to the load. Although the converter operates
under soft-switching condition and have significant volume
reduction and consequent power density improvement, it does
not have power factor correction capability.
To obtain a single stage PFC in a LED driver, the
alternative structure, which combines a DC-DC converter with
an inductor and a coupled inductor as a current shaper can be
considered. By properly selecting a DC-DC converter, many
advantages, such as high power factor and minimum DC Bus
voltage can be achieved.
0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
To achieve this goal, a half-bridge series resonant converter is
selected as a DC-DC converter to use in the LED driver as
shown in Fig. 1. The series resonant converter in DCM mode
(fres>2.fsw) acts as a constant current source without using
any current sensor [30]. In the suggested LED driver, the
converter switches are turned on under ZCS condition and
turned off under ZVZCS condition. The voltage of the bus
capacitor and the switches voltage stress is approximately
equal to the peak of input voltage which is much lower as
compared to the converters proposed in [24] and [25]. Also,
the resonant converter provides an output current independent
of output voltage in this mode and therefore, current feedback
is not required.
3
Db1
Db3
Vin
Lf
+
S1
VC1
- na2
C1
na1
*
Lin
Dr1
*
Ta
Dr3
T
Lr
Co
+
Vo
-
Co
+
Vo
-
Cr
Cf
n1:n2
Db4
Db2
+
VC2
-
C2
S2
Dr4
Dr2
(a)
Db1
S1
Db3
S3
na1
Vin
Lf
Lin
Dr1
na2
*
*
Ta
Cf
C
Db4
Db2
Lr
+
VC
-
S4
Dr3
T
Cr
n1:n2
S2
Dr4
Dr2
(b)
II. PROPOSED LED DRIVER OPERATING PRINCIPLES
Fig.2 (a) shows the proposed LED Driver circuit. In the
driver, C1 and C2 are the DC bus and also the half bridge
converter input capacitors. Cr, Lr, T, Co, Dr1-Dr4 and S1 and S2
are series resonant converter resonant capacitor, resonant
inductor, transformer, output capacitor, output rectifier diodes
and switches respectively. The input and output voltages are
illustrated as Vin and Vo respectively. The inductor Lin and
transformer Ta are power factor correction elements. In the
proposed driver, a near-unity power factor is achieved by
DCM operation of input inductor Lin. Since the input current is
discontinuous, a LC filter is added to remove the highfrequency harmonics.
For converter operation analysis, illustrate the filtered Vin is
shown by Vac in Fig.1. Also, transformer Ta is modeled by an
ideal transformer, na1 and na2. The magnetizing inductance Lm
and also the leakage inductance of this transformer is Lin. To
simplify the analysis, it is assumed that all semiconductor
elements are ideal. Also, the value of capacitors C0, C1 and C2
are large enough so that their voltages can be considered
constant. Magnetizing inductance of Transformer T is very
large so it can be neglected. The switching frequency fs is
much larger than the line frequency fl so that the input voltage
can be considered fixed in a switching cycle. Also, (fsw/fr)<1/2
and thus, the series resonant half bridge converter operates in
DCM mode. The ratio of transformers are considered (n1/n2) =
n and (na1/na2) = na. By considering the mentioned
assumptions, the proposed converter has 7 distinct modes in a
switching cycle at steady state condition. The theoretical
waveforms of the converter are shown in Fig. 2 and the
converter equivalent circuit in each mode is shown in Fig. 3.
It is assumed that the circuit is at steady state and both
switches S1 and S2, and diodes Db1~Db4, Dr3 and Dr4 are off.
The current iLr equals to na.iLm and na.iLm current is charging
the output capacitor. Since Lm is large, its current iLm is
considered constant and relatively small.
Interval I [t0 – t1]: At t0, the switch S1 is turned on and the
voltage VC is applied to na2 and also to the resonant circuit. As
a result, S1 current gradually increases from zero and thus, S1
is turned on under ZCS condition. Since, VC1 is applied to na2,
(vac+na.VC1–(VC1+VC2)) is placed across Lin and its
current increases linearly. At the same time, a resonance starts
between Lr and Cr through C1-S1-Lr-Cr-n1-C2. Fig.2 shows the
Fig. 1. The proposed LED driver. (a) Half bridge, (b) Full bridge.
VGS1
t
VGS2
t
VS1
IS1
VC1+VC2
t
VS2
IS2
VC1+VC2
t
ILin Peak
ILin
t
IDb1,
IDb2
ILin Peak
t
ILr
t
IDr
t
t0
t4
t5
t6
t7
t1
t2
t3
Interval Interval Interval Interval Interval Interval Interval
I
II
III
IV
V
VI
VII
Fig. 2. The key waveforms of the proposed half bridge LED deriver during a
switching cycle.
converter equivalent circuit. When S1 current becomes zero
this interval ends.
Interval II [t1 – t2]: This mode starts when iLr direction
changes. At the beginning of this interval, the rectifying
diodes Dr1 and Dr2 are reverse biased and Dr3 and Dr4 are on.
0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
IDr
Interval I
Db1
Db3
na1
Vac
C1
*
ILin Lin ILm
+
S1
VC1
- na2
Dr1
+ VCr -
*
ILr
Lm
Lr
Dr3
T
Cr
Co
+
Vo
-
n1:n2
Db4
Db2
Db1
Db3
C2
+
VC2
-
C1
+
S1
VC1
- na2
na1
*
ILin Lin ILm
Dr4
Dr2
Dr1
Dr3
IDr
Interval II
Vac
S2
*
+ VCr ILr Lr
Lm
T
Cr
Co
+
Vo
-
n1:n2
Db4
Db2
Db1
Db3
C2
+
VC2
-
C1
+
S1
VC1
- na2
Interval III
na1
Vac
*
ILin Lin ILm
S2
*
Dr2
Dr1
Dr3
IDr
+ VCr ILr Lr
Lm
Dr4
T
Cr
Co
+
Vo
-
n1:n2
Db4
Db2
Db1
Db3
C2
+
VC2
-
C1
+
S1
VC1
- na2
na1
*
ILin Lin ILm
Dr4
Dr2
Dr1
Dr3
*
+ VCr ILr Lr
Lm
T
Cr
Co
III. ANALYSIS
+
Vo
-
n1:n2
Db4
Db2
Db1
Db3
C2
+
VC2
-
C1
+
S1
VC1
- na2
na1
*
ILin Lin ILm
S2
Dr4
Dr2
Dr1
Dr3
IDr
Interval V
Vac
+ VCr -
*
ILr
Lm
Lr
T
Cr
Co
+
Vo
-
n1:n2
Db4
Db2
Db1
Db3
C2
+
VC2
-
C1
+
S1
VC1
- na2
S2
Dr4
Dr2
Dr1
Dr3
IDr
Interval VI
na1
Vac
*
Lin ILm
+ VCr -
*
ILr
Lm
Lr
T
Cr
Co
+
Vo
-
n1:n2
Interval VII
Db4
Db2
Db1
Db3
na1
Vac
C2
+
VC2
-
C1
+
S1
VC1
- na2
*
Lin ILm
condition. In this interval, the voltage –VC2 is applied across
the resonant circuit, and another resonance between Lr and Cr
begins. Also, the voltage (vac–na.VC2 – (VC1+VC2)) is applied
across Lin and reduces its current. This interval continues until
the direction of Lr current changes.
Interval V [t4 – t5]: This mode starts when the direction of
Lr current changes and S2 can be turned off under ZVZCS
condition. Dr3 and Dr4 are off and Dr1 and Dr2 are on. The
current of Lin decreases to zero at the end of this interval.
Interval VI [t5 – t6]: During this interval, S2 body diode
current decreases and at the end of this interval reaches zero.
Interval VII [t6 - t7]: When S2 body diode turns off, this
mode begins. In this mode, Lm current flows via na1 and na2
and so, iLr equals to na.iLm. Also, iLr charges Co through the
transformer T. Since Lm is large enough, during this interval iLr
is approximately constant. In this interval both switches are
off and the diodes Db1~Db4, Dr3 and Dr4 are reverse biased.
After this interval, one switching period is completed.
IDr
Interval IV
Vac
S2
4
S2
Dr2
Dr1
Dr3
IDr
+ VCr -
*
ILr
Lm
Dr4
Lr
T
Cr
Co
+
Vo
-
n1:n2
Db4
Db2
C2
+
VC2
-
S2
Dr4
Dr2
Fig. 3. Equivalent circuit of the proposed half bridge LED driver during a
switching cycle.
During this interval, S1 current is negative and this switch
can be turned off under ZCZVS condition. The current of Lin
increases similar to the previous interval. The resonance
between Lr and Cr is also similar to the previous interval
except that iLr is negative. This interval ends when the S1 body
diode current reaches zero.
Interval III [t2 – t3]: In this mode, both switches are off. The
converter is in this situation until S2 switch is turned on.
Interval IV [t3 – t4]: At the beginning of this interval, S2 is
turned on. Consequently, the switch current increase from zero
in a sinusoidal form and thus, S2 is turned on under ZCS
In this section a detailed analysis of the proposed converter
is presented and design procedure for its elements are
discussed. It is assumed that the line voltage has a sinusoidal
waveform as Vin (t) =Vm.sin(ωl.t) where ωl=2.π.fl. According to
the proposed driver structure, capacitors C1 and C2 are
connected in series and placed after the diode bridge and input
inductor, hence VC1+VC2 is equal to the maximum value of the
line voltage Vm. It should be noted that na1 voltage is a square
wave voltage and its average is zero and when the sum of na1
voltage and input voltage is higher than DC bus voltage, a
current pulse is injected to DC bus. This pulse changes the
voltage of C1 and C2 negligibly. By assuming VC1=VC2=VC, VC
is equal to Vm/2. Depending on the input voltage, two different
states are possible for Lin current waveform as shown in Fig. 1.
If
the
input
voltage
become
greater
than
[Vm+Vx -2.(Vm+Vx).Tr/Ts], the current of Lin become as Fig. 4(a)
and if the input voltage is smaller than the mentioned value the
current can be depicted as Fig. 4(b). In the figure, Vx is as
follows
(1)
Vx  2.(n.Vo  VCr1 )
In Fig. 4(a) the current rate in A1, B1 and C1 areas are as
follows
Vm
dI Lm A1 Vin  (na  2). 2
(2)

dt
Lin
dI Lm B1
dt
dI Lm C1

Vin  Vm  Vx
Lin
Vin  (2  na ).
(3)
Vm
2 .
(4)

dt
Lin
In order to have a proper PFC operation, when Vin(t) is zero,
ILin should be equal to zero, hence equation (2) can be set
equal to zero, as a result, na is obtained as 2. By selecting 2
for na, the current rate equations become as follows
0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
5
Vin
.Tr
Vin  Vm  Vx
V  2.Vm
T3.( in
)  I2
Lin
T2 .  
ILin
I1
I2
A1
B1
C1
T3  
Tr
(½).Ts-Tr
∆T3
ILin
I1
A2 B2
∆T2
(b)
T
[Vm  Vx  2. r .(Vm  Vx )]
Ts
T
(b) when Vin< [V  V  2. r .(V  V )]
m
x
m
x
Ts
(a) when Vin>
ILin
I1
2
Tr
B1
(½).Ts-Tr
A1
Tr
Fig. 5. The current waveform of Lin at the peak of line voltage.
A1
dI Lm B1
dt

Vin
Lin
(5)

Vin  Vm  Vx
Lin
(6)
Vin  2.Vm .
(7)
dt
Lin
The current rate of Lin in A2 and B2 areas are similar to A1
and B1, respectively. From (5), (6) and (7), I1 and I2 can be
obtained as follows
V
(8)
I  in .T
dI Lm C1
1
Lin
I 2  I1 


r
Vin  Vm  Vx Ts
.(  Tr )
Lin
2
(Vm  Vx ).Tr  (Vin  Vm  Vx ).
(9)
Ts
2
Lin
Also, form Fig. 1, ∆T2 and ∆T3 are
T2 .
Ts
2 .
1
1
Vin
(18)
.T2 .I1  .
.Tr2 .
2
2 (Vm  Vx  Vin ).Lin
In series resonant converter VCr1 is as follows
(19)
VCr1  2.n.Vo
By selecting 1/2 for n, from (2), Vx would be equal to zero.
(20)
Vx  0
According to (20), equations of Lin current rate in the peak
of Vin are as follows:
dI Lm A1 Vm
(21)

dt
Lin
dI Lm B1
(22)
0
dt
dI Lm C1  Vm
.
(23)

dt
Lin
And the current waveform of ILin at the peak of Vin is shown
in Fig. 5. According to Fig. 5, the area of A1, B1 and C1 at peak
of Vin are as (24) and (25).
V
1
(24)
A1  C1  .Tr2 . in
2
Lin
V
T
(25)
B1  in .Tr .( S  Tr ) .
Lin
2
In order to simplify the calculations and present a practical
relation for designing Lin it can be assumed that for
π/4<θ<3.π/4 the line voltage is near its peak voltage and (24)
and (25) can be used to calculate the average of ILin. Also,
equation (17) and (18) are used to obtain Lin average current in
0<θ<π/4
and
B2 
A1
dt
(Vm  Vx ).Tr  (Vin  Vm  Vx ).
T
[(Vm  Vx ).Tr  (Vin  Vm  Vx ). s ]2
1
1
2 . (16)
C1  .T3.I 2  .
2
2
Lin .(2.Vm  Vin )
The area under the current waveform in Fig. 4(b) is equal to
the sum of A2 and B2, where A2, B2 are as follows:
1
V
(17)
A2  .Tr2 . in
2
Lin
Fig. 4. The current waveform of Lin.
dI Lm
(12)
(13)
Vin  2.Vm
The area under the current waveform in Fig. 4(a) is equal to
the sum of A1, B1 and C1 where A1, B1 and C1 are as follows:
1
V
(14)
A1  .Tr2 . in
2
Lin
1 T
B1  .( s  Tr ).( I1  I 2 )
2 2
(15)
Ts
(Vin  Vx  Vm ).Tr  (Vin  Vm  Vx ).
1 T
2]
 .( s  Tr ).[
2 2
Lin
(a)
Tr
(11)
Vin  Vm  Vx
  I1
Lin
(10)
0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
3.π/4 <θ<π. Hence, Lin average current during the half period
of line voltage is
 1 Tr2
Vm
T 3
T
 2.L . T 1  sin( .t ) . sin( L .t ) 0  t  l , .Tl  t  l
8 8
2
L
 in s
I in avg (t )  

Vm .Tr
Tl
3.Tl
t 
. sin( L .t )

2.Lin
8
8

Tl
(27)
by substituting (26) in (27) the input power average can be
obtained as follows

1
Pin 

 Vm2 .
.[Vm2 .
2 4
r
1 T
.
Lin Ts
 sin
2
.
0
1
.d
1  sin 
Tr
. sin 2  .d ]
2.Lin 
4

4
1
2
0 sin  . 1  sin  .d 
3.
2
 sin
2
 .d ] (29)

4
2
m
V .Tr
T
.[2. r .k1  k 2 ] .
2. .Lin
Ts
where k1 and k2 are
Pin 
k1 
3. 2 
 1
2
4
k 2  0.5 
(30)
(31)
.
(32)
4
Neglecting the converter losses and assuming that the input
power average is equal to the output power, Lin can be
calculated as
V 2 .T
T
(33)
Lin  m r .[2. r .k1  k 2 ]
2. .Po
Ts
Also the average output power is
2 T V
(34)
I o avg  . r . m
 TS Z o
where
Lr
Cr
(35)
The capacitor C1 can be selected from the following equation
C1 
2.Po
l .Vm .VC1
(36)
The capacitor C2 can be selected as C1. The output capacitor
can be chosen like the output capacitor of half bridge series
resonant converter as
T
I o .( s  Tr )
2
(37)
Co 
VCo
I Dr avg 
(28)
3.
2
V 2 .T
T
Pin  m r .[2. r
2. .Lin
Ts
Zo 
(26)
The input power average is
2 2
Pin 
. Vin (t ).I in avg (t ).dt
TL 0
6
Tr Vm .
(38)
 .Ts Z r
IV. EXPERIMENTAL RESULTS
In order to verify the theoretical analysis, a laboratory
prototype is designed and implemented to supply 50W/70V
LED module from 220Vrms/50Hz ac mains. The LED module
is composed of series connection of 21×3.3V white LEDs. The
nominal switching frequency of 200 kHz is selected. Lin is
obtained by using (34) as 570 µH. From (36), capacitors C1
and C2 are obtained as 100 µF and Co is selected 100 µF using
(37). The key parameters of the implemented prototype are
expressed in Table I. As seen from Fig. 6(a), the input current
and voltage waveforms are sinusoidal and in-phase. The
voltage waveform of VCr and the current waveform of ILr are
shown in Fig. 6(b). As observed from these waveforms, the
converter operates in DCM (fsw/fr<1/2). The voltage waveform
of the S1 (S2) and the current waveform of Lr are shown in Fig.
6 (c). According to this figure, switches are turned on under
ZCS and turned off under ZVZCS condition. The voltage
stress on the switches is limited to VC1+VC2 which is
approximately equal to the peak of the line voltage.
The switching frequency range is changed from 100 to 230
kHz for half to full load and 185 to 265 Vrms input voltage
range to test the converter. However, the converter can operate
without any feedback with constant switching frequency for
limited range of input voltage changes and for also for
constant output power load such as LED string. The
harmonics of input current under three different line voltages
(185Vrms, 220Vrms, and 265Vrms) are measured and depicted in
Fig. 7. Also, the IEC 61000-3-2 class C standard is illustrated
ZVZCS Turn off
ZCS Turn on
(a)
(b)
(c)
Fig. 6. The experimental waveforms of the implemented LED driver. (a) Input voltage (top) and current (bottom) waveforms. (Vertical scale is 200 V/div or
0.5 A/div and time scale is 5ms/div), (b) Voltage waveform of Cr (top) and current waveform of Lr (bottom) (Vertical scale is 200 V/div or 5 A/div and time
scale is 0.5µs/div), (c) Voltage waveform of S1 (top) and resonant current waveform (bottom) (Vertical scale is 200 V/div or 5 A/div and time scale is
0.5µs/div)
0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
7
Fig. 8. Implemented prototype converter.
TABLE I
PARAMETERS OF THE IMPLEMENTED PROTOTYPE
Fig. 7. Measured input current harmonics of the implemented prototype under
three different line voltages compared with the IEC 61000-3-2 class C
standard.
in this figure which shows the compliance of the proposed
LED driver with this standard. Based on the measured
harmonics, at 220Vrms line voltage, the total harmonic
distortion (THD) and power factor (PF) of the implemented
prototype are 16.43% and 98.0%, respectively.
Fig. 8 shows the shows the implemented prototype
converter. Fig. 9(a) and (b) show the DC bus voltages for
various input voltages and loads, respectively. The efficiency
of the implemented prototype converter is reported in Fig 9
(c). The measured efficiency of the implemented prototype at
full load is 89.4%.
The above conditions are tested in order to verify the
converter operation in a closed loop manner. However, the
main goal of the paper is to design a converter without any
feedback for limited range of input voltage and also fix output
power. The proposed converter is not suitable for universal
applications due to lack of current feedback in final
implementation.
Parameter
Value
Output power (Po)
Output voltage (Vo)
Output current (Io)
Input voltage (Vin)
Nominal Switching
frequency (fsw)
Switches (S1 and S2)
Diodes Dr1~Dr4
Diodes Db1~Db4
n
na
Inductor Lin
Magnetizing inductance Lm
Inductor Lf
Inductor Lr
Capacitor Cf
Capacitors C1 and C2
Capacitor Cr
Capacitor Co
50 W
70 V
700 mA
220 Vrms
200 kHz
SKP10N60A
BYV28-200
MUR460
0.5
2
570 H
4 mH
1 mH
5 H
220 nF / 400V
100 F / 250 V
6 nF / 800V
100 F / 100 V
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For LED driver application, a single-stage soft-switching
PFC which combines a series resonant half bridge converter
and a transformer and an inductor, is introduced in this paper.
The proposed driver provides an output current independent of
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0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
(a)
8
(b)
(c)
Fig. 9. (a) DC bus voltages for various input voltages, (b) DC bus voltages for various loads, (c) Implemented prototype converter efficiency.
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Hosein Khalilian received the B.S.
degree in electrical engineering from
University of Tabriz, Tabriz, Iran in
2004 and the M.S. degree in electrooptical engineering from Malek Ashtar
University of Technology, Shahin
Shahr, Iran, in 2008. He is currently
working toward the Ph.D. degree at the
Department of Electrical and Computer
Engineering, Isfahan University of Technology, Isfahan, Iran.
His current research interests include power converter
analysis, PFC design, LED power supplies and digital control
of switching power supplies.
Hosein Farzanehfard was born in
Isfahan, Iran, in 1961. He received the
B.S. and M.S. degrees in Electrical
Engineering from the University of
Missouri, Columbia, in 1983 and 1985,
respectively, and the Ph.D. degree from
Virginia Polytechnic Institute and State
University, Blacksburg, in 1992. Since
1993, he has been a faculty member in
the Department of Electrical and Computer Engineering,
Isfahan University of Technology, Isfahan, Iran.
0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
This article has been accepted for publication in a future issue of this journal, but has not been fully edited. Content may change prior to final publication. Citation information: DOI 10.1109/TIE.2017.2764841, IEEE
Transactions on Industrial Electronics
IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS
9
Prof. Farzanehfard is the author or coauthor of more than 150
technical papers published in journals and conference
proceedings. His current research interests include highfrequency soft-switching converters, power factor correction,
bidirectional converter, active power filters, high-frequency
electronic ballasts and pulse power applications.
Morteza Esteki (S’16) was born in
Isfahan, Iran, in 1989. He received the
B.S. degree in Electrical Engineering
from the University of Bonab, East
Azerbaijan, Iran in 2012 and the M.S.
degree in Electrical Engineering for
Isfahan University of Technology,
Isfahan, Iran in 2015. Since 2015, he
has been a research assistant and
laboratory engineer in the Department of Electrical and
Computer Engineering, Isfahan University of Technology. His
research interests are dc-dc converters and power-factor
correction ac-dc converters.
He received the best master thesis award from IEEE Iran
section, 2016.
Ehsan Adib was born in Isfahan, Iran,
in 1982. He received the B.S., M.S.,
and Ph.D. degrees in electrical
engineering
from
the
Isfahan
University of Technology, Isfahan,
Iran, in 2003, 2006, and 2009,
respectively. He is currently a Faculty
Member in the Department of
Electrical and Computer Engineering,
Isfahan University of Technology. He
is the author of more than 100 papers in journals and
conference proceedings.
His research interests include dc–dc converters and their
applications and soft-switching techniques.
Dr. Adib received best PhD dissertation award from IEEE Iran
section, 2010.
0278-0046 (c) 2017 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.
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