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This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination.
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
1
High Power Integrated Photonic W-Band Emitter
Keye Sun , Jesse Moody , Qinglong Li, Steven M. Bowers, Member, IEEE, and Andreas Beling
Abstract— High power W-band integrated photonic emitters
were successfully demonstrated by the integration of modified
unitraveling-carrier photodiodes (PDs) with high directional gain
Vivaldi antennas. An impedance matching network was designed
to achieve conjugate matching between the PD and the antenna
for better power delivery. An effective isotropic radiation power
as high as 5 dBm at 110 GHz was measured for the emitter
with a 5-µm-diameter PD. The impedance matching was shown
to significantly affect the effective isotropic radiation power.
Index Terms— Integrated photonic emitter, modified
unitraveling-carrier (UTC) photodiode (PD), Vivaldi antenna.
I. I NTRODUCTION
T
HE everlasting demand for high-speed data transmission rates is getting greater. Since the low frequency
band is too crowded for high-speed wireless link applications, it becomes inevitable to extend carrier frequencies to
the millimeter-wave (MMW) band above 60 GHz (V-band)
or even to 100 GHz (W-band) [1], [2]. Realization of such
high-speed system in electronics has intrinsic drawbacks,
such as high propagation loss in coaxial cables and narrow
bandwidth of electronic circuits, which is incompatible with
system operation at both V and W frequency bands. Developing MMW wireless links using photonic techniques seems
advantageous due to the low loss of the optical fiber and
inherent large bandwidth of photonic microwave components.
Recently, significant interest has been paid to microwave
photonic wireless emitters that tightly integrate a high-speed
photodiode (PD) and an antenna [2], [3].
To this end, the unitraveling carrier (UTC) PD was proposed
in [5] owing to its excellent high-speed capability. In this
structure, only fast electrons are required to drift across the
depletion region as the slower holes relax within their short
dielectric relaxation time [6]. Consequently, the space charge
effect in the depletion region is reduced as compared with
the conventional p-i-n PD, which suffers from the slow hole
drift velocity [6]. As a result, the bandwidth performance and
the power handling capability of the UTC PD are superior
to that of the p-i-n PD. To further enhance device bandwidth
and power handling capability of the UTC PD, a depleted
region and a quasi-electric field generated by graded doping
are incorporated into the absorber layer to switch the carrier
transport mechanism from slow diffusion to fast drift [7], [8].
Manuscript received May 17, 2017; revised August 24, 2017; accepted
September 10, 2017. (Corresponding author: Keye Sun.)
The authors are with the Department of Electrical and Computer Engineering, University of Virginia, Charlottesville, VA 22904 USA (e-mail:
ks2kz@virginia.edu; jm6ne@virginia.edu; ql2tr@virginia.edu; smb9cz@
virginia.edu; ab3pj@virginia.edu).
Color versions of one or more of the figures in this paper are available
online at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TMTT.2017.2755008
This improved version of the UTC PD is called modified
UTC (MUTC) PD [9]–[14].
The other essential component of the photonic MMW
emitter is the antenna, which converts the electrical current
generated by the PD into an electromagnetic wave. In order
to compensate for the attenuation of MMW radiation in the
atmosphere, antennas with high directional gain are desired
to concentrate the beam in a certain direction. Commercial horn antennas and Si lenses have been used for this
purpose [3], [15]. Although a high directional gain can be
achieved, they are usually bulky in size and not compatible
with large-scale integrated systems. Planar structures, including slot, log-periodic toothed, Vivaldi antenna, and bow-tie
antenna, as well as traveling wave antenna structures are
preferred for the purpose of system miniaturization, planar
integration, and wide bandwidth [3], [4], [16].
In order to achieve high performance as well as reliable
operation, the photonic emitter needs to have high mechanical
strength, good thermal dissipation, and carefully designed
RF characteristics. Among them, the RF characteristics are the
most critical. In previous reports, PDs were directly connected
to the antenna without an impedance matching network. However, in order to reach maximum output power, a conjugate
impedance match between the antenna and the PD should
be achieved. The effect of impedance matching has recently
been discussed in [17] and was shown to have a significant
effect on the radiation power. In this paper, we successfully
demonstrate the W-band photonic emitters by the integration
of the planar Vivaldi antennas with flip-chip bonded highpower high-speed MUTC PDs. The high gain of the Vivaldi
antenna and the usage of a superstrate eliminate the need for
the horn antenna or Si lens. AlN was used as the substrate,
which has high mechanical strength and high thermal conductivity. An impedance matching network was designed to
achieve a conjugate impedance matching between the MUTC
PD and the antenna. Owing to these improvements, a recordhigh effective isotropic radiated power (EIRP) of 5 dBm was
achieved using the planar structure.
II. MUTC PD D ESIGN AND FABRICATION
The epitaxial layer structure and the band diagram of the
MUTC PD are shown in Fig. 1 [14]. A 1000-nm heavily
doped n-type InP layer acts as the n-contact layer. A charge
compensated InP drift layer was then grown on top of the InP
contact layer. The cliff layer is used to maintain a high electric
field in the depleted absorber. The following quaternary layers
are used to smooth the band discontinuity at the heterojunction
interface, and thus avoid charge accumulation at the barriers.
The absorber is composed of depleted and undepleted InGaAs
layers. Layers with step-graded doping levels are used to
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2
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
Fig. 1.
(a) Epilayer structure and (b) band diagram schematic of the
MUTC PD.
generate a quasi-electric field in the undepleted absorber. As a
result, electron transport is facilitated, which further increased
the bandwidth of the PD. The top quaternary and InP layers
were grown to facilitate the hole collection and block the
electrons from entering the anode. Finally, a heavily doped
InGaAs layer is deposited as the p-contact layer.
The PDs were fabricated using a double mesa process flow.
The first mesa was dry-etched to the InP n-contact layer.
The larger n-mesa was then formed by dry-etching down to
the semi-insulating InP substrate. SiO2 surface passivation
was deposited by plasma-enhanced chemical vapor deposition (PECVD). AuGe/Ni/Au and Ti/Pt/Au metal stacks were
used for n- and p-metal, respectively. The top p-contact was
connected to the bonding pads on the InP substrate by an air
bridge. The metal layers and the air bridge were Au-plated
to a thickness of 1.5 μm. A 220-nm-thick SiO2 antireflection
coating was deposited by PECVD on the backside of the PD
to reduce the optical reflection at 1.55-μm wavelength. The
PDs with the mesa diameters of 5, 6, 14, and 20 μm were
fabricated. A schematic and an optical image of the PD are
shown in Fig. 2.
III. A NTENNA AND I MPEDANCE M ATCHING N ETWORK
D ESIGN , FABRICATION , AND I NTEGRATION
The Vivaldi antenna was chosen for its wide RF bandwidth,
highly directive endfire beam, and planar profile, all of which
are suitable for integration into large-scale phased arrays.
The Vivaldi antenna is a tapered slot line antenna, which
has similar impedance characteristics to continuously tapered
slot line matching networks. It presents an approximately real
impedance over a very broad range of frequencies [18].
However, the planar Vivaldi antenna has a distorted radiation pattern due to the substrate on which it is placed. A simple
method for distortion suppression is to place a superstrate
on top of the antenna, which equalizes the electromagnetic
wave propagation velocity in air and substrate [19]. While
this limits the antenna bandwidth, it enables higher directional
gain. The simulated radiation patterns using the commercial
software (ANSYS HFSS) are shown in Fig. 3. The radiation
pattern with superstrate has a narrow main lobe pointing
forward. For comparison, the pattern without superstrate has a
Fig. 2.
the PD.
(a) Cross-sectional schematic and (b) top view optical image of
Fig. 3. Simulated antenna radiation pattern on E-plane at various frequencies (a) with and (b) without superstrate. Insets: simulated 3-D radiation
patterns of the antenna with and without superstrate.
wider main lobe and shows significant distortion. The antenna
design was detailed in [20].
In order to maximize the radiation power from the antenna,
a resonant impedance matching technique was employed
where the dc bias lines for the PD were used in conjunction
with a quarter wavelength RF short circuit to synthesize an
inductance, which resonates out the PD junction capacitance.
This resonance was optimized between 100 and 110 GHz for
maximum extraction of RF power generated by the PD.
AlN was used for the antenna substrate due to its high thermal conductivity, which significantly increases the dc power
dissipation of the PD [21]. The antenna and the impedance
matching network pattern were fabricated on the AlN substrate
using electron-beam metal deposition, photolithography, and
dry etching processes. The PDs were then flip-chip bonded
on the pattern using an Au–Au thermocompression bonding
process at 340 °C. Fig. 4(a)–(c) shows a cross-sectional view,
optical image, and 3-D schematic of the integrated photonic
emitter, respectively.
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SUN et al.: HIGH POWER INTEGRATED PHOTONIC W-BAND EMITTER
3
Fig. 4. (a) Cross-sectional schematic and (b) optical image of the photonic
integrated emitter. (c) 3-D schematic of the integrated photonic emitter.
Fig. 6. (a) Circuit model of flip-chip bonded PDs for S11 fitting. (b) Measured
and fit S11 parameters of 5-, 6-, 10-, and 20-μm-diameter PDs.
Fig. 5.
Experimental setups for (a) antenna gain measurement
and (b) integrated photonic emitter radiation measurement.
two distributed feedback laser diodes (Mitsubishi,
FU-68PDF-5) with wavelengths around 1.55 μm were
heterodyned to generate an optical beat signal with nearly
100% modulation depth. The frequency of the beat signal
was controlled by thermally tuning the wavelength of one
laser diode. The wavelengths of the two laser diodes were
monitored by a multiwavelength meter (Hewlett Packard,
86120C). An erbium-doped fiber amplifier (EDFA) (Keopsys)
was used to amplify the optical signal. The output optical
signal incident on the PD was adjusted by a variable
optical attenuator (Agilent, 81577A) following the EDFA.
A commercial horn antenna (QuinStar, QGH-WPRR00) with
23-dB gain was used as the receiving antenna. A W-band
mixer (Pacific Millimeter, WM harmonic mixer) was used
to convert the W-band RF signal down to an IF frequency
around 2 GHz. The IF signal was fed into a spectrum
analyzer (Agilent, E4440A) from which the power was read.
B. PD Impedance Characterization
IV. E XPERIMENTAL R ESULTS AND D ISCUSSION
A. Experimental Setup
The experimental setups for measuring the antenna gain
and the radiation characteristics of the integrated photonic
emitter are shown in Fig. 5(a) and (b), respectively. For the
antenna gain measurement, a constant CW power was fed
into the antenna from a PNA network analyzer (Keysight,
N5227A) equipped with frequency extender module (Keysight,
N5251AL20). For the photonic emitter radiation measurement,
Knowledge of the PD impedance is critical to the design
of the matching network. To this end, the S11 scattering
parameters of the PDs flip-chip bonded onto the coplanar waveguide (CPW) were measured using a PNA up
to 100 GHz. Parameter fitting was done in the Advanced
Design System (ADS) software using the circuit model shown
in Fig. 6(a). C j , Rs , and L s represent the junction capacitance,
series resistance, and series inductance associated with the
air bridge, respectively. The effect of the CPW structure was
calculated by the electromagnetic simulator in ADS.
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4
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
TABLE I
E XTRACTED C IRCUIT E LEMENTS OF THE PD S
Fig. 6(b) shows the measured and fit S11 results of PDs
with different mesa sizes. The fit parameters were summarized
in Table I. The parameters of the PD with 14-μm diameter
were calculated from the impedance trend line of the PDs
of 5-, 6-, 10-, and 20-μm diameters as discussed in [14].
C. Antenna and Impedance Matching Network Measurement
The schematic of the antenna connected to the impedance
matching network is shown in Fig. 7(a). The RF short is
realized by a quarter wavelength open stub. The transmission
line of length L 1 + L 2 after the RF short is effectively
an inductor to resonate with the PD junction capacitance. The
detailed dimensions are listed in Table II. The S11 scattering
parameters were measured by a PNA network analyzer from
95 to 110 GHz, and the results are shown in Fig. 7(b). Since
the 5- and 6-μm-diameter PDs have capacitive impedances
in this frequency range, the antenna with matching network
should have an inductive input impedance to achieve conjugate
matching. The Vivaldi antennas in designs 1, 2, 3, 4, and 7 are
parallelly connected with longer matching lines (L 1 +L 2 ),
and thus, they are more inductive and closer to the optimum
conjugate impedance matching condition than designs 5 and 6
with shorter matching lines. The real part of the antenna
impedance was estimated to be 50 at around 100 GHz for
design 2.
The directional gain of these antenna designs was measured
at 100 GHz in the far-field at a distance of 57 cm. The gain
was calculated from the path loss, mixer loss, and receiving
horn antenna gain using the Friis equation. Both the simulated
and measured results are shown in Table II. The measured
gain varied from 1.7 to 5.5 dBi. There is some discrepancy
between the simulated and measured results. Potential reasons
for the discrepancy include additional parasitics and a slight
misalignment between the transmitting Vivaldi antenna and
the receiving horn antenna during the measurement. Design 2
has the highest measured gain of 5.5 dBi. The directional
gain of this antenna was measured over the frequency band
from 95 to 110 GHz, and the result is shown in Fig. 8. The
HFSS simulation results are also shown in Fig. 8, and they
reasonably agree with the measurement values. The antenna
gain reaches 3 dBi or more over a bandwidth of 10 GHz.
Finally, design 2 was chosen to be integrated with the PDs due
to its highest measured antenna gain and optimum impedance
matching with the PDs. It should be noted that the S11 of
the 14-μm-diameter PD is inductive instead of capacitive.
Fig. 7. (a) Schematic and dimension of the antenna plus matching network.
(b) S11 scattering parameters for different antennas plus impedance matching
networks and 5-, 6-, and 14-μm-diameter PDs.
To conjugately match its S11, a capacitive matching circuit
should be connected to the PD. The metal lines of a capacitive
matching circuit (L 1 + L 2 ) would be very long, and the
loss associated with it would be large at such high frequency.
So the conjugate matching of the 14-μm-diameter PD was
not pursued. But the same design was also integrated with the
14-μm-diameter PD to verify the importance of impedance
mismatching on the EIRP.
D. EIRP of the Integrated Photonic Emitter
The radiation power at 100 GHz from the emitters with
5- and 14-μm-diameter PDs biased at −3 V and 10-mA photocurrent was measured at different distances from the integrated
photonic emitter. As shown in Fig. 9, the received radiation
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SUN et al.: HIGH POWER INTEGRATED PHOTONIC W-BAND EMITTER
5
TABLE II
A NTENNA D IMENSIONS AND D IRECTIONAL G AIN OF D IFFERENT A NTENNAS AT 100 GHz
Fig. 8. Measured and simulated directional gain of antenna design 2 between
95 and 110 GHz.
Fig. 9.
Received radiation power at 100 GHz from the emitters with
5- and 14-μm-diameter PD at 10-mA photocurrent and −3-V bias voltage
at different distance.
power decreases with increasing distance due to a larger path
loss and enters far-field at 57 cm. The received power for
the emitter with the 5-μm-diameter PD is 8 dB higher than
Fig. 10. EIRP at 100 GHz from the emitters with 5-, 6-, and 14-μm PD
biased at −2 V at different photocurrents.
the one with the 14-μm-diameter PD at the same photocurrent, which can be explained by an improved impedance
matching.
The EIRP at different photocurrents was measured in
the far-field. The results are shown in Fig. 10. The EIRP
increases with increasing photocurrent as expected. The EIRP
from the emitters with the 5- and 6-μm-diameter PDs is
7.5 and 6 dB higher than the power from the emitter with
the 14-μm-diameter PD, respectively. The higher power is
attributed to the better impedance matching between the
5- and 6-μm-diameter PDs and the antenna as shown on
the Smith chart in Fig. 7(b). The input impedance of the
antenna with the matching network appears inductive and
is located at the conjugate loci of the impedance of the
5- and 6-μm-diameter PDs on the Smith chart. On the other
hand, the impedance of the 14-μm-diameter PD appears
capacitive, which is far away from the point of optimum
conjugate matching. As a result, the EIRP of the emitter with
14-μm-diameter PD is around 6 dB lower than the emitter
with 6-μm-diameter PD. The emitter with 5-μm-diameter PD
has an EIRP about 1.5 dB higher than the emitter with
6-μm-diameter PD does. Based on the fact that the impedance
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6
Fig. 11.
EIRP of the integrated photonic emitters with 5-, 6-,
and 14-μm-diameter PDs biased at 10-mA photocurrent from 95 to 110 GHz.
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
Fig. 12. E-plane radiation patterns of the integrated photonic emitters with
6- and 14-μm-diameter PDs, the antenna without PD, and the HFSS simulation results. Inset: experimental setup for radiation pattern measurement.
matching condition for the 5- and 6-μm-diameter PDs is
similar, the possible reason for this difference is that the
5-μm-diameter PD has a slightly higher −3-dB bandwidth
than the 6-μm-diameter PD [14]. The EIRP does not increase
linearly at high photocurrent, which indicates the power compression due to the space charge effect in the PD at high
photocurrent.
E. Radiation Bandwidth of the Integrated Photonic Emitter
The radiation bandwidths of the integrated photonic emitters
were measured by tuning the temperature of one laser diode
to sweep the beat signal frequency from 95 to 110 GHz.
The results are shown in Fig. 11. The EIRP is around
3, −3, and −7 dBm for the photonic emitters with
5-, 6-, and 14-μm-diameter PDs, respectively. The difference
of the EIRP from the emitters with different PDs is due to the
impedance matching condition as shown in Fig. 7(b). Moreover, the EIRP varies by several dB for each photonic emitter
across the frequency band. This variation might result from
the varying input impedance of the antenna plus matching
network. As can be seen in Fig. 7(b), the S11 of the antenna
plus matching network circles near the optimum conjugate
matching loci. Consequently, the impedance matching condition varies across the frequency band, which, in turn, causes
the variation of the EIRP. The −6-dB power bandwidths for
all three emitters are around 10 GHz.
F. Radiation Pattern of the Integrated Photonic Emitter
The E-plane radiation patterns of the integrated photonic
emitters were measured by changing the receiving horn
antenna position in the far-field region. The results are shown
in Fig. 12. Also shown in Fig. 12 is the E-plane radiation
pattern of the Vivaldi antenna only for comparison. All measurements revealed a half main lobe of about 15°. As expected,
the antenna radiation pattern does not change when the PD
is integrated. The HFSS simulation of the antenna radiation
pattern on E-plane is included in Fig. 12 for comparison. The
simulation and measurement results agree reasonably well.
Fig. 13. (a) Equivalent circuit model for calculating the EIRP of the integrated
photonic emitter. (b) Alternative equivalent circuit model for calculating
the EIRP.
V. C OMPARISON W ITH S IMULATION R ESULTS
An equivalent circuit model was used to calculate the
EIRP of the integrated photonic emitter in the ADS software.
The circuit model is shown in Fig. 13(a). The PD model
includes an ideal current source, junction capacitance C j ,
series resistance Rs , and series inductance L s associated with
the air bridge. The carrier transit time effect in the PD was
taken into account by the fact that the photocurrent is a
function of the beat signal frequency ω, as shown in [22]
I (ω) = I0 ·
1 − exp( j ωtr )
j ωtr
(1)
where I0 is the average photocurrent, ω is the beat signal
frequency, and tr is the electron transit time across the drift
layer, which is calculated to be 3.5 ps [14]. The antenna
plus the impedance matching network were represented by the
measured S11 parameters. Finally, the EIRP was calculated by
the total power delivered to the load multiplied by the antenna
gain (5.8 dB) as a function of frequency.
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SUN et al.: HIGH POWER INTEGRATED PHOTONIC W-BAND EMITTER
Fig. 14. Measurement and ADS calculation results of the EIRP of integrated
photonic emitters with (a) 5-μm, (b) 6-μm, and (c) 14-μm-diameter PDs at
10-mA photocurrent.
The ADS calculation and measurement results using the
circuit model in Fig. 13(a) are shown in Fig. 14 for the integrated photonic emitters with 5-, 6-, and 14-μm-diameter PDs.
For the emitters with 5- and 6-μm-diameter PDs, the ADS
calculations were about 10 dB higher than the measurement results. The difference is 4 dB for the emitter with
14-μm-diameter PD.
There are several possible reasons for this discrepancy. First,
since the layout of the antenna substrate is different from the
CPW substrate, different parasitics can be expected once the
PDs flip-chip bonded on the two substrates. In other words,
the parasitics for the PD bonded on the antenna substrate
could be significantly larger than the value extracted from
the S11 parameters of the PD bonded on the CPW substrate.
In order to show the effect of parasitics on the calculated EIRP,
7
an alternative circuit model shown in Fig. 13(b) was used. The
capacitance Cstray represents the contribution from parasitics.
Other circuit elements have the same meaning and values as
the circuit model in Fig. 13(a). A stray capacitance of 30 fF
was assigned to Cstray in the circuit model in Fig. 13(b). As can
be seen from the calculation results in Fig. 14, even a small
parasitic capacitance of only 30 fF can have significant impact
on the EIRP, which can be several decibels lower than without
parasitics.
Second, the S11 of the PD was measured under dark
condition. It has been shown that the input impedance of the
PD shows a significant change under strong light illumination
due to the space charge effect [23]. Like microwave amplifiers, the S-parameters based on small input signal conditions
may not be adequate to characterize the PD. Other more
sophisticated techniques, such as load-pull measurement, may
help to characterize the optimum load for the device. Since
the impedance matching network is designed based on the
fitting parameters of our simple circuit model with small PNA
stimulus signal under dark condition, the actual matching
condition might be worse than the one shown in Fig. 7(b).
Based on the fact that the impedance matching at such high
frequency range is highly sensitive and depends on the knowledge of the accurate impedance of the PD under strong light
illumination deembedded from the CPW structure, a nonlinear
circuit model and a 3-D full-wave modeling will be developed
in the future. In order to show the effect of illuminationinduced impedance change on the EIRP, the circuit model
in Fig. 13(a) with a different junction capacitance C j was
used for calculation. Under strong illumination, the junction
capacitance tends to increase as a result of electric field
screening caused by photogenerated carriers. To illustrate this
effect, we used a light-induced change of junction capacitance
of 3 fF and recalculate the EIRP. As can be seen from the
results in Fig. 14, a difference around 1 dB can be induced by
this 3-fF increase in junction capacitance. A better matching
between the calculation and measurement results can be found
by using the circuit model extracted from S11 parameters
under illumination.
Finally, the circuit model used to represent the MUTC-PD in
this paper is oversimplified. The model includes only junction
capacitance, series resistance, and air-bridge inductance. More
sophisticated circuit models have been used to represent the
MUTC-PDs. In [12], a serial connection of resistance and
capacitance was used to model the carrier transit time effect,
and two separate capacitances were used to represent the
collector and electrical field suffer layer. In [23], two parallelconnected resistance and capacitance are used to represent
the absorber and drift layers. In [24], parallel connections of
resistance and capacitance were used to model the effect of
quaternary grading layers in order to precisely match the magnitude and the phase of the S11 parameters. According to [17],
a circuit model composed of a simple junction capacitance
and a series resistance may lead to substantial disagreement
between calculated and measured radiation power. In order
to show the effect of variations in the circuit model on
calculated EIRP, the circuit model shown in Fig. 13(b) was
used. All the circuit element values were extracted from the
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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
VII. C ONCLUSION
A W-band photonic emitter was successfully demonstrated
by the integration of high-power flip-chip bonded MUTC PD
and Vivaldi antenna. An EIRP as high as 5 dBm has been
achieved at 110 GHz. The impedance matching has a large
effect on the EIRP. The emitter has a wide −6-dB radiation
bandwidth of around 10 GHz. The main lobe of the radiation
pattern is 30°. The compact integrated photonic emitter is
fully compatible with planar integration. The EIRP is among
the highest that have been reported around 100 GHz in the
literature.
Fig. 15. Radiation power of integrated photonic emitters reported in the
literature.
momentum fitting of S11 of the flip-chip bonded PDs as
discussed in Section IV-B. The EIRP was then calculated
using this circuit model with the best fitting parameters and
is shown in Fig. 14. As can be seen, the calculated power is
already around 1 dB less just by using this slightly different
circuit model. As can be expected, the discrepancy between the
calculation and measurement results could be further reduced
if more sophisticated circuit models were used as the literature
suggested.
In summary, the EIRP is sensitive to all the effects, including light-induced impedance change, parasitics, and the circuit
model representing the PD. We believe that a combination
of these effects can explain the discrepancy between the
calculation and measurement results. Other minor effects
include the orientation and the polarization misalignment of
the transmitting and receiving antenna in the far-field region.
VI. C OMPARISON W ITH L ITERATURE R ESULTS
Fig. 15 gives an overview of the radiation power in this
paper and results that have been reported in the literature.
At around 100 GHz, the EIRP from our photonic integrated
emitter is among the highest. In [28], [39], and [41], the
photonic emitters were connected to a horn antenna with high
directional gain. In addition, either teflon or Si lens were used
in [27], [38], [39], and [42] to achieve high directional gain by
focusing the emitted beam. Up to 12-dB additional gain can be
acquired by a Si lens as reported in [27]. In contrast, the emitters in this paper use a planar Vivaldi antenna. As a result,
the footprint is greatly reduced, and the device is fully compatible with planar integration and suitable for phased array
applications. Other types of planar antenna structures were also
used in the literature to avoid the Si lens and horn antenna,
such as Yagi antenna [25], [26], log-periodic antenna [40],
bow-tie antenna [17], [38], [40], slot antenna [27], [42], and
traveling-wave ring radiator [4]. Among all these reported
photonic emitters, the EIRP in this paper is the highest owing
to the optimized impedance matching network and excellent
high-power-handling capability of the MUTC PDs.
ACKNOWLEDGMENT
The authors would like to thank Prof. J. C. Campbell for
fruitful discussions and P. Yao for his help with the flip-chip
bonding process.
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Keye Sun received the B.S. degree in physics
from Shanghai Jiao Tong University, Shanghai,
China, in 2007, and the Ph.D. degree in electrical and computer engineering from the University of Virginia, Charlottesville, VA, USA,
in 2015.
He is currently a Research Scientist with the
Electrical and Computer Engineering Department,
University of Virginia. His current research interests include high-power and high-speed photodetectors, wireless photonic transmitters, high-frequency
antennas, microwave photonics and circuits, Si photonics, solar cells,
Ge-based devices, and high-power laser-based material processing.
Jesse Moody received the B.S. degree from the
University of Southern Florida, Tampa, FL, USA,
in 2014.
He joined the Electrical and Computer
Engineering Department, University of Virginia,
Charlottesville, VA, USA, in 2015, as a Graduate
Researcher. His current research interests include
ultralow power wake-up receivers for IoT
applications and photonically driven radiators.
Qinglong Li was born in Tianjin, China, in 1985.
He received the bachelor’s degree from Nankai University, Tianjin, in 2008, and the M.Sc. degree from
the Rochester Institute of Technology, Rochester,
NY, USA, in 2013. He is currently pursuing the
Ph.D. degree in electrical and computer engineering at the University of Virginia, Charlottesville,
VA, USA.
He worked within the semiconductor manufacturing industry in China for one year and a half. His
current research interests include high-power and
high-speed photodetectors for optical communication applications.
This article has been accepted for inclusion in a future issue of this journal. Content is final as presented, with the exception of pagination.
10
Steven M. Bowers (GS’08–M’12) received the B.S.
degree in electrical engineering from the University
of California at San Diego, La Jolla, CA, USA,
and the M.S. and Ph.D. degrees in millimeter-wave
circuits and systems from the California Institute of
Technology, Pasadena, CA, USA.
He joined the faculty of the Charles L. Brown
Department of Electrical and Computer Engineering, University of Virginia, Charlottesville, VA,
USA, in 2014, where he is currently an Assistant
Professor.
Dr. Bowers is a member of the HKN and TBP. He was the recipient of the
Caltech Institute Fellowship in 2007 and the Analog Devices Outstanding
Student Designer Award in 2009. He was a recipient of the IEEE RFIC
Symposium Best Student Paper Award in 2012, the IEEE IMS Best Student
Paper Award in 2013, and the 2015 IEEE MTT Microwave Prize.
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
Andreas
Beling
received the Dipl.-Phys.
degree (M.S.) in physics from the University of
Bonn, Bonn, Germany, in 2000, and the Dr.-Ing.
degree (Ph.D.) in electrical engineering from the
Technical University of Berlin, Berlin, Germany,
in 2006.
He was a Staff Scientist with the Photonics
Division, Fraunhofer-Institut für Nachrichtentechnik,
Heinrich-Hertz-Institut, HHI, Berlin, from 2001 to
2006, and a Research Associate with the Department
of Electrical and Computer Engineering (ECE),
University of Virginia (UVA), Charlottesville, VA, USA, from 2006 to
2008. He possesses two years of industry experience as a Project Manager
involved in optoelectronic receivers for high-speed fiber optic communication
systems. He became an Assistant Professor with the Department ECE, UVA,
in 2013. He has authored or co-authored over 130 technical papers and 2
book chapters. He holds four patents.
Dr. Beling is a Senior Member of the OSA. He has served on the Technical
Program Committees of the Optical Fiber Communication (OFC) Conference
from 2010 to 2012, the International Conference on Indium Phosphide and
Related Materials in 2014, the Microwave Photonics Conference in 2015 and
2016, and the Integrated Photonics Research conference in 2016. He was a
Technical Program Subcommittee Chair for Subcommittee 8 (Optoelectronic
Devices) at OFC in 2013. Since 2014, he has been an Associate Editor of
the IEEE/OSA J OURNAL OF L IGHTWAVE T ECHNOLOGY.
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