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Temperature behavior of active circuits at microwave frequency

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(pkD 'l7Usis
‘Rfa- Wo.: 5024. 19* January, 2004.
S C Vera, Qujarat University, India.
i
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Th'D ‘Thesis
Teg. 9{p.: 5 0 2 4 .1 9 ,l January, 2004
5 C S e r a , (jujarat U niversity, India
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9{p.. $ 0 2 4 .1 9 th Ja n u a ry, 2004
XV
1-1 5 4
S C 3cra, Qujarat University, India.
iVecCaration
I hereby declare that the thesis entitled “Temperature Behavior o f
Active Circuits at Microwave Frequency” is a genuine record of
research work carried out by me and no part of this thesis has been
submitted by anybody to any University or Institution for the
award of any degree or diploma.
(Subhash Chandra Bera)
Space Applications Centre
Ahmedabad
July, 2006
a><® Thesis
%pg. Oio.: 5024. 19 “ January, 2004.
O e u w ic w t e
This is to certify that the thesis entitled “Temperature Behavior of
Active Circuits at Microwave Frequency ” is a bona-fide record of
research work done by Shri Subhash Chandra Bera, Registration
No.-5024, dated 19th January 2004, in fulfillment of requirement
for the degree of “Doctor of Philosophy” (in Engineering)
submitted to the Gujarat University, Ahmedabad.
I certify that this work has not been submitted by anyone for any
other degree or diploma to any other University/Institution.
(Dr. S B Sharma)
PhD Supervisor
Space Applications Centre
Ahmedabad
July, 2006
ipT.sft. ? F rf/S . B. SHARMA
T Q D a ? ! * / Dy. Director
TJ^TT g ^T T c fr
st
Antenna Systems Area
‘r fW rhcsu.
: \ a : 5024.
January. 2004.
II
ftc W K P <WL,E*DQ‘E M tE fX?
During design, development and characterization o f various solid-state
circuits and subsystems fo r payload projects o f different Indian National
Satellite System (INSAT), it is realized that more understanding is required
to properly operate the microwave semiconductor devices and circuits at
varying temperature environment o f spacecraft. This feeling tempted me to
understand the temperature behavior o f microwave semiconductor active
devices and circuits to develop a concept o f proper temperature
compensation
mechanism,
suitable fo r
airborne
and
space-based
applications. I should express my sincere thanks to our former Director Dr.
K N Sankara, present director Dr. R R Navalgund, Deputy Director, Shri V
K Garg, and Group Director, Shri R V Singh, fo r assignment o f various
payload project works, providing technical support and encouragement to
complete these works.
I feel much delighted in expressing my foremost and deepest gratitude to
my thesis guide Dr. S B Sharma, Deputy Director, Antenna systems Area,
Space Applications Centre (ISRO), fo r his invaluable guidance and whole­
hearted cooperation throughout the thesis work. I greatly feel that without
his encouragement and unconditional support it would not be possible to
complete the work and submit the thesis.
It becomes my great advantage to work in the field o f solid-state circuits
under Shri V K Garg and Shri R V Singh. Their long experience and
expertise in the field o f microwave solid-state circuits and systems helps me
to complete my work within very short time. From Shri V K Garg, I have
learned always to start with a new and better ways rather than in
conventional way, that leads many novel and innovative designs in various
iPA'D Thesis
%eg 3£o„ 5024.19,h January, 2004.
in
S C Sera, gujarat University, India.
microwave fields.
I always get moral support and boost from Shri R V
Singh during the failure in my work and I often learn more from my failure
than my success.
I express my sincere thanks to Shri V K Jain, Head Channel Amplifier
Division (CAD) and Shri Virender Kumar for their invaluable help and
guidance.
I am one o f the rare persons supported in various ways and encouraged
by my colleagues about the completion of my thesis work. I would like to
mention some of them, Mr. P S Bhardhwaj, Mr. R P Doshi, Mr. S P
Vaishnav, Mr. D R Prajapati, Mrs. Swati A Jaiswal, Mrs. Lina B Shah and
Mrs. Darshna U Raval.
/ would also like to acknowledge Dr. S B Chakrabarty and Mr. B K
Pandey o f Antenna Systems Area, for their valuable suggestions and
discussions during active antenna related works.
I should also recognize the generous support of the device assembly and
packaging facility (DAPF) and mechanical fabrication facility (MFF) of
Space Applications Centre that made the work possible by fabricating
different circuits and subsystems.
The section will be incomplete if I do not acknowledge the criticism from
some o f my friends that makes me more sincere, determined and stubborn to
complete my thesis work within scheduled time.
Space Applications Centre
Indian Space Research organization
Th'D Thesis
%fg. 'Ho.: 5024. 19li January, 2004.
IV
'
Bio-data o f the ftuthor
S ubhash Chandra Bera was bom on 19th September 1968 at Rampur,
Midnapur, West Bengal, India. He received B. Sc. in Physics (Honors) from
Presidency College, Calcutta, B. Tech, and M. Tech, in Radio Physics and
Electronics from Institute of Radio Physics and Electronics, University of
Calcutta, in 1989, 1992 and 1994 respectively. Since 1994, he has been with
the Space Applications Centre, Indian Space Research Organization,
Ahmedabad, India. He has been involved in the design and development of
various
solid-state
power
amplifier,
channel
amplifier
and
other
communication subsystems working in different INSAT and GSAT series of
communication payloads. His field of interest is related to solid-state power
amplifier, channel amplifier, linearizer and attenuator. He has several
publications in international journals related to design, development and
temperature compensation of different payload subsystems. He is also an
inventor of driver circuits for temperature compensation of diode based RF
circuits and light emitting diodes that are applied for international patent.
List o f Publications
[1]
S B Sharma, S C Bera, V K Garg, R V Singh, S B Chakrabarty, “A Digital Beam-forming
Network for Satellite Applications,” Manuscript under preparation.
[2]
S B Sharma, S C Bera, S B Chakrabarty “Active Antenna for Improved Efficiency and
Reduced Harmonic Radiation”. Communicated.
[3]
S C Bera, R V Singh, V K Garg, S B Sharma “Design Issues of Solid-State Power
Amplifier Integrated with Planar Antenna,” Communicated.
[4]
S C Bera, R V Singh, V K Garg, S B Sharma “Optimum Bias Load-Line Compensates
Temperature Variation of Junction Diodes RF Resistance,” Communicated.
[5]
S C Bera, R V Singh, V K Garg, ‘Temperature Behavior and Compensation of Schottky
Barrier Diode,” Communicated.
<Pfi!D th e sis
%pg. 9(p.: 5024. 19l* January, 2004.
S C 'Bera , (yujarat U niversity, India.
V
[6]
S C Bera, R V Singh, V K Garg, “A Driver Circuit for Temperature Compensated LED
Brightness,” Applied fo r International Patent.
[7]
S C Bera, R V Singh, V K Garg, “Design and Temperature Compensation of a Ku-Band
Channel Amplifier with ALC for a Satellite Transponder,” Microwave Journal, Vol. 49,
No.-4, April 2006, pp 68-82.
[8]
S C Bera, “Remarks on ‘PAE Improvement and Compensation of Small-Signal Gain Drift
due to Temperature on Power Amplifiers through Active Biasing’,” Microwave and
Optical Technology Letters, Vol.48, No. 4, April 2006, pp 827-828.
[9]
S C Bera, R V Singh, V K Garg, N S Arora & S S Nair, “A Novel PBG Structure for Filter
Application,” Microwave and Optical Technology Letters, Vol.-48, No.-l, Jan.-2006, pp
188-190.
[10] S C Bera, R V Singh, V K Garg, N S Arora & S S Nair, “PBG Effect On A Modified
Wilkinson Power Divider,” In Proc. InCMARS- Dec.- 2005.
[11] S C Bera, R V Singh, V K Garg, “Temperature Behavior and Compensation of Light
Emitting Diode,” IEEE Photonics Technology Letters, Vol.-17, No.-l 1, Nov.-2005, pp
2286-2288.
[12] S C Bera, P S Bhardhwaj, R V Singh “A Control Circuit for Diode Based RF Circuits,”
International Patent, Publication number “WO 2005/013029” Feb. 2005.
[13] S C B era, R V Singh, V K Garg, “A Temperature Dependent PIN Diode Model Leads to
Simple Temperature Invariant Attenuator Circuit,” Microwave Journal, Vol.- 48, No.-02,
Feb-2005, p l0 4 -l 16.
[14] S C Bera, P S Bharadhwaj “Insight into PIN Diode Behavior Leads to Improved Control
Circuit,” IEEE Transactions on Circuits and Systems-H Vol.-52, No.-l, Jan.-2005, pp 1-4.
[15] S C Bera, R V Singh, V K Garg “A compact Ku-Band Linearizer for Space Application,”
In Proc. APMC-Dec. 2004, New Delhi, India.
[16] S C Bera, R V Singh “A Temperature-compensated Closed Loop Overdrive Level
Controller for Microwave Solid-state Power Amplifiers,” Microwave Journal, April -2004,
Vol.- 47, No.- 4, p i 14-122. Also correction in Microwave Journal, June-2004, Vol.-47,
No.-6, pl56.
[17] S C Bera, P S Bhardhwaj, R V Singh, V K Garg “ A Diode Linearizer for Microwave
Power Amplifiers,” Microwave Journal, Nov-2003, Vol.- 46, No.-l 1, ppl02-l 13.
[18] S C Bera, B M Raval, R V Singh, V K Garg “Electronically Controlled Continuously
Variable Four Quadrant Microwave Phase Shifter,” 31st Mid Term Symposium on
Emerging Wireless Communication Technology & Systems, IETE April 8-9,2000.
[19] V K Garg, R V Singh, V K Jain, S C B era “New Amplifier Design Eliminates Chip
Capacitors,” IETE Technical Review, March-April 1999, VoL-16, No.- 6, ppl97-201.
sjc $ $
2*hD Thesis
%pg. 9 {p : 5024 19‘* January, 2004
S C $ e r a , Qujarat ‘U .nivtrsity, India.
VI
S
ummary
Temperature-behavior o f microwave semiconductor devices
and circuits has received little attention even though under
certain conditions effects can be severe. Intent o f the thesis
work is detail understanding o f temperature behavior o f
microwave semiconductor active devices and circuits to
develop a concept o f proper temperature compensation
mechanism,
suitable
fo r
airborne
and
space-based
applications. The thesis work addresses temperature behavior
and compensation technique o f PIN diode, Schottky barrier
diode and FET based microwave circuits. The research work
proposed and demonstrates a novel temperature-compensation
technique, based on optimum bias load line technology, which
compensates temperature behavior o f PIN and Schottky barrier
diode based microwave circuits, which is more accurate,
simpler and more reliable than previous reported circuits. The
work shows that the same technology is also applicable fo r
temperature compensation o f light emitting diode. The thesis
work also presents and demonstrates practical and theoretical
design details o f temperature compensation mechanisms o f
various microwave circuits and subsystems based on MESFET,
HEMT, PIN diode and Schottky barrier diode.
<Bk<D Thesis
2{o. 5024 1S ' * January, 2004
5 C Viera, Qujarat University, India
VII
CONTENTS
Titles
Page No.
Chapter-I
1-11
1.0
Introduction
2
1.1
Statement of problem
8
1.2
Organization of thesis
9
Chapter-O
12 - 74
2.0
Microwave Junction Diodes
13
2.1
PIN Diode
14
2.1.1
Principle of operation
15
2.1.2
Temperature behavior
18
2.1.3
Temperature compensation
20
2.1.3.1
Conventional temperature compensation circuits
20
2.1.3.2
Analog compensation scheme
21
2.1.3.3
Digital compensation scheme
22
2.1.3.4
Specially fabricated PIN diode for temperature invariant RF resistance
24
2.1.3.5
Proposed optimum bias load line scheme for temperature compensation
25
2.1.3.5.1
Temperature dependent PIN diode model
26
2.1.3.5.2
Theoretically achieving equiresistance curve of PIN diode
27
2.1.3.5.3
Accuracy of optimum load line technique over temperature
31
2.1.3.5.4
Experiment and test results
33
2.1.3.5.4.1
Extraction of PIN diode parameter values and Vopt
34
2.1.3.5.4.2
Experimental determination of Vopt
37
2.1.3.5.4.3
Proposed improved control circuit for PIN diode based step attenuator
2.1.4
Conclusion
44
2.2
Schottky Barrier Diode
45
2.2.1
Principle of operation
46
2.2.2
Temperature characteristic
48
2.2.3
Temperature compensation techniques
51
2.2.3.1
Proposed novel temperature compensation technique
51
Tfl'D Thesis
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S C Wera, Qujarat U niversity, India
VIII
38
2.23.1.1
Theoretically determination of optimum bias' load line
53
2.2.3.1.2
Accuracy of optimum load line technique over temperature
54
2.2.3.13
Experiment and test results
55
2.2.4
Conclusion
60
2.3
Light Emitting Diode
61
2.3.1
Principle of operation
62
2.3.2
Temperature characteristic
63
2.3.3
Temperature compensation technique
65
2.3.3.1
Proposed novel temperature compensation technique
66
23.3.1.1
Theoretically determination of optimum bias load line
69
2.33.1.2
Accuracy of optimum load line technique over temperature
69
23.3.1.3
Experiment and test results
72
233.1.4
Proposed LED driver circuit
74
233.1.5
Effect of supply voltage variation
75
2.3.4
Conclusion
75
2.4
Conclusion
76
Chapter-Ill
77-88
3.0
GaAs FETs and HEMTs
78
3.1
Temperature behavior of MESFET and HEMT
79
3.1.1
Temperature effects in DC characteristics
81
3.1.2
Temperature effects in RF characteristics
82
3.1.3
Temperature behavior of HEMTs
84
3.2
Temperature compensation
85
Chapter-IV
90 -139
4.0
Some Microwave Subsystems
91
4.1
92
4.1.1
Temperature Compensated Vector Modulator for Beam Forming
Network
Design approach
4.1.2
Details of circuit design
94
4.1.3
Realization of vector modulator
97
4.1.4
Temperature behavior and compensation of vector modulator
100
4.1.5
Conclusion
102
T fesis
%Pg <H°-: 5024 19t(t January, 2004
s C S e r a , qujarat U niversity, India
rx
93
4.2
Diode Based Linearizer
103
4.2.1
Design and development of a novel diode based linearizer
105
4.2.2
Experiment and tested results of a Ku-band linearizer
109
4.2.3
Evaluation of a S-Band diode linearizer with SSPA
110
4.2.4
Temperature behavior and compensation of diode linearizer
112
4.2.5
Conclusion
115
4.3
A Temperature Compensated OLC Circuit for SSPAs
116
4.3.1
Different OLC scheme
116
4.3.2
Operation of closed loop OLC
117
4.3.3
Temperature behavior of closed loop OLC
119
4.3.4
Temperature compensation of closed loop OLC
121
4.3.5
Circuit realization and test result
123
4.3.6
Conclusion
125
4.4
Temperature Compensated Channel Amplifier with ALC
126
4.4.1
Block schematic of the channel amplifier
127
4.4.2
Realization of individual modules
128
4.4.3
Temperature compensation of the channel amplifier
132
4.4.3.1
Temperature compensation for Fixed Gain Mode (FGM)
132
4.4.3.2
Temperature compensation for Automatic Level Control (ALC) mode
134
4.4.3.3
Temperature compensated step gain control
136
4.4.3.4
Measurement and test results
137
4.4.4
Conclusion
139
4.5
Conclusions
140
Chapter-V
141 -147
5.0
Summary Conclusion and Future Scope
142
5.1
Summary and conclusion
142
5.1.1
Temperature dependency of bandgap potential
143
5.1.2
Temperature dependency of Schottky barrier height
144
5.1.3
Temperature dependency of bias resistor
144
5.1.4
Self-heating and thermal runaway
146
5.2
Future scope o f the work
147
References
148-154
!PIt'D ‘Thesis
9{o 5024 19‘6 January, 2004
S C (Bera , gujarat University, India
X
L IS T O F F IG U R E S
Figure - 2.1.1
Impurity, electric field and space charge profile of PIN diode.
Figure - 2.1.2
Equivalent circuit of PIN diode.
Figure - 2.1.3
Plot of normalized PIN diode resistance versus temperature using carrier
lifetime coefficient m as a parameter.
Figure - 2.1.4
Schematic of a temperature compensation circuit
Figure -2.1.5
Schematic of a digital temperature compensation circuit.
Figure - 2.1.6
The temperature coefficient of carrier lifetime (m) verses junction
capacitance of the diode, diode passivation material as the parameter.
Figure - 2.1.7
Equiresistance curve and load line of PIN diode.
Figure - 2.1.8
Equiresistance curves for different/? values.
Figure - 2.1.9
Equiresistance curves for different m values.
Figure - 2.1.10
Equiresistance curves and load lines for different attenuation settings.
Figure - 2.1.11
A simple diode bias circuit for PIN diode based attenuator.
Figure - 2.1.12
Schematic circuit diagram and simple RF equivalent circuit of the forward
biased beam-lead PIN diode
Figure - 2.1.13
MIC layout & assembly drawing of PIN diode attenuator.
Figure - 2.1.14
Photograph of the PIN diode based attenuator circuit.
Figure - 2.1.15
I-V characteristic of PIN diode (MPND 4005).
Figure - 2.1.16
Attenuation versus bias current of the diode MPND-4005.
Figure - 2.1.17
Attenuation versus frequency at different current and temperature.
Figure - 2.1.18
Measured PIN diode I-V data and load lines to determine Vopt-
Figure - 2.1.19
PIN diode drier circuit for p # (2-m).
Figure - 2.1.20
A simple bias circuit for PIN diode based variable attenuator.
Figure - 2.1.21
PIN diode driver circuit for p = 2-m.
Thesis
%eg.
S 0 2 4 .19“ January, 2004
S C (Bera, Qu-jarat Zlniperstty, India_
XI
Figure 2.1.22
PIN diode driver circuit to eliminate effect of source resistance
Figure 2.1.23
PIN diode driver circuit to eliminate effect of on resistance of switch
Figure
2.2.1
Schottky diode structure.
Figure
2.2.2
Energy band diagram of Schottky diode.
Figure •2.2.3
Depletion layer of Schottky diode.
Figure
2.2.4
Fermi level alignment of Schottky diode.
Figure
2.2.5
RF equivalent circuit of forward biased Schottky diode.
Figure
2.2.6
Simulated RF resistance variation with temperature at fixed current bias
and fixed voltage bias condition.
Figure
2.2.7
I-V characteristic and load lines of Schottky diode in presence of RF
power.
Figure
2.2.8
Equiresistance curve and optimum load line.
Figure
2.2.9
Simple bias circuit of the Schottky diode.
Figure
2.2.10
Schematic circuit diagram of Schottky diode circuit.
Figure
2.2.11
Photograph of the Schottky diode circuit.
Figure
2.2.12
Measured Schottky diode’s V-I data and load lines for different attenuation
at Pm= - 30 dBm.
Figure ■2.2.13
Measured Schottky diode’s V-I data and load lines for different attenuation
at Pm= - 25 dBm.
Figure ■2.2.14 Measured Schottky diode’s V-I data and load lines for different attenuation
at Pm= - 20 dBm.
Figure
2.2.15
Measured S21 variation over RF power level for Von = 0.75 Volts.
Figure •2.3.1
Simulated brightness variation with temperature for fixed voltage bias and
fixed current bias condition for LED.
Figure •2.3.2
Equiintensity curve and load line of LED.
Figure ■2.3.3
Equiintensity curves and load lines for different brightness levels of LED.
Figure
2.3.4
Equiintensity curves for different values of temperature coefficient (!TC).
Figure
2.3.5
Equiintensity curves for different values of p.
Figure
2.3.6
A simple bias circuit of LED.
TAD TAesis
<Hs>■S 0 2 4 .19“ January, 2004.
S C $ e r a , Qujarat U niversity, India .
XII
Figure - 2.3.7
Calculated light intensity variation with temperature.
Figure - 2.3.8
Measured equiintensity curves of the diode 1N6092 over the temperature of
-20 to +80 °C.
Figure - 2.3.9
Measured light intensity variation with temperature of the diode 1N6092.
Figure - 2.3.10 Series and parallel combination of several LEDs to achieve sufficient
brightness.
Figure - 3.1
Typical I-V characteristic of MESFET with change of temperature.
Figure - 3.2
The equivalent circuit model of MESFET and HEMT.
Figure - 3.3
MIC assembly drawing and photograph of a 3-Stage C-band amplifier of
MESFET (NE-13783).
Figure - 3.4
Photograph of a 3-Stage Ku-band amplifier of MESFET(NE67383).
Figure - 354
MIC assembly drawing and photograph of a 3-Stage Ku-band amplifier of
pHEMT (CFY6708).
Figure - 4.1.1
Block schematic of the vector modulator.
Figure - 4.1.2
Schematic circuit of variable attenuator with phase shift
Figure - 4.1.3
S21 variation of PIN diode over diode resistance.
Figure - 4.1.4
RF equivalent circuit of forward biased PIN diode.
Figure - 4.1.5
Simulated S21 plot of the variable attenuator.
Figure - 4.1.6
Schematic circuit diagram of the proposed vector modulator.
Figure - 4.1.7
Simulated S21 of vector modulator
Figure - 4.1.8
Photograph of vector modulators.
Figure - 4.1.9
Measured polar S21 plot of S-band vector modulator
Figure - 4.1.10
Schematic circuit of the temperature compensated analog vector modulator
Figure - 4.1.11
Schematic circuit of the temperature compensated digitally controlled
vector modulator
Figure - 4.2.1
Measured I-V characteristics of Schottky diode in presence of RF power.
Figure - 4.2.2
Simulated RF resistance variation with temperature.
Figure - 4.2.3
Schematic of the proposed diode based linearizer.
Figure - 4.2.4
RF equivalent circuit of forward biased junction diode.
Thesis
%fg 9{p. 5024.19“ January, 2004.
S C “Sera, gujarat University, India
XIII
Figure - 4.2.5
Calculated amplitude and phase of linearizer.
Figure - 4.2.6
MIC layout and assembly diagram of Ku-band linearizer.
Figure - 4.2.7
Photograph of Ku-band linearizer.
Figure - 4.2.8
AM/AM & AM/PM characteristics of the linearizer for different bias
conditions.
Figure - 4.2.9
Photograph of S band linearizer.
Figure - 4.2.10
Amplitude and phase characteristic of SSPA with linearizer.
Figure - 4.2.11
IMD of SSPA with and without linearizer.
Figure - 4.2.12
Measured l&zl variation of distortion generator circuit over RF power
level under constant current bias condition.
Figure - 4.2.13
Measured S21 variation of distortion generator circuit over RF power level
for constant current bias condition.
Figure - 4.2.14
Measured S21 variation with temperature for optimum load biasing.
Figure - 4.3.1
Block diagram of proposed OLC.
Figure - 4.3.2
Photograph of C-Band SSPA used to demonstrate OLC scheme.
Figure - 4.3.3
Closed loop OLC response under different control bias voltages.
Figure - 4.3.4
I-V characteristics of SBD over temperature.
Figure - 4.3.5
OLC response due to temperature variation of the detector diode.
Figure - 4.3.6
Block diagram of OLC circuit.
Figure - 4.3.7
Complete OLC circuit response over the temperature variation.
Figure - 4.4.1
Basic block diagram of the channel amplifier with ALC.
Figure - 4.4.2
Block diagram with control signals.
Figure - 4.4.3
MIC assembly of 3-stage amplifier.
Figure. - 4.4.4
MIC assembly of PIN diode attenuator.
Figure - 4.4.5
Photograph of Ku-band channel amplifier with ALC.
Figure - 4.4.6
Equivalent circuit for
Figure - 4.4.7
Plot to determine Vrj.
Figure - 4.4.8
1-0 characteristic without temperature compensation.
Vh'D Tkests
Vr i .
!Rfy 9{p : S024 IS'* January, 2004
S C (Bern, ^ ajarat ‘University, India.
XIV
Figure - 4.4.9
1-0 characteristic of channel amplifier with temperature compensation.
Figure -5.1
Measurement to consider effect of temperature dependency of analogswitch, resistor, Eg, and fo.
Figure - 5.2
Plot of load line to determine Vqpt-
LIST OF TABLES
Table-2.1.1
Test results of the PIN diode based attenuator for Voft= 1.19 Volts.
Table-2.1.2
Test results of PIN diode based Attenuator (With Mechanical Switch).
Table-2.1.3
Test results of PIN diode based attenuator (With Analog Multiplexer,
CD4051).
T h D Thesis
Keg
5024 13’* January, 2004
S C ‘3 era , Cjujaral ’University, India.
XV
C
SU'D Thesis
hapter
%pg. 9{p.. S024 19'* January, 2004.
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S C S era, (jujarat University, India
-
1
-
1.0
IN T R O D U C T IO N
Well performance in wide-ranging thermal environment is always demanded from
airborne and space-based microwave electronic circuits and systems. The success of
electronic system for these applications relies on the ability to design high performance,
high reliable circuits which function in demanding thermal environments. To accomplish
this, designers require greater understanding about the temperature behavior of
microwave devices and circuits. However, the temperature-behavior of microwave
semiconductor devices and circuits has received little attention even though under certain
conditions effects can be severe.
Performance parameter of microwave diodes and transistors changes with
temperature. Attenuation and phase shift of PIN diode based microwave circuits and
limiting power level, attenuation & phase shift of Schottky barrier diode based circuits
are all functions of temperature. Gain, saturated output power and nonlinear parameters
of GaAs FETs and HEMTs are also changes with temperature.
PIN diode and Schottky barrier diodes are popularly used in various microwave
circuits and systems such as power limiter, variable attenuator, phase shifter, linearizer,
automatic level control system, etc.
PIN diodes have been used for several decades [1] - [7]. This device used for
attenuator, phase shifter and limiter from microwatt to megawatt level [7] of power at the
frequency range of RF, microwave and extended to mm wave range. Though silicon PIN
diodes are more popularly used, GaAs PIN diodes are also available for fast switching
applications [8]. GaAs PIN diodes are more suitable to operate at mm wave frequency
range.
Although extensive work has been done for accurate modeling of the PIN diode to
design high frequency circuits for high power applications [9] - [11], the design of bias
!PH<D ‘Thesis
9{p ■S0Z4 19* January, 2004.
S C Bern, gujarat University, India
-
2
-
networks for electronically controlled PIN diode based attenuators, phase shifters, etc.
has remained something of a black art. This is particularly true when accurate control is
required over a wide operating temperature range. The reason is that the effect of
temperature on the RF resistance of PIN diodes was not well understood in the past In
the early 1980s, Alfa industries Application Note No. 80200 [2] had this to say; “We do
not label the [Temperature versus RF resistance] curves because we do not yet have
enough data and do not want to mislead you. We want you to realize that you will have
to do your own experimenting with your diode in your application.” Indeed, it was not
until 1993 that Caverly and Hiller [12] - [14] showed that the RF resistance of these
diodes varies as the (2-m)-th power of the temperature, where the power “m ” lies
between 0 and 2 for practical PIN diodes. Caverly and Hiller have suggested using
specially fabricated PIN diode of m = 2, to achieve temperature insensitive PIN diode
attenuation [14]. However, this technique is not applicable to large varieties of
commercially available microwave PIN diodes for which m
2. Moreover, the PIN
diodes of m = 2, have very high parasitic capacitance values for oxide passivation and
not workable for high range of attenuation at microwave frequency range [12] - [13].
Therefore, suitable temperature compensation circuit is required to achieve temperature
invariant RF performance of PIN diode based RF circuits.
There are numerous temperature compensation mechanisms for PIN diode based
circuits [15] - [20]. All the conventional compensation circuit uses temperature sensor to
sense the temperature of the diode and produce a signal that is function of temperature.
Among the various compensation mechanisms, some of the compensation mechanism
based entirely on analog control signal [15] - [18], while others contain digital devices
acting at discrete temperature steps [19], [20]. The entire conventional temperature
compensation schemes using temperature sensor having their own disadvantages of
compensation errors due to self-heating of the diodes when they are operated under high
power condition.
Some analog compensation circuit used thermistor [15], [16] as the temperature
sensing elements and other uses diode [17] as the temperature sensor. Analog
compensation circuit sense the ambient temperature of the diodes and produce
temperature dependent control signal. Change of thermistor’s resistance and change of
diode’s voltage drop with the change of temperature produces temperature dependent
Thesis
%fS £V > . 5024 19* January, 2004.
S C <Bera, §ujarat U niversity, India.
-3 -
control signal. In case of analog compensation circuits, temperature dependent control
signal controls diode current continually over the ambient temperature to provide
temperature invariant performance. Some analog compensation circuits are simple,
consisting with only resistive networks with their less accurate performance whereas
some circuits [18] are complicated containing several amplifiers and comparators to
provide performance that is more accurate. Moreover, circuit complexity increases when
many number of step attenuation is required.
Digital compensation circuits also use same type of temperature sensors as used for
analog compensation circuits. Some digital compensation circuit uses ADC/DAC
combination [19] whereas other circuit uses microprocessor based control [20]. Digital
control circuits are more accurate but require more number of components and time
consuming for setting. Microprocessor based compensation circuit [20] uses look-up
tables. With the variation of ambient temperature address of the memory changes and
predetermined data from memory produces control voltage for PIN diode based circuits
to provide temperature invariant performance. Main drawback of the digital
compensation circuit is that the performance of the circuit controlled in a stepwise
fashion, means produce oscillation for digital quantization.
The thesis work investigated the consequences of the temperature behavior of PIN
diodes by designing several PIN diodes based circuits [21] - [24] and proposed a novel
compensation technique to achieve temperature insensitive RF performance. It is shown
that the proposed compensation scheme, based on the novel optimum bias load line
technology, can provide a setting accuracy of 0.2 dB over a very wide temperature range
o f-20 to +70 °C for all practical PIN diodes. The proposed scheme achieved temperature
insensitive RF performance of PIN diode based circuits without using any temperature
sensor or separate compensating mechanism, but responds directly to the junction
temperature of the diodes. This prevents any errors caused by temperature gradients, or
by self-heating of the diodes due to operation at high RF power level.
Schottky barrier diodes are widely used for microwave applications [26], [27]. This
device is used as RF detector, attenuator, mixer, limiter, etc. Among the various types of
linearizers [28] - [41], Schottky diode based linearizers are gaining popularity due to
their low power consumption, compact size, and ease of operation. Schottky diode is
used to generate non-linear characteristic for linearizer applications [28] - [39] and also
(PHD Thesis
Xj,g 9{p.. 5024 I9'f‘ January, 2004.
S C S e r a , Qujarat V m vcrsity, India.
-4 -
are used as microwave phase shifter for microwave beam forming network. Cut-in
voltage and RF resistance of the Schottky barrier diode is known to vary with
temperature. Thus, attenuation, limiting power level, non-linear characteristic and phase
shift of Schottky diode based circuits will change with temperature.
Some Schottky diode based linearizers are series [28], [29] diode based circuit and
others are parallel diode based circuits [30], [31]. Series diode based linearizer circuits
are biased by nearly constant voltage source whereas parallel diode based linearizers are
biased by nearly constant current source to achieve proper amplitude and phase
characteristic of the linearizer to linearize nonlinear characteristics of SSPA and TWTA.
Under constant voltage or constant current bias condition, Schottky diode based
linearizer’s performance changes with temperature since RF resistance of the diode is
function of temperature. The circuits where diodes are biases with constant voltage bias
condition will be suffered severely with change of temperature whereas constant current
biased diode circuits will be less affected.
Although constant voltage or constant current biased Schottky diode based circuit’s
performance affected over the change of temperature, surprisingly there is no open report
addressing this effect. However, same temperature compensation mechanisms applicable
for PIN diode based circuits are also can be applicable to compensate Schottky diode
based circuits with their merits and demerits. The thesis work proposed and successfully
demonstrate that the “optimum bias load line technology” applicable for temperature
compensation of PIN diode based circuits are also applicable for temperature invariant
performance of Schottky barrier diode based circuits with their merits of simplicity,
compactness and ease of operation and no error due to self heating of the diode [21],
[41].
The thesis work extended the proposed temperature compensation scheme “optimum
bias load line technology” used for PIN diode and Schottky barrier diode based circuits,
used for temperature compensation of output light intensity (brightness / illumination)
variation of Light Emitting Diode (LED). Today, LEDs are used widely for domestic
lighting and for signal lighting in automobile applications. LEDs are also used as
reference source for colour printer and measuring instruments [42], [43]. Light intensity
of LED decrease exponentially with the increase of diode temperature [43]. Thus for
solid state lighting application, LEDs light becomes dimmest during hot summer day and
!PA© Thesis
$j.g. Oio.: 5024. IS'* January, 2004.
S C S e r a , Qujarat U niversity, India
-5 -
brightest during cold winter day which are contrary with the requirement. In colour
printer and instrument applications, where it is used as reference light source, intensity
variation creates change of colour and measurement errors respectively. There are
various temperature compensation mechanisms, which are based on ambient temperature
sensor [43] - [46], with their demerits of circuit complication and compensation error
due to self-heating of the diode. The thesis work [47], [48] proposed and successfully
demonstrated theoretically and practically that “optimum bias load line technology” can
provide temperature invariant light intensity of LEDs with its merit of simple circuit and
no error due to self-heating of the diode.
Microwave transistors such as MEtal Semiconductor Field Effect Transistors
(MESFETs), and pseudo-morphic High Electron Mobility Transistors (pHEMTs) play an
important role in microwave applications [49]-[52], These devices are becoming more
prevalent as the demand grows for high-speed data transfer and internet access. As a
result, users are demanding smarter circuits with temperature insensitive performance for
the present systems.
Several researcher conducted extensive research about temperature behavior on
MESFET and HEMT devices [53] - [56]. DC performance [53] as well as RF
performance [54] - [56] of MESFET and HEMTs is changes with temperature. Linear
gain, saturated output power level, and efficiency of GaAs-based FETs depend strongly
on temperature [52]. At fixed drain current and voltage bias, transconductance (gm) as
well as FET unity current gain cut-off frequency (ft) decreases with increasing
temperature. Therefore, in case of fixed drain current bias, small signal gain, saturated
output power level and efficiency of the amplifier decreases with increasing temperature.
It has been known from the equivalent circuit model and verified by the experiments on
various amplifier circuits [57] that the linear gain of GaAs-based FETs and HEMTs
decreases approximately by 0.015 dB/°C for each stage as the channel temperature
increases. Thus, FET gain tends to decrease greatly with temperature, even for FETs
biased at a constant current bias. Such a decrease will significantly degrade the system
performance.
Decrease of small signal gain and output power level with the increase of device
temperature is the main concern for microwave amplifiers. There are various methods to
compensate small signal gain variation of MESFET and HEMTs [58] - [61]. Some
'J’h'D Thesis
'Kf'fi-
50Z4
januanj, 2004
s C ‘B rra , (jujarat U niversity, India
-
6
-
temperature compensation technology includes on-chip compensation mechanisms [58],
[59] and other technology uses separate variable gain blocks in the amplifier line up to
compensate overall performance. The thesis work demonstrates [61], [83] temperature
compensation mechanism using variable PIN diode attenuator to compensate overall
gain variation of the amplifier.
There are various microwave subsystems based on PIN diode, Schottky barrier diode,
MESFET and HEMTs [62] - [83] used for satellite and other applications. In the thesis
work temperature behavior and compensation of various spacecraft subsystems, such as,
PIN diode based beam forming network [70], [71] diode based linearizer [39], [40] over
drive level control system for SSPAs [80] and Ku-band channel amplifier with automatic
level control system [83] are discussed. These subsystems are designed using GaAs FET
and HEMT devices, PIN diodes and Schottky barrier diodes. Temperature compensation
schemes of the subsystems are proposed, mathematical and practical procedures are
given to determine the component values. These subsystems are characterized over
operating temperature range and test results are presented
Vh® Thesis
V jg 9(p . 5 6 2 4 .19li January, 2004
S C Vera , gujarat U niversity, India.
-
7
-
1.1
STATEMENT OF PROBLEM
One of the intent of the research is proper understanding of temperature behavior of
microwave semiconductor devices such as PIN diode, Schottky barrier diodes and GaAs
based MESFET and HEMTs. Another intent of the research is to explored the
temperature compensation mechanisms for various microwave circuits and systems,
based on semiconductor active devices, suitable for high reliable air borne and space
based application
The temperature behavior and compensation mechanisms, suitable for airborne and
space-based microwave active circuits and systems, addressed in the present research
work are categorized as below:
•
Circuits based on RF resistance of microwave diodes such as PIN diode and
Schottky barrier diode.
•
Circuits based on pHEMT and MESFET devices.
•
Complete subsystems based on various microwave diodes and transistors.
Pn(D *ffiests
tRjg J’lp-- S 0 2 4 . J a n u a r y , 2004
S C "Sera, gujarat U niversity, Indus
-
8
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1.2
ORGANIZATION OF THESIS_______________
The thesis is organized in five chapters addressing temperature behavior and
compensation mechanisms of different microwave diodes and transistors. Popularly used
microwave diodes used for satellite microwave communications are PEN diode and
Schottky barrier diode. A novel technique is invented for temperature compensation of
these microwave diodes. It is also presented that the same technique is also applicable to
the temperature compensate of output light intensity of light emitting diode. Among the
microwave transistors temperature behavior of GaAs MESFET and pHEMT are
addressed. At last design development, temperature characterization and compensation
techniques of various microwave circuits and subsystems based on solid-state active
devices are also presented.
Chapter-I consists of introduction containing literature survey and statement of the
problems addressed in the thesis.
Chapter-II addresses temperature behavior and compensation technique of various
junction diodes. Different temperature compensation techniques of RF resistance of PEN
diode and Schottky barrier diodes are discussed and a novel technique is proposed to
achieve temperature insensitive RF resistance, which leads to temperature invariant
attenuation of diode based RF circuits. Here it is investigated the consequences of the
temperature behavior of PIN diodes by designing several PIN diode based circuits and
proposed a simple technique to achieve temperature insensitive RF performance. It is
shown that the proposed temperature compensation technique, based on the novel
optimum bias load line technology, can provide an attenuation setting accuracy of 0.2 dB
over a very wide temperature range of -20 to +70 °C for all practical PIN diodes.
Different circuits operating at frequencies S, C and Ku-bands are realized with the
optimum bias load line scheme using different types of PIN diodes. It is verified that the
proposed load line biasing scheme provide nearly temperature invariant RF resistance for
all practical available PIN diodes at all the operating frequency range. This temperature
insensitive RF resistance is achieved without using any temperature sensor or separate
compensating mechanism, but responds directly to the junction temperature of the
diodes. This prevents any errors caused by temperature gradients, or by self-heating of
the diodes due to operation at high RF power level.
Tfl(D ‘Tkests
9(o.. 5024. IS 11* ')anuary, 2004
s C ‘B tr a , Qujarat ‘University, India
-9 -
The thesis work offers a fresh look into the temperature behavior and compensation
mechanism of Schottky barrier diode at radio frequency operation by designing various
Schottky barrier diode based circuits. Here it is shown that, temperature invariant RF
resistance of Schottky diode can be achieved by proper selection of the bias load line.
Thus, without any separate temperature sensor and compensation circuits, as used in
conventional temperature-compensation scheme, it is possible to achieve temperature
compensated RF resistance of Schottky diode. Mathematically it is shown, and verified
by the measurements that, variation of RF resistance, which is nearly 24 % in case of
fixed current bias over the operating temperature range of -10 to +60 °C, can be
minimized to nearly zero by the proposed optimum load line biasing scheme.
Temperature characterization and compensation of Schottky barrier diodes performance
over varying RF power level is also presented.
The temperature compensation scheme used to compensate PIN and Schottky barrier
diode
is
extended
for
temperature
compensation
of
output
light
intensity
(brightness/illumination) variation of Light Emitting Diode (LED). Light intensity of
LED decreases with increase of diode temperature at fixed current bias condition. Today,
LEDs are used widely for signal lighting in automobile applications. During hot summer
day at noon, LEDs light becomes dimmest contrary with the requirement of brightest
intensity, and during cold winter day at night, LEDs light becomes brightest contrary to
the requirement of dimmest intensity. In color printer application, light intensity
fluctuation results in a change of color. In the measuring instrument application, where it
is used as a reference intensity light source, intensity variation creates measurement
errors. In the research work, temperature behavior of light-emitting diode’s parameters
investigated mathematically and exploited the temperature variation of diode current to
improve the temperature dependency of LED’s light intensity. It is shown, that the
equiintensity curves of LEDs can be considered as nearly a straight line over the
operating range of diode temperature. Therefore, by selecting the best-fitted load line
along the equiintensity curve leads to nearly temperature invariant light intensity. Test
results confirms the same and verified by the measurements that, peak to peak variation
of light intensity , which is nearly 99% in case of conventional constant current bias,
over the temperature range of -20 to +80 °C, can be minimized to nearly 6% by selecting
proposed optimum load line biasing scheme.
i ’h'l) Thesis
%py <Xp • 5 0 2 4 .19 ‘* January, 2004
S C (Bera, Qujarat ‘University, India
-
10
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Chapter-in addresses temperature behavior of GaAs based MESFET and HEMTs.
Through different experiments, it is shown that linear gain of GaAs-based FETs and
HEMTs decreases approximately by 0.015 dB/°C for each stage as channel temperature
increases. Also discusses different techniques for temperature compensation of gain and
output power variation.
Chapter-IV addresses temperature behavior of some microwave subsystems. In this
work, temperature behavior and compensation of various spacecraft subsystems, such as,
PIN diode based beam forming network, diode based linearizer, over drive level control
system for SSPAs and Ku-band channel amplifier with automatic level control system
are discussed. These subsystems are designed using GaAs FET and HEMT devices, PIN
diodes and Schottky barrier diodes. Temperature compensation schemes of the
subsystems are proposed, mathematical and practical procedures are given to determine
the component values. These subsystems are characterized over operating temperature
range and test results are presented.
Chapter-V contains summary conclusion and future scope.
Tft'D m usts
X p • 5024.19“ January, 2004
S C (Bern, Qujarat University, India
-
11
-
C hapter - I I
Ph'D ‘Thesis
Ups ‘Hp . 5024 IS“ January, 2004
S C Hera, Qujarat University, India.
12
-
2.0
JUNCTION DIODES
Junction diodes popularly used for microwave circuits and systems are PIN diode
and Schottky barrier diode [1] - [5]. PIN diodes are used for controlling, attenuating,
limiting microwave signal. Schottky barrier diodes are used for various microwave
applications such as RF detector, attenuator, mixer, limiter etc. This device is also used
to generate non-linear characteristic for linearizer applications [28] - [35]. Both the
diodes are also used for microwave phase shifter for beam forming network [68] - [71].
Performance of PIN diode and Schottky barrier diode based circuits is mainly
determined by RF resistance of the diodes, which is determined by DC bias current.
DC I-V characteristic and RF resistance of PIN diode and Schottky barrier diode is
known to vary with temperature. Thus, performance parameters such as attenuation,
limiting power level, non-linear characteristic, and phase shift of PIN diode and Schottky
diode based circuits will change with temperature.
This chapter presents different past and present temperature compensation techniques
of junction diode and diode based circuits with their merits and demerits. This chapter
also presents the proposed novel “optimum bias load line technology” for the diode
based microwave circuits to provide temperature invariant RF performance. Different
diode based circuits are realized operating at different frequency bands and demonstrate
the experimental results under optimum bias load line condition. Here it is also shown
that same optimum bias load line technique is also applicable to temperature compensate
the output light intensity of Light Emitting Diode.
Pfi'-D ‘TUtsis
%fg 9{p.. 5024 I 3 ‘k January, 2004
S C (Bern, Qujarat ‘U niversity, India.
-
13
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2.1
PIN DIODE
PIN diode is a useful control element at microwave frequencies. Although PIN
diodes have been used for several decades [1] - [7], design of bias networks for
electronically controlled, PIN diode based attenuators, phase shifters, etc. has remained
something of a black art. This is particularly true when accurate control is required over
a wide temperature range. The reason is that the effect of temperature on the RF
resistance of PIN diodes was not well understood in the past. In the early 1980s, Alfa
Industry’s Application Note No. 80200 [2] had this to say: “We do not label the
[Temperature versus RF resistance] curves because we do not yet have enough data and
do not want to mislead you. We want you to realize that you will have to do your own
experimenting with your diode in your application.” Indeed, it was not until 1993 that
Caverly and Hiller [12] - [14] showed that the RF resistance of these diodes varies as the
(2-m)-th power of the temperature, where the power m lies between 0 and 2 for practical
PIN diodes. Caverly and Hiller have suggested using specially fabricated PIN diode to
achieve the temperature insensitive PIN diode attenuation [13], [14]. However, this
technique is not applicable to large varieties of commercially available diodes.
Here it is investigated the mathematical consequences of the temperature behavior of
PIN diodes and by designing, testing several PIN diode based circuits [21]-[24],
proposed a simple bias network to achieve temperature insensitive RF performance. It is
shown that the proposed bias network, based on the novel optimum bias load line
technology, can provide attenuation setting accuracy of 0.2 dB over a very wide
temperature range of -2 0 to +70 °C for all practical PIN diodes. It is verified that the
proposed load line biasing scheme provide nearly temperature invariant RF resistance for
all practical available PIN diodes at all the operating frequency range. This temperature
insensitive RF resistance is achieved without using any temperature sensor or separate
compensating mechanism, but responds directly to the junction temperature of the
diodes. This prevents any errors caused by temperature gradients, or by self-heating of
the diodes due to operation at high RF power level.
<PA'D ‘Thesis
%.eg 9{p : 5024 19‘‘ January, 2004
S C ‘9 era, CJajar at U niversity, India
-
14
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2. 1. 1.
P R IN C IP L E O F O P E R A T IO N
A PIN diode is a useful control element at microwave frequencies [1] - [7]. These
diodes differ considerably from ordinary p-n junction diodes. A PIN diode has a heavily
doped p-region and a heavily doped n region separated by a layer of high resistivity
nearly intrinsic layer. The thickness of the intrinsic layer is usually in the range of 10 to
200 pm. The intrinsic layer has a resistivity of about 1000 ohm-cm and is either lightly
doped p-type or n-type. This lightly doped layer is called 7C-type or v-type depending
Space charge density
©
Impurity Concentration
o
upon whether the conductivity is p-type or n-type, respectively.
p-i-n structure
p-Jt-n structure
Fig.-2.L1: Impurity, space charge and electric field profile of PIN diode.
At zero bias, two space-charge regions are formed in p and n layers adjacent to the
intrinsic layer because of the diffusion of holes and electrons across the junctions. The
thickness of these regions is inversely proportional to the impurity concentration. In the
case of an ideal diode, the intrinsic layer has no impurities, i.e. it is totally depleted of
iP&$) *T&tsts
<
Ka$- ‘Hp - 5024 19l* January, 2094
S C 'Bern ? Q ujarat ‘U nwcTsitu^ l sidm
-15-
mobile charge carriers. Thus p-layer have a region of fixed negative charge and n-layer
have a region of fixed positive charge with equal charges in the two regions. When a
reverse bias is applied, the space-charge regions in the p and n layers become wider. A
uniform electric field exists in the intrinsic region, dropping linearly to zero through the
depletion regions in the n and p layers. This situation is shown in Figure-2.1.1 [5].
When the high-resistivity layer is actually a tt-layer, the situation is somewhat
different and is shown in Figure-2.1.1 [5] by dotted lines. With no applied bias, the
diffusion of holes and electrons across the n-tt-junction produces a very thin depletion
region in the n-layer and a thicker depletion region in the Tt-layer. The two depletion
regions contain equal but opposite fixed charges. In this case, the tt-layer is not
completely depleted of mobile carriers. As the reverse bias is applied, the depletion
region in 7t-layer becomes more and more thick till the entire tt-layer is swept free of
mobile carriers. The small negative bias required for this purpose is called the bias
required to swept out the tt-region. With increasing reverse bias, a thin depletion region
appears in p-Iayer also. The electric field rises steeply in the n-region, decays slowly
across the tt-region and then drops steeply to zero in the p-depletion region.
When a forward bias is applied to the diode, carrier injection into the tt-layer takes
place. Electrons are injected into the tt-layer from the n-layer, and holes are injected from
the p-layer. The carriers diffuse into the tt-layer, their concentration diminishing with
depth into the tt-layer because of recombination. The diffusion of carriers causes the
carrier concentrations in the tt-layer to increase above their equilibrium levels, and the
resistivity of this layer drops as the forward bias is increased. The hole and electron
concentrations are about equal throughout the tt-layer. If the carrier lifetimes in the ttlayer are relatively long and the tt-layer is not too thick, the tt-layer becomes flooded
with carriers at a reasonable forward bias level. When this happens, the diode exhibits a
low resistance and, in fact, appears as a virtual short circuit across a microwave
transmission line. For this characteristic of the PIN diode, it is essential that the lifetime
of carriers in the tt-layer be greater than the time period at the operating frequency. For
this reason, PIN diodes cannot be operated usually as a bias controlled linear resistor at
lower radio frequencies.
The effect of controlling the carrier density, and hence the resistance in the /-layer,
by varying the forward bias is known as conductivity modulation. This phenomenon
2>fi© rh ts is
%t3 Oio : 5 0 2 4 .1 9 * January, 2004.
S C 'Beta, g ajar at U niversity, India
-16-
plays the key role in the use of PIN diodes as electronically controlled attenuators and in
amplitude modulation. It may be recalled that the ordinary p-n junction diodes have a
relatively thin depletion layers and do not exhibit the phenomenon of conductivity
modulation as seen in PIN diodes.
The relationship for the microwave (actually for frequencies much greater than the
inverse of the I-region carrier lifetime) resistance of a PIN diode contains terms that are
related to the forward current, device geometry, and electronic properties of its
semiconductor material. These parameters are related to the PIN diode resistance by the
following simple expression (2.1.1) [1]:
W2
R ,=
( 2. 1. 1)
2W j l
where, W is the I-region thickness, / 4 is the I-region ambipolar mobility, ra is the
ambipolar carrier life time, / is the dc forward current of the diode and the value of
current exponent p is nearly equal to 1.
The generalized equivalent circuit of the PIN diode is shown in Figure-2.1.2 [5]. The
Fig.-2.L2: Equivalent circuit of PIN diode
characteristic features of a PIN structure are shown enclosed in a dotted loop. The
characteristic of the PIN structure contains: (a) Ri and C/ representing the resistance and
the capacitance of the portion of the high resistivity layer exclusive of the swept out
region, and (b) diffusion capacitance Co representing the charge storage caused by the
2Vt© ‘Thesis
3(t:g. 9{p.. 5024 19,e January, 2004
5 C Vera , Qujarat 41 nwersity, India
-
17
-
current flow through the intrinsic region. Other components in the equivalent circuit
correspond to the model of a P-N junction diode. Rj and Cj are junction resistance and
junction capacitance, respectively. Rs is the Ohmic series resistance of the diode and CF
is the fringing capacitance associated with the device chip. Series inductance of the
bonding wire is represented by Ls and the capacitance CP accounts for the package
capacitance.
Under forward bias condition, the diode’s intrinsic region series resistance R/
changes inversely with the diodes direct current.
2 .1 .2 .
T E M P E R A T U R E B E H A V IO R
PIN diode attenuators are commonly used as microwave control devices. The
attenuation level is mainly dependent on the value of PIN diode resistance, which is
primarily established by the forward bias current at room temperature. The performance
of PIN diode based RF circuits varies with the variation of diode temperature is due to
the temperature dependence of RF resistance of PIN diode. Thus the attenuation level
and other RF parameters that is determined by the RF resistance of the PIN diode is
affected by the change of environment temperature as well as self-heating of the PIN
diode when PIN diode based circuits are operated in a wide temperature variation
environment..
I-region ambipolar mobility /4, and the ambipolar carrier life time Ta, are both
functions of temperature, therefore, high frequency resistance of the PIN diode, given by
the equation (2.1.1), also depends upon temperature.
The temperature dependence of the mobility in silicon has been extensively studied
[25] and it has been established that mobility decreases with increasing temperature in
the temperature range of interest. This temperature dependence can be approximated by:
f j,
M(T) = ju(T0)
T
\ x <>
V”
(2 .1 .2 )
y
in the temperature range of 223 to 473 Kelvin (-50 to 220 °C) and with a value of n In
the range of approximately 2 to 2.2.
V h V Thesis
<Hfg V{p . 5024 19,k January, 2004
5C Vera ,
-
18-
(jujarat ‘University, India.
Based on the measurements and works on silicon devices, earner lifetime has been
found to increase with increasing temperature. The temperature dependence of carrier
lifetime may be modeled by the following expression [12,13]:
t (T)
= t (T0)\
(2.1.3)
combining equation (2.1.1), (2.1.2) and (2.1.3), the temperature dependence of a PIN
diode resistance can be written as follows:
i?i(r)=z.-
(2.1.4)
where, z is a constant depends upon the geometry of the diode and independent of
temperature.
F igure-2.13 shows the normalized resistance with respect to the value at
temperature 25 °C over the temperature range of -50 to +80 °C using the carrier lifetime
1 A
—
4,
Kj
bo
o
O
Normalized Resistance
Normalized to 25 °C
-70
-20
30
80
Temperature (°C)
Fig.-2.1.3: Plot o f normalized PIN diode resistance versus temperature using
earner lifetime coefficient m as a parameter.
coefficient m as a parameter. The figure illustrates that, depending on the temperature
dependence of the carrier lifetime, the resistance of the PIN diode may increase, decrease
or remain constant over a wide variation in temperature. The value o f m equal n (=2)
would indicate no change in resistance with temperature.
Tfi’D Thesis
mpg
5024 15‘‘ January, 2004.
S C Vera , Qujarat University, India
-
19 -
2.1.3.
TEMPERATURE COMPENSATION
When the PIN diodes are operated under conventional fixed current bias, the RF
resistance of the diode increases with the increase of temperature as suggested by the
equation (2.1.4) since practically (n-m) * 0, and n>m. There are different approaches to
achieve temperature invariant RF resistance of the PIN diode:
1.
The temperature dependent control voltage generated using separate temperature
sensor. The temperature sensor sense the ambient temperature of the diode,
accordingly the control circuit controls the forward current of the diode in such a
manner that the factor
of equation (2.1.4) remains constant with the
change of temperature [15] - [20].
2.
The PIN diodes are specially fabricating so that the temperature exponent (n-m) of
equation (2.1.4) becomes zero as proposed by Caverly and Hiller [12] - [14].
3.
Optimum bias load line technique, as proposed in the thesis work [21]-[24].
The following sections will discuss these different types of temperature
compensation mechanisms with their merits and demerits to achieve temperature
invariant RF resistance of PIN diode.
2 .1 3 .1 .
T e m p e r a tu re C o m p e n s a tio n U s in g S e p a r a te
T e m p e r a tu re S e n s o r
Typically, PIN diode attenuators have been controlled by bias current and operated by
biasing through constant current source. This is caused by the fact that PIN diode is
current controlled device and the attenuation level is dependent on the value of the PIN
diode’s RF resistance, which is primarily established by the forward bias current. There
are numerous conventional compensation mechanisms [15] -
[20], which uses
temperature sensor to sense the ambient temperature of the diode and produce a signal
that is function of temperature. This signal modified the current applied to the diode in
such a manner that the diode provides same RF resistance at all operating temperature of
the diode. Some of the compensation mechanism based entirely on analog control signal
Thm Thesis
Kj-g. sy>.. 5024 19‘‘ January, 2004
S C Vera, (jujarat ‘University, Indus.
-
20
-
[15] - [18], while others contain digital control signal acting at discrete temperature steps
[19], [20],
2 . 1 .3.2
A n a lo g C o m p e n s a tio n S c h e m e
There are numerous analog compensation mechanisms, which uses temperature
sensor to sense the ambient temperature of the diode and produce a continuous varying
signal that is function of temperature. This signal modified the current continuously
applied to the diode in such a manner that it maintain the same RF resistance of the diode
at all temperature [15] - [18].
R3
Fig.- 2.1.4: Schematic block diagram of an analog
temperature compensation circuit.
As an example, Figure-2.1.4 shows the schematic diagram of such an analog
temperature compensation circuit. The circuit contains a temperature controlled voltage
generator containing thermistor associated with some resistive networks. The operation
principle of the temperature-compensation circuit is as follows:
1. When the ambient temperature rises, the resistance of the thermistor becomes
changes.
2. The offset voltage Von** changes and trace the temperature variation of the
PIN diode forward voltage drop. Thus, Voffs* can effectively cancel the
forward voltage drop of the diode.
ThTf Thesis
%eg. 7{p.: 5024. 19,k January, 2004.
S C (Bern , Qujarat U niversity, India.
3. The input control voltage Vconuoi is injected to the inverting input of the
differential amplifier to adjust the magnitude of the temperature control
.voltage.
4. Offset voltage is also injected to the non-inverting input of the differential
amplifier (OPAMP). Therefore, the differential amplifier output voltage Vf
has forward voltage variation over operating temperature range to maintain
RF resistance of the PIN diode constant over the change of temperature.
With the analog compensation mechanism, the reported [15] attenuation accuracy is
0.5 dB over the temperature range of -15 to +65 °C, for fixed attenuation of -12 dB.
However, we believe that the successful adjustment of such control circuits is a time
consuming task, often involving much trial and error; and the circuits themselves are
relatively complex.
The above circuit is for a single value of attenuation. However, there are various
applications where attenuation in discrete steps is required e.g. to control the onboard
gain control of the communication transponder by giving command from ground. The
typical commandable gain-control requirement is up to 22 dB in steps of 2 dB. The
temperature compensation scheme as discussed in this section is for only one value of
attenuation, the circuit will be more complex and successful adjustment of the
component values will be difficult, when attenuation is required in number of steps.
2 .1 .3 .3
D igital Compensation Schem e
There are also numerous digital compensation mechanisms [19], [20], which uses
temperature sensor to sense the ambient temperature of the diode and contain digital
devices acting at discrete temperature steps. Temperature dependent discrete control
signal modified the current discretely applied to the diode in such a manner that the PIN
diode maintain the same RF resistance at all discrete temperatures.
T h V Thesis
S sg . 9{p.. 5024 19‘li January, 2004
S C 'Sera, Qujarat University, India.
-
22-
As an example, Figure-2.1.5 shows the schematic block diagram of such a digital
temperature compensation circuit. The circuit consists of a thermistor to sense the
temperature of the diode, and a programmable memory device (PROM) associated with
analog to digital converter (ADC), digital to analog converter (DAC) and some resistors.
The operation principle of the digital temperature-compensation circuit is as follows:
+v
R2
$
Ri
DAC
-AMV
ADC
Rth
M
ATTN.
Fig.-2.1.5: Schematic of a digital temperature compensation circuit
1. When ambient temperature increases, resistance of the thermistor changes
depending upon the temperature coefficient of the thermistor.
2. The output of the ADC changes which determine the address of the PROM.
3. The pre-stored data in the memory will be the input of the DAC.
4. DAC output will provide the bias current to the PIN diode through the resistor
r 2.
Depending upon the temperature of the diode, which is sensed by the thermistor, the
temperature dependent pre-determined discrete stored data will provide the proper bias
current to the diode to achieve temperature invariant RF resistance of the diode.
However, the proper determination of the memory data of such control circuits is a
time consuming task, often involving much trial and error. The schematic digital
compensation circuit as shown in Figure-2.1.5 is only for a fixed attenuation value.
However, the circuits themselves become relatively complex, and time required to
acquire data will be more when attenuation is required in number of steps.
‘Ph'D 'Thesis
X$g. Oip. $024 IS * January, 2004
S C B e r a , Qujarat 11nwersity, India
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23-
2 .1.3.4
S p e c ia lly F a b r ic a te d D io d e
As shown in the Figure-2.1.3, the resistances described by equation (2.1.4), are
plotted with respect to temperature using simulated values of carrier lifetime coefficient
m as a parameter. The figure illustrates that, depending on the temperature dependence of
the carrier lifetime, the resistance of the PIN diode may increase, decrease or remain
constant over a wide temperature range. A value of m equal to 2 would result in no
change in resistance with temperature.
A variety of factors influences the carrier-lifetime temperature characteristic. Some
of the important factors are [12] - [ 14]:
•
The diode geometry, specially the I-region width and diameter.
•
The type of material used to passivate the diode surface.
•
Whether impurities have been intentionally introduced into the intrinsic layer
in order to reduce carrier lifetime.
O f these three factors, the first two factors are typical of PIN diodes most commonly
used in microwave attenuator design.
Caverly and Hiller showed [12]-[14] that lifetime temperature coefficient of the PIN
diode depends upon both the passivation material and geometry i.e., I-region area to I
region width of the diode.
2.5
on J
s 1.0-
0.1
C(pF)
1.0
10.0
Fig.-2.1.6; Temperature coefficient of carrier lifetime (m) verses junction
capacitance of diode, passivation material as the parameter.
S>A© 1 /tests
9{p : 5024 19li January, 2004
-
24-
Figure-2.1.6 shows the dependence of carrier lifetime on junction capacitance of the
diode and passivation material as the parameter. Where, junction capacitance at punch
through is the measure of the diode geometry, which is directly proportional to ratio of Iregion area to I-region width. This figure shows, that for large capacitance, nitride
passivated diodes, carrier lifetime increases approximately linearly with temperature i.e.
m - 1. For large capacitance diode with glass passivation and silicon dioxide passivation,
the temperature coefficient of lifetime m = 2. Thus, for large area PIN diode with glass or
silicon dioxide passivated diode has RF resistance with zero temperature co-efficient.
For small area diode i.e. diode with lower capacitance (0.1 pF) and with silicon dioxide
passivation, the m value is nearly 1.7, means temperature coefficient of the RF resistance
of the PIN diode is very small.
Thus, it can be concluded that with proper choice of diode passivation material and
geometry, it is possible to achieve low temperature coefficient /temperature invariant RF
resistance of the diode with the compromise of capacitance value of the diode. However,
high value diode capacitance increases the circuit insertion loss and limits the available
range of attenuation. Capacitance effect will be more serious as the frequency of
operation increases.
2 .1 .3 .5
P ro p o s e d O p tim u m L o a d - L in e B ia s in g T e c h n iq u e
Caverly and Hiller have suggested [12]-[14] that a specially fabricated diode with
proper passivation material and area to width ratio, it is possible to having RF resistance
of PIN diode of temperature index (2-m) is nearly equal to zero. Thus, use of this
specially fabricated diode for attenuator or other circuits; it is possible to achieve a
nearly temperature invariant RF performance when the diodes are biased at a constant
forward current condition. It seems unlikely, however, that this requirement can be met
without sacrificing other aspects of the PIN diode’s performance. In any case, we find
that typical commercially available diodes have m close to unity, which means that
constant-current biasing will not provide acceptable temperature stability. Therefore
proper temperature compensation mechanism is required to achieve acceptable
temperature stability of the circuits where performance depends upon the RF resistance
of the PIN diode.
Ph<D 'Thesis
%eg. 9(p.: S024. 19,k January, 2004.
S C 'Beta , Qujarat University, India.
-
25-
There
are
numerous
conventional
compensation mechanisms,
which
uses
temperature sensor to sense the temperature of the diode and produce a signal that is
function of temperature. This signal modified the current applied to the diode in such a
manner that it maintains the same RF resistance of the diode at all temperature. Some of
the compensation mechanism based entirely on analog control signal, while others
contain digital control signal acting at discrete temperature steps as discussed in the
previous sections. However, the successful adjustment of such control circuits is a time
consuming task, often involving much trial and error; and the circuits themselves are
relatively complex, specially, when attenuation is required in number of steps.
The thesis work offers a fresh insight into the behavior of PIN diodes over
temperature [21] - [24]. A simple but accurate biasing and control circuit for attenuators
and other PIN diode applications which depends upon the RF resistance of the PIN diode
is described. The circuit can easily be optimized for a high attenuation setting accuracy
over a wide temperature range. It is investigated some mathematical consequences of the
PIN diodes equation and shows that a very simple bias network, designed on this
principle, can provide a setting accuracy of 0.2 dB over a temperature range of -2 0 to
+70 °C for all practical values of m. This temperature stability is achieved without using
any temperature sensor or compensating circuit. Practical implementation and test results
for such circuits are also presented.
2 .1 .3 .5 .1
T e m p e r a tu re D e p e n d e n t P IN D io d e M o d e l
The forward biased PEN diode can be modeled as a bias and temperature dependent
RF resistance given by equation (2.1.5) as discussed to derive the equation (2.1.4) [12],
[13].
rp 2 —m
s ' (,' ’r ) = z ' K O T
" C2I-5)
Whereas, forward biased junction resistance Rj is determined by the current voltage
relationship of the PIN diode is given by the equation (2.1.6) [1].
I d(Vd, T ) = A J \ E x p
Th<D ‘Thesis
ijkT
%pg 9{p ■S0Z4 19* Sanitary, 2004
••( 2 . 1.6 )
S C $ e r a , Qujarat ‘University, India
-
26-
Eb is the band gap potential of the semiconductor in volts, 7] is the ideality factor of
the diode, q is the electron charge, k is the Boltzman’s constant, n is temperature
exponent, Rs is the equivalent series resistance of the diode and A is a constant depends
on the area of the diode but independent on diode temperature T.
Under forward bias condition, diode capacitance can be considered as independent
on bias voltage as well as temperature. Other parasites e.g. lead inductance, package
capacitance and fringing capacitance of the diode can be considered as independent on
temperature.
2 .1 .3 .5 .2
A c h ie v in g E q u ire s is ta n c e C u rve O v e r T e m p e r a tu re
Caverly and Hiller have suggested that a carefully fabricated PIN diode having a
temperature index 2-m = 0, can be used in an attenuator to achieve a zero temperature
coefficient when biased at a constant forward current. It seems unlikely, however, that
this requirement can be met without sacrificing other aspects, e.g. high frequency
operation due to high capacitance, of the PIN diode’s performance. In any case, we find
that typical commercially available diodes have m close to unity, which means that
constant-current biasing will not provide acceptable temperature stability.
We now discuss what requirements the bias network must meet in order to maintain a
constant RF resistance over temperature without imposing narrow constraints on the
characteristics o f the diodes used. If RIO is the desired RF resistance and Rj the actual
resistance at a given current and temperature, then the bias network must ensure the
relation:
R, —Rl0
•(2.1.7)
Combining this constraint (2.1.7) with equation (2.1.5) and (2.1.6), we can write [21]:
.......(2X 8)
V
1i.k.T
J
Where Vd(T) is given by equation (2.1.9).
~>2—m
Vd{T) —Eb +
R„
Tti$> “T h esis
% eg.
p n n .kT T
^
Rs + —-----.Ln
q.p
A pRI0
■5 0 2 4 .1 9 “ J a n u a r y , 2 0 0 4 .
(2-~n:~np)
5 C 'B e r n ,
-27
•(2.1.9)
Q u ja r a t U n iv e r s it y , I n d ia .
This parametric equation is the equation of the ideal bias-point locus that will satisfy
R,
=
R,0 across the temperature range. We can call this locus the “equiresistance curve"
for the required RF resistance RioIf the constant-resistance V-I locus happens to be highly linear within the
temperature range of interest, then the load line of a simple bias circuit can be adjusted to
coincide with this curve, and will then maintain the desired RF resistance over
temperature as shown in Figure (2.1.7). This is indeed the case for the practical ranges
of values of p and m. Figure (2.1.8) and (2.1.9) show these curves for various values of p
and m for an attenuation of 8 dB. Numerical calculations using a typical range of m
1.2
Eb= 1.12
i? = 2
Oo
P
n= 1
m= 1
P= 1
-20 °C \
P
Bias Load
line
-K
Equiresistance
curve
O
Id (mA)
70 °C X
T. ............
0
xr n r l . \
Vd (Volts)
X.
J*
1
0 .8
|7 /
Fb
—'
1
Fig-2.1.7: Equiresistance curve and load line
Fig.-2.1.8i Equiresistance curves for different p values
'Thesis
%f£. 1{p : 5024.19* January, 2004
S C'Sera, gujarat University, India
-28-
values have confirmed that a temperature stability of less than 0.2 dB is easily achievable
between -2 0 and +70 °C.
Once the optimum load line has been determined, it is a simple matter to design the
bias network for a given attenuation step. As shown in Figure-2.1.7, the circuit’s opencircuit output voltage (in the sense of Thevenin’s theorem) must be equal to the voltage
intercept of the load line, and its Thevenin output resistance must be the reciprocal of the
o
*
slope.
Id(mO)
co
o
CM
o
.9
Vd (V olt)
Fig.-2.1.9: Equiresistance curves for different m values
I-V Curve at +70 °C
I-V Curve at +25 °C
I-V Curve a t-20 °C
ld(m ty
EquiResistance
Curves for 2 to
16 dB
attenuation
m=1.0251
p=0.9545
Eb=1.2175
Voltage
Intercept of
Load Lines
1
0 .4
V d (V o lt)
0 .8
1.2
Fig.-2.1.10: Equiresistance curves and load lines for different attenuation settings
‘PH'D Thesis
•
5024 19"’ January, 2004
S C (Bera , Qujarat U niversity, India
29
When this exercise is repeated for a number of RF resistance (attenuation) values, we
observe the interesting fact that the voltage intercept of the equiresistance curves is
almost constant for different attenuation values, and the voltage intercepts o f the
corresponding optimal load lines are also within a few millivolts of each other. This is
shown in Figure (2.1.10) for attenuation values from 2 dB to 16 dB.
This is no mere coincidence, as we can see if we determine the load line along the
equiresistance curve.
Combining equation (2.1.8), (2.1.9), (2.1.5) and eliminating temperature parameter T,
another form of equation of the equiresistance curve can be written as:
V d ~ E b + I dR S +
{ _ \ 2~m
tfk R m idp 2-ra
.In
pn
\ Rio j
A .lf m
•(2 . 1. 10)
Differentiating the constant resistance locus (2.1.10), we can write the slope (M0) of
the curve at the bias point (V*,, Id0) corresponding to temperature T0 as:
_ L = i5iL = fi
M.
SI,
+!t El. 2 - m ■ q'l*
pn
q{vtdo
2 —m
2 ~m
-h o R s
■Eb)
(2.1.11)
T}kT0
Thus, the equation of tangent passing through the bias point (1Idm Ido) at temperature
T0 corresponds to RF resistance Rio is given by
Vd=VJo+ - ± - { l d- l J
M0
( 2. 1. 12)
Putting value of M0 from equation (2.1.11) to equation (2.1.12), we can write the
equation of tangent to the equiresistance curve at temperature T„ corresponding to diode
bias (Vdo, Ido) is given by:
vd(id)=vdo+
f
- £ - ( E b -V do+ I doRs) + & ( - ^ ~ - l ) - l dt,Rs
-m
q \2~m )
JV
I '
(2.1.12A)
■*do
Taking this tangent to be our optimum load line, its voltage intercept Vopt can be
obtained by putting Id= 0 in equation (2.1.12A):
Vorr = V*
(BfcD ‘T&esis
+ - £- ( E b -V * + / * i O + — f ^ - l l
2 -m
q \2~m
J
9{p . 5 0 2 4 .19th January, 2004
(2.1.13)
5 C $ e r o , Ojujarat U niversity, India.
30
Putting voltage drop across the junction of the diode
VdJ0
= V do-
Id<Ms,
the optimum
voltage can be expressed as:
Vqpt ~ Vd,o
-{Eb - V djo)+
2 -m
'
+
"
VkTo r pn
-m
q
1
(2.1.13 A)
The intercept is seen to be practically independent of the selected bias point (V do, Ido),
implying that the optimum no-load output voltage of the bias circuit is the same for all
attenuation values. W hen p = 2-m this “magic” optimum voltage becomes [21], [23]:
^on-= £ » + —
(»-!)
(2.1.14)
<1
It is very close to the bandgap potential Eb of the semiconductor, since the last term is
of the order of a few millivolts. In the general case when p ^ m, the voltage intercept is
given by (2.1.13A).
Many commercial integrated voltage references are based on the bandgap principle
and provide outputs close to the Eb of the semiconductor. These parts, with a provision
for minor trimming of the voltage, may be used conveniently as bias sources in this
application.
2 .1 .3 .5 .3
A c c u ra c y o f O p tim u m B ia s L o a d L in e T e c h n iq u e o v e r
T e m p e ra tu re
Now we will investigate the accuracy of the optimum bias load line technique over
the operating range of temperature. Temperature coefficient of RF resistance variation,
say Tc /°C, of the diode can be written from equation (2.1.4) as:
Fig.-2.1.11: A simple bias circuit for PIN diode
fPft(D ‘Tfiests
!%££*
5024 .19** January, 2004
$ C tBera, (jujarat Zlntpersity, India
31
From the bias circuit of Figure-2.1.11, we can write
V -V
h
=
~
(2 -1. 16)
^
Combining equation (2.1.15) and (2.1.16), we can write
T =
2 —rti
P dVj
V - V d dT
(2.1.17)
Differentiating equation (2.1.9) with respect to temperature T
dVd _
dT
Eb - V d
T
pn + m —2 jjk
p
q
J]kT dRd
qpRd dT
Putting this expression in equation (2.1.17), we can write
(2-m )(V -Vd) -p (E b - V d) - { Pn + m - 2 ) ^
__________________________ q_
T{{V-Vd) + ^ T / q ]
(2.1.19)
This is the expression of temperature coefficient of RF resistance variation of the
diodes. For m = p =1 and n = 2, these are the typical valued of PIN diodes, this
expression reduces to:
T _ 1 V - E b -T]kT/q
c T V —Vd + JjkT/q
(2.1.20)
When, diodes are biased by constant current source, V->°o, and Tc = 1/T / °C.
Now, when diodes are biased by proposed optimum bias load line condition then
putting V= Vo p t in equation (2.1.20) from equation (2.1.14), temperature coefficient will
be:
T _1
( l ~ T / T 0)nkT0/ q
c T ( E b - V d)+{l + T / T a)nkT0/ q
P M ) Thesis
•&£. 9{p 5024 IS'* January, 2004
(2.1.21)
S C “B ern, gujarat V ntvcrsity, India
32
This expression shows that at T - Tg, temperature coefficient of RF resistance
becomes zero. Now we will investigate the temperature coefficient at other temperatures
putting typical parameter values of the PIN diodes.
For silicon PIN diodes, putting Eb = 1.12V, tj = 2.0, T0 = 298 K (25 °C), temperature
coefficient will be increases from 0 to 0.015 times when temperature changes from 25 °C
to 25±45 °C with respect to the variation in fixed current biasing condition.
Thus, proposed optimum bias load-line technique provides excellent temperature
accuracy of PIN diodes RF resistance with very simple circuit suitable for spacecraft
applications.
2 .1 .3 .5 .4
Experim ent an d Test Results
Figure-2.1.12 shows the schematic circuit diagram of the PIN diode circuit that has
been used for experiment on various PIN diodes. Isolated port of a 3-dB hybrid coupler
is used as the output of the circuit, two diodes are connected at the direct port and
coupled port of the hybrid and other port is used as the input port. R is load resistor to
drive the circuit from a voltage source V. Depending upon RF resistance of the diodes,
determined by the current through the diode, some portion of RF power will be
dissipated within the diode and rest of the power will be reflected from the diode. The
Fig.-2.1J2: Schematic circuit diagram and simple RF equivalent circuit of the
forward biased beam-lead PIN diode
'Thesis
%pg 9{o ■5024.19'6 January, 2004
S C Vera, gujarat University, India
33
reflected power from both the diode will be combined at the output port of the circuit.
Thus, S21 of the circuit is given by:
S 2 \ = (Zd - Z 0)/(Zd + Z a)
(2.1.22)
where, Zj is the RF impedance of the diode, i.e. parallel combination of RF resistance Rd
and capacitance Q
with series combination of parasitic series resistance (RJand
inductance (Ls). The circuit is realized in 25mil alumina substrate (£r = 9.9). Lange
coupler is used for 3-dB coupler at the centre frequency of 3.7 GHz for experiment at C
band frequency and layout is made for beam lead diodes. The assembly drawing and
photograph of the realized circuit is shown in Figure-2.1.13 and 2.1.14 respectively.
This circuit is used to demonstrate the temperature variation of RF resistance for PIN
diode at C-band frequency.
Fig.-2.1.13: MIC layout & assembly drawing
of PIN diode attenuator
2 .1 .3 .5 .4 .1
Fig.-2.1.14: Photograph of the PIN
diode MIC attenuator circuit.
Extraction o f PIN Diode P aram eters
To extract the parameter values of the PIN diode the circuit is kept in a temperature
controlled thermal chamber and DC as well as RF parameters of the circuit is measured
at different stable temperatures of the chamber. Beam lead PIN diode MPND 4005 of
Metalics is used to demonstrate the extraction of the diode parameter values [23],
Figure-2.1.15 shows the tested and simulated I-V characteristic of the PIN diode at three
3>fi® Thesis
-Keg. 9{p.: 5024. 19“ January, 2004.
S C Vera , Qujarat University, India.
34
Fig.-2.1.15:1-V characteristic of PIN diode (MPND 4005)
different chamber temperatures. The fitting parameters of the diodes I-V relationship are
n = 2 , E b = 1.12 Volt and t]=2.1. Voltage and current of the diode is measured by an
automatic test setup by applying pulse voltage to the diode, to avoid self-heating effect of
the diodes.
Figure-2.1.16 shows the tested and simulated attenuation verses DC bias current of
the diode at three different temperatures at frequency of 3700 MHz [23]. The fitting
parameters of the tested diode MPND 4005 are m - 1.15, p = 0.88, Rs = 4.5 Cl Cj = 0.02
pF.
S
Oi
o
A tte n u a tio n (d B )
oS
+60
0 .2
Id ( m A )
0.6
0 .4
0.8
Fig.-2.1.16: Attenuation Vs bias current of the diode MPND-4005
(PAD Thesis
%tg. 'H.a.: 5024. 19'* January, 2004.
5 C 'Bern , Qujarat U niversity, India.
35
a 60 d e g C T e s t e d
----- 60 d e g C S im u la t e d
22
-22
-10
-10
o
1
o
ff
cn
§
o
3
®
tn
w
A ttenuation (d B )
o
IO
25
0 -I----------------------!----------------------J -------------------?
3 35
3.55
3.75
3.95
F re q u e n c y (G H z )
Fig,-2.1.17i Attenuation Vs frequency at different current and temp.
Figure-2.1.17 shows the tested and simulated attenuation performance over the
frequency band of 3400 to 4000 MHz at three different constant bias current (0.18, 0.38,
0.68 mA) settings at the three different operating temperatures (-10, +22, +60 °C) [23].
The plot shows how attenuation varies when PIN diode biases by constant bias current.
Setting of 20 dB attenuation at -10 °C, decreases to 15 dB attenuation, at temperature of
60 °C, when the bias current of each diode kept constant at 0.68 mA.
The plot of Figures 2.1.15, 2.1.16 and 2.1.17 shows good agreement between
simulated and tested results. That proves the validity and accuracy of the extracted
parameter values £*,, m, n and p.
Table-2.1.1: Test results of the attenuator for V = 1.19 Volts
Temp.(°C)
Freq (GHz)
AUiu
(dB)
l= >
+25
-10
+60
3.40 3.55 3.70 3.85 4.00 3.40 3.55 3.70 3.85 4.00 3.40 3.55 3.70 3.85 4.00
5 01 496 495
994 9.92 9.96
14.85 1483 14 87
2015 1996 19 86
497
10 (X)
1496
19 83
502
1014
15 14
19 92
5 03
9.98
1495
20 30
499 498 501 507
9.97 1001 1009 10.21
1493 1499 1510 15.28
2011 20 02 20 00 20.09
499 496 496
1000 999 10.03
1498 1497 15.03
20 31 2014 20 05
499
1011
15 15
20 04
506
10.24
15.35
20.15
Putting these parameter values to expression (2.13A), optimum no load voltage
becomes 1.19 Volts for the diode MPND4005. The tested results of the attenuator when
diodes are biases by a voltage source of 1.19 Volts are shown in Table-2.1.1 [23].
Different attenuation setting from 5 to 20 dB in steps of 5 dB are achieved by setting
different resistance values ,of resistor R. This table shows that variation of all the step
27itP 'Thesis
9{p.: 5024 1916January, 2004
S C ®era, gujarat University, India
36
attenuation are less than 0.3 dB over the temperature range of -10 to +60 °C and over the
whole frequency band of 3.4 to 4.0 GHz.
Thus, from the above discussions it can be concluded that, for conventional constant
current bias condition, attenuation of the PIN diode attenuator changes drastically with
temperature. However, when the PIN diode biases by the derived optimum voltage
source of value = 1.19 Volts (for MPND4005 diode), current through the diode
controlled in such a way that RF resistance of the diode remains invariant over the
change of temperature.
2 .1 .3 .5 .4 .2
Experim ental D eterm ination o f Optim um Voltage (V opt)
The optimum open circuit voltage (V opt) can also be determined experimentally by
graphical method. To determine V0PT, the circuit is kept in a thermal chamber, voltage
and current of the diode is measured at three different chamber temperature (say at -10,
+25, +60 °C) for three attenuation setting (say 20,15 and 10 dB) at frequency of 3.7 GHz
by adjusting the current of the diodes. Figure-2.1.18 shows the plot of these data and
load lines for 10, 15 and 20 dB attenuation. The plot shows that all three load lines are
intersect the voltage axis at nearly 1.2 Volts, is the optimum voltage V opt- This agrees
with the value obtained by parameter extraction method in previous section.
1.0
+60 °C
s - +25 °C
20 dB
o o o Measured
Data
15 dB
0.5
Load Lines
10 dB
/-•■s
1
^
0.0
0.2
0.4
0.6
0.8
1,0
1-2
1.4
Vd (Volts)
Fig.-2.1.18: Measured PIN diode I-V data and load lines to determine V opt-
TUT) Thesis
%eg 9{p 5024.19‘* January, 2004.
37
2 .1 3 .5 .4 .3
Proposed Im proved Control Circuit fo r P IN Diode Based
S tep A ttenuator
The important characteristics of the equiresistance curve of PIN diodes are:
(a) It is nearly linear over a very wide range of temperature. Thus, a simple bias
circuit of bias load line along the equiresistance line can provide temperature
invariant attenuation of the PIN diode based attenuator circuit.
(b) The case where p & 2-m, no load voltage depends upon diode voltage Vj Thus,
for different setting of attenuation, optimum open circuit voltage will be different
and given by the equation (2.1.13A)
QVS
Ral
VoPTI
Rbl
Rj2‘
H
R b2
H
H
Ra
Rb3
H
Ra4
H
4
R»3
RbS
•K
PIN D iode
Circuit
RaN
■K
VoPTN
tr
Digital Data for
Selection of Resistor
Fig.-2.1.19: PIN diode driver circuit for p * (2-m)
‘Thesis
Hp • 5024 IS* January, 2004.
S C ‘B tra ,
38
y itjarat H nw crstty,
In if:
(c) The case where p = 2-m, no load voltage is independent on diode voltage VdThus, for different setting of attenuation, optimum open circuit voltage will be
same and given by equation (2.1.14).
Therefore, diodes with p * (2-m), the control circuit [24] for stepwise variable
attenuation is shown in Figure-2.1.19. Here, optimum open circuit voltages Vom ,
Vopn,......... V optn are derived from a stable voltage source Vs and with bank of potential
dividers. Here, optimum voltages given by:
R■bN
Vv OPTN = Vv s
(2.1.23)
RaN + ^bN
And, equivalent series resistance is given by:
_
RaN X
RaN
(2.1.24)
^bN
This optimum voltage ( V optn) and resistance (RN) through an analog switch will bias
the PIN diode based circuit. The analog switch will provide DC current path connecting
one of the optimum voltages (say Voptn) and associated equivalent resistance (say R N) to
the PIN diode circuit. Analog switch may be a mechanically controlled switch, or digital
switch controlled by digital data. Depending upon digital data, PIN diode circuit will be
biased by a particular current determined by V optn and R n and provide particular
attenuation.
To determine the combination of resistor values R0n and RbN, one has to taken into
account the finite on resistance (say R on) of the analog switch. Thus overall series
resistance of the stepwise control circuit will be (R n + R on).
For the PIN diodes of p = 2-m, the optimum open circuit voltage is independent on
diode voltage V* means VOPT is independent on the value of attenuation setting. Thus,
the bias control circuit must provide a constant no-load output voltage and a varying
output resistance to get temperature insensitive variable attenuation. Therefore, we can
use either (a) a constant voltage source in series with a variable resistor, or (b) a constant
voltage source in series with a bank of pre-adjusted resistors, from which one resistor at
a time is switched into the circuit. The circuit of first option (a) is shown in Figure-
V(i<D Thesis
%eg. 9{o • S0Z4 19‘* January, 2004
S C "Bern, gujarat University, India.
39
2.1.20. The latter option, which is well suited to digital electronic control, is shown in
Figure-2.1.21 [21]-[24]. The constant voltage V0pt is provided by a bandgap reference
with associated trimming resistors.
Vopt
Q
Analog
Switch
R3
R4
Rs
Rn
PIN Diode
Circuit
TT
Digital Data for
Selection o f Resistor
Fig.-2.1.21: PIN diode driver circuit for p = 2-m
!P/t© Thesis
Vj.y. Up • S024 13“ January, 2004
S C Vera, (jujara t University, India
40
Table 2.1.2 [22] shows the test results of PIN diode attenuator with a mechanical
switch as an analog switch. The attenuation accuracy remains within the range of 0.11
dB for all attenuation level up to 30 dB and over the temperature range of -10 to +70 °C.
As analog switch, analog multiplexer such as IC 74HC4051 can be used. This part
has an ON resistance of a few tens of ohms, which must be taken into account when
adjusting the resistor network. Table 2.1.3 [22] shows the test results of the same PIN
diode attenuator, with two numbers of digital multiplexer 74HC4051 as analog switch.
The attenuation accuracy remains within the range of 0.2 dB for all attenuation level up
to 24 dB and over the operating temperature range of -10 to +70 °C.
Table-2.1.2 : Test results of PIN diode Attenuator
(With Mechanical Switch)
Binary
Data
0000
0001
0010
0011
0100
0101
0110
0111
1000
1001
1010
1011
1100
1101
1110
Attenuation
setting
30
28
26
24
22
20
18
16
14
12
10
8
6
4
2
Measure Attenuation
+70 °C
+25 °C
-10 °C
3000
29.98
30.11
28.08
28.00
27.95
26.03
26 00
26.00
24.03
23.99
24.00
22.00
21.99
22.00
19.98
19.99
19.99
17.98
18.01
17.99
15.97
15.99
16.01
13.99
14.00
14.01
11.95
11.98
1199
10.02
9.98
10.00
7 98
8.00
8 01
6 02
5.98
6.00
3.99
4.00
4.01
2.00
201
2.02
Max. Error
dB
0.11
008
0.03
0.03
0.01
0.02
0.02
0.03
0.01
0.05
0.02
0.02
0.02
0.01
0.02
Table-2.1.3 : Test results PIN diode Attenuator
(With Analog Multiplexer, 4051)
(PfitD ‘Thesis
0000
0001
0010
0011
0100
0101
0110
0111
1000
1001
1010
1011
24
22
20
18
16
14
12
10
8
6
4
2
’
Measure Attenuation
+70
+25 °C
°C
23.8
23.9
23.9
22.1
22.2
21.9
20
201
19.8
18.1
18
17.8
15.9
16
15.9
14.2
14.1
14 0
12.2
12.2
11.8
10.1
10.1
9.9
8.2
8.1
7.8
6.1
6.1
5.9
4.1
4.0
3.8
2.0
1.9
2.0
n
Attenuation
setting
o
o
Binary
Data
5024 19 *^ January, 2004-
Max.
Error
dB
0.2
0.2
0.2
0.2
0.1
0.2
0.2
01
0.2
0.1
0.2
0.1
5 C © era , (pujarat ‘University, India
41
Practically, optimum voltage (Vopl) ralized from standard voltage source V„ (say +5V)
with potential divider (Ra, Rb) as shown in Figure- 2.1.22. To avoid high curren drain
by the potential divider, value of resistor Ra and Rb to be kept sufficiently large. But for
high value of these resistors, current to the diode will be limited and may not be
sufficient to drive PIN diodes. To avoid this problem unity gain buffer may be used to
eliminate the effect of this potential divider. The buffer will enable to use high value
-W
i
resistor values of the potential divider without limiting current to the diodes.
Rn
--
-PIN Diode
Circuit
Digital Data for
Selection o f Resistor
Fig.-2.1.22: PIN diode driver circuit to eliminate effect of source resistance
There are different parts used, as analog mux are CMOS IC CD4051 and 74HC4051.
Analog mux CD4051 has high on resistance nearly 250 to 300 Q. at ±5V supply.
Whereas on resistance of 74HC4051 is less, nearly 60 12. Moreover, on resistance of
these ICs has positive temperature coefficient. Thus, use of low on-resistance mux is
preferable.
2>fi© Thesis
t f S 'Hfi- S024 19* January, 2004
S C Tern, gujarat University, India
42
One way to compensate the effect of temperature coefficient of mux resistance is to
determine modified Vop, to take into account the effect of on-resistance. In graphical
method as discussed in previous section, plotting attenuator current (Iam) versus addition
of attenuator voltage (Vam) and voltage drop across the switch (V™) at three different
temperatures, and determining the intercept point of the load line to voltage axis (Vam, +
Vsw), modified Vop, can be determined. Using the modified Vopt for biasing, effect of
temperature coefficient of switches on-resistance can be compensated.
Another way to eliminate the effect of on-resistance of the switch and potential
1divider is the use of analog buffer circuit for each step values as shown in Figure.2.1.23. This circuit will eliminate the effect of on-resistance of mux. This circuit will
also eliminate the current limitation by the use of high resistor values of potential
divider.
Q vs
D
Ri
>
■
5>
VOPT
$ ■
r2
r3
R4
&
Rn
&
PIN Diode
Circuit
tr
Digital Data for
Selection of Resistor
Fig.-2.1.23: PIN diode driver circuit to eliminate effect of on-resistance of switch
271® Thesis
X ty ’H p •' 5024 19li January, 2004
S C Sera, Qujaral University, India.
43
2 .1 .4 .
CONCLUSION
A hitherto unexplored mathematical property of PIN diode has been discussed. This
property is exploited in a simple and easily adjusted stepwise bias control circuit that
provides excellent attenuation setting accuracy and temperature stability. It is shown that
this approach can be used for a wide range of practically available diodes. The circuit
uses no separate temperature sensor or compensating mechanism, but responds directly
to the junction temperature of the diodes. This prevents any errors caused by temperature
gradients, or by self-heating of the diodes due to high RF power levels.
Vk'D Thesis
9(o . 5 0 2 4 . 1 9 January, 2004
S C S e r a , Qujarat U niversity, India
44
2.2
SCHOTTKY BARRIER DIODE
Schottky barrier diodes are widely used for microwave applications. This device is
used as RF detector, attenuator, mixer, limiter, etc [26], [27]. This is also used to
generate non-linear characteristic for linearizer applications, and as microwave phase
shifter for microwave beam forming network [28] - [36]. DC I-V characteristic, Cut-in
voltage and RF resistance of Schottky barrier diode is known to vary with temperature.
Thus, attenuation, limiting power level, non-linear characteristic and phase shift of the
Schottky diode based circuits will change with temperature.
Schottky barrier diodes are conventionally biased by constant current source as well
as by constant voltage source for various RF applications though in both the bias
conditions RF resistance changes with temperature [28]- [36]. The thesis work offers a
fresh look into the temperature behavior and compensation mechanism of Schottky
barrier diode at radio frequency operation by designing various Schottky barrier diode
based circuits [39] - [41] operating at microwave frequency.
RF resistance of Schottky diode changes drastically when it is biased by constant
voltage source and there is comparatively less variation when the diodes are biased by
constant current source. Here it is shown that, temperature invariant RF resistance of the
Schottky diode can be achieved by proper selection of the bias load line. Thus, without
any separate temperature sensor and compensation circuits, as used in conventional
temperature-compensation scheme, it is possible to achieve temperature compensated RF
resistance of the Schottky diode. Mathematically it is shown, and verified by the
measurements that [21], [41], variation of RF resistance, which is nearly 24 % in case of
fixed current bias over the operating temperature range of -10 to +60 °C, can be
minimized to nearly zero by the proposed optimum load line biasing scheme.
Temperature characterization and compensation of Schottky barrier diodes performance
over RF power level is also presented [21 ], [41].
‘Thesis
‘T ip : 5 0 2 4 .19‘* January, 2004.
S C ® era, Qujarat ‘U niversity, India.
45
2 .2 .1 .
P R IN C IP L E O F O P E R A T IO N
Schottky barrier diodes are used for detector, mixer, linearizer, harmonic generator,
etc [26] - [36]. It contains a metal-semiconductor contact by deposition of a metal layer
on a semiconductor as shown in Figure-2.2.1. The semiconductor material may be GaAs
or silicon. The heavily doped N+ layer is used to reduce the series resistance Rs.
Schottky
Contact
Ohmic
Contact
Fig.-2.2.1: Schottky diode structure
Energy band diagram [1], [26] of an n-type Schottky barrier diode is shown in
Figure-2.2.2, for n-type Schottky diode, the work function of metal (<f)m) is higher than
that of semiconductors (<}»s). When metal and semiconductor comes into contact, then
electron flows from semiconductor to metal to establish the thermodynamic equilibrium.
n type semiconductor
t
t
qz
q<te
Ec
-t"
E fs
metal
q't’m
-F m -
>
<f>S
Fig.-2.2f2: Energy band diagram of Schottky diode
A depletion region (W) formed adjacent to the metal in the semiconductor as shown in
Figure-2.2.3. The positive charge due to uncompensated donor ions within W, matches
the negative charge on the metal.
<PHV Thesis
%f.g Via.. 5 0 2 4 .1 9 “ January, 2004
S C V e ra , Qujarat U niversity, India
46
SEMICONDUCTOR
_
+++++*+
EE+++++++
= +4- +++ - M •*— w —+
Fig.-2.2.3: Depletion layer
Due to the charge transfer, Fermi levels align and barrier potential ($,, =
- %)
developed between metal to semiconductor junction as shown in Figure-2.2.4. Resulting
the equilibrium contact potential V0, which prevents further net electron diffusion from
the semiconductor conduction band into the metal, is the difference in work functions
and given by Va = ?((#»- $)■
METAL
semiconductor
Fig.-2.2.4: Fermi level alignment
The potential barrier height
for electron injection from the metal into the
semiconductor conduction band is ((j)m - %), where q% (called the electron affinity) is
measured from the vacuum level to the semiconductor conduction band edge. The
potential difference Va can be decreased or increased by the application of either forward
or reverse bias voltage.
When a forward-bias voltage V applied to the Schottky barrier, the contact potential is
reduced from Va to (V0-V). As a result, electrons in the semiconductor conduction band
can diffuse across the depletion region to the metal. This gives rise to a forward current
(metal to semiconductor) through the junction. Conversely, a reverse bias increases the
barrier to (V0+Vr), and electron flow from semiconductor to metal becomes negligible. In
2>A® Tfitsis
<Ho : 5024 19<*January, 2004
s C (Bern, gujarat U niversity, India
47
either case flow of electrons from the metal to the semiconductor is retarded by the
barrier
Thus, the forward current is due to the injection of majority carriers from the
semiconductor into the metal. The absence of minority carrier injection and the
associated storage delay time is an important feature of Schottky barrier diodes.
Although some minority carrier injection occurs at high current levels, these are
essentially majority carrier devices. Their high frequency properties and switching speed
are therefore generally better than typical p-n junctions.
Fig.-2.2.S: RF equivalent circuit of forward biased junction diode
Simplified RF equivalent circuit [27] of a Schottky barrier diode is shown in Figure2.2.5. Here, Rj is the junction resistance of the diode and Q is the depletion layer
capacitance. Rs is equivalent series resistance of the diode caused by the part of N layer,
N+ layer and Ohmic contact.
2 .2 .2 .
TEM PERA TU R E CHARACTERISTICS
The forward DC current of the Schottky diode is related to the forward voltage by this
well-known equation [1], [27] (neglecting a term corresponding to the reverse saturation
current):
Id(Vd,T) = A T 2exp v(vd I A
rjkT
*Pk*D 'Thesis
%pg ${p , 5 0 2 4 .19s6 January, 2604
( 2 .2 . 1)
S C (Bern, Qujarat 'University, India.
48
where, q is the electron charge, k is Boltzman’s constant, fa is the barrier height in Volts,
7} is the ideality factor and Rs is the equivalent series resistance of the diode. A is the
product of modified Richardson constant with junction area of the diode.
At small signal operating condition, RF resistance Rd of the diode will be given by [27]:
RA,- TH k - - z-7 jb )
"
<2-2 -2)
here, z = ijk/q is a constant independent on temperature. Equation (2.2.2) shows that, RF
resistance of Schottky diodes linearly increases with the increase of temperature when
the diodes are biased through constant current source i.e. Id(Vd,T) maintained constant
over the operating temperature range.
"
oi
o
~
-10
10
30
so
70
8
o
o
o
o
Rj (Normalized)
Fixed Voltage bias
15
i
Rd (Normalized)
Fixed Current Bias
1.5
Temperature (°C)
Fig.-2.2.6: Simulated RF resistance variation with temperature at
fixed current bias and fixed voltage bias.
Figure-2.2.6 shows the simulated RF resistance (normalized to the value at 30 °C)
variation over the diode temperature variation of -20 to +80 °C. This figure shows that
RF resistance increases with the increase of temperature when diodes are biased by
constant current source but RF resistance decreases drastically when the diodes are
biased by a constant voltage source bias. Thus, performance of the diode based RF
circuits, where constant voltage bias used, will be affected severely over the change of
temperature. Whereas, the performance of constant current bias circuits, will be less
affected with change of temperature.
S’<© 'Thesis
%f.g. Tifi.: 5024. 19“ January, 2004.
S C Vera , Qujarat University, India.
49
Schottky diodes are used for detector, mixer, linearizer, harmonic generator, etc [26,
36]. Among various types of linearizer [28]-[38], Schottky diodes are used as linearizer
[28] - [36] where RF resistance of the diode determines the performance of the circuit.
Especially, Schottky diodes are used as distortion generator for RF/microwave linearizer
applications for its low power consumption and compact size.
In linearizer applications, RF resistance of Schottky diode increases or decreases with
the increase of RF power level and generates amplitude and phase non-linearity to
compensate the nonlinearity of the high power amplifier. Typical I-V characteristic of a
Fig.-2.2.7: I-V characteristic and load lines of Schottky diode in
presence of RF power
Schottky diode in presence of RF signal is shown in Figure-2.2.7. Two types of bias
load line for nearly constant current bias and constant voltage bias are also shown in the
figure. It is clear from the figure that RF resistance of diode increases with the increase
of RF power level when the diodes are biased by constant current bias condition.
Whereas, RF resistance of diode decreases with increase of RF power level when the
diodes are biased with constant voltage bias source. This property enables two types of
diode-based linearizer, one is series diode linearizer [28], [29], and another is parallel
diode linearizer [30], [31]. As shown in Figure-2.2.6, series diode based linearizer will
be affected severely in temperature varying environment, since these are biased by a
constant voltage source. Comparatively, there will be less performance variation of the
constant current biased parallel diode linearizer.
I’fl'D Thesis
%f3
5024 19“ January, 2004
S C ‘B era, Qujarat U niversity, India.
50
2 .2 .3 .
TEMPERATURE COMPENSATION TECHNIQUE
RF circuits based on RF resistance of Schottky barrier diodes are affected severely
with change of temperature under constant current or constant voltage bias conditions.
However, there are many reported Schottky diode based circuits operated with constant
voltage and constant current bias condition. Generally, conventional temperature
compensation techniques as used for PIN diodes also can be used for Schottky barrier
diode based circuits. These schemes uses temperature sensor to sense the temperature of
the diode and produce a signal that is function of temperature. This signal modified the
current applied to the diode in such a manner that it maintains the same RF resistance of
the Schottky diode at all temperature. Some of the compensation mechanism based
entirely on analog control signal, while others contain digital control signal acting at
discrete temperature steps as discussed in the previous chapter. However, the successful
adjustment of such control circuits is a time consuming task, often involving much trial
and error; and the circuits themselves are relatively complex.
The thesis work offers a fresh look into the behavior of Schottky diodes over
temperature. A simple but accurate biasing scheme for Schottky diode based RF circuits
that depends upon the RF resistance of the Schottky diode is described. The circuit can
easily be optimized for a high accuracy over a wide temperature range. It is investigated
some mathematical consequences of the Schottky diodes equation and shows that a very
simple bias network, designed on this principle, can provide a very high temperature
stable performance over very large range of temperature. This temperature stability is
achieved without using a temperature sensor or compensating circuit. Practical
implementation and test results for such circuits are also presented.
2.2.3.I.
P ro p o s e d N o v e l T e m p e r a tu re -C o m p e n s a tio n T e c h n iq u e
Now here it is investigated the voltage and current requirement to achieve the
temperature invariant Schottky diodes RF resistance.
To achieve constant RF resistance say
Rd =
/?*„ the current-voltage relationship will
be given by the equation (2.2.3).
Thesis
%fg, 3{p.. 5024 19“ January, 2004
S C •Beta, Qujarat University, India
51
Id(Vd,T)=AT*Exp
g(vd - I dR , - ? B)
ijkT
(2 .2 .3 )
where Vj(T) will be given by the equation (2.2.4).
Vj(r)=A+i ^ +^
•(2.2.4)
Rdo
O
Ii(mA)
Ol
q
r*^
O
Q
Load Line
°
Vd (Volts)
0.8
04
Fig.-2.2.8: Equiresistance curve and optimum load line.
R
9
Fig.-2.2.9: Simple bias circuit o f Schottky diode.
The plot o f the equation (2.2.3) with the constraint for temperature invariant RF
resistance given by equation (2.2.4) is shown in Figure-2.2.8 over the temperature range
o f -273 to 300 °C. This curve is called equiresistance curve o f the Schottky diode, which
intersect the voltage axis at the barrier height voltage ($ ,) o f the diode. From this plot, it
is clear that the equiresistance curve can be considered as a straight line over a very wide
!Pfi?D Chests
9{p • 5024 19t&January, 2004,
S C S e r a , (Jujarat ‘University, India
52
\
range of temperature. Therefore, a simple bias circuit as shown in Figure-2.2.9 can
provide temperature invariant RF resistance of the diode if the load line of the bias
circuit passes along the equiresistance line over the temperature range of interest.
Figure-2.2.8 also shows such a bias load line.
2 .2 .3 .I.I.
T h e o re tic a l D e te r m in a tio n o f O p tim u m B ia s L o a d - U n e
Combining equation (2.2.2), (2.2.3) and (2.2.4) and elimination T we can write
another form of equiresistance curve as:
Vj
= 0b
.Ln
+ IdR. +■
z
\ R do J
•(2.2.5)
A.Ia
According to above discussion, the equation of optimum bias load line will be the
tangent of equiresistance curve.
Differentiate the constant resistance locus, equation (2.2.5), we can write the slope
(M0) of the curve at the bias point (Vdo, ho) corresponding to temperature T0 as:
1
31
d V dQ - I dQRS - ^ )
VkT0
9 Ido
/
( 2 .2 .6)
Thus, the equation of tangent to the equiresistance curve passing through the bias
point (Vdo, Ido) at temperature T0 corresponds to RF resistance Rio is given by:
y d = y do+-^r {id - i do)
Mo
(2.2.7)
Putting the value of M0 from equation (2.2.6) to equation (2.2.7), we can write the
equation of tangent to the equiresistance curve at temperature T„ corresponding to diode
bias (Vdo, Ido) is given by:
vd(id)=vd0+ its
L
/
i - j Ll
+ /< * ,* > - - «
9
JV
J
(
- V
d
o
( 2 .2 .8)
* d o
Taking this tangent to be our optimum bias load line, its voltage intercept Vopt can be
obtained by putting h - 0 in equation (2.2.8):
‘Ifiesis
7{o . S024. I S '6 January, 2004.
S C fBera, Qujarat University, India.
53
The voltage intercept is seen ideally independent o f the selected bias point (Vdo> ho),
( 2-2.9)
9
implying that the optimum no-load output voltage o f the bias circuit is same fo r all RF
resistance values i.e. fo r all attenuation values.
Second term o f this equation is nearly 0.026 Volts, which is negligible compared to
the barrier height o f the diode <j>g. Thus the optimum open circuit voltage is depends only
upon the physical parameter, barrier height, o f the Schottky diode.
2 .2 .3 .I.2 .
Accuracy o f Optim um Bias Load-Une Technique
Over Tem perature
Now we will investigate the accuracy of the optimum bias load line technique over
the operating range of temperature. Temperature coefficient of RF resistance variation,
say Tc !°C, of the diode can be written from equation (2.2.2) as:
i
t
i
i , ar
(2.2. 10)
From the bias circuit of Figure-2.2.9, we can write
/-*A =
-y -A
R
(2 .2 .11)
Combining equation (2.2.10) and (2.2.11), we can write
i
T = - +
T V - V d dT
(2 .2 .12)
Differentiating equation (2.2.4) with respect to temperature T we can write:
3V, _
dT
<j>b - V d
T
rjk
q
JjkT dRd
qRd dT
(2.2.13)
Putting this expression in equation (2.2.12), we can write
tPA® 'Ifirsts
J C H era, gujarat U niversity, India.
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54
T = J_ V —0B —rjkT!q
c T V - V d +r]kT/q
(2.2.14)
This is the expression for temperature coefficient of RF resistance variation of the
diode.
When, Schottky diodes are biased by constant current source, then V—» °°, and Tc =
1/T / °C.
Now, when Schottky diodes are biased by proposed optimum bias load line condition
then putting V = V opt in equation (2.2.14) from (2.2.9), temperature coefficient of RF
resistance will be:
1
°
(1- T I T 0)nkT0l q
T{</>B - V d)+{l + T / T 0)rjkT0 / q
This expression shows that at T ~ T0, temperature coefficient of RF resistance
becomes zero. Now we will investigate the temperature coefficient at other temperatures
for Schottky diode, putting typical parameter values of the diodes.
For silicon Schottky diode, putting
= 0.69 Volts, T) = 1.2, T0 = 298 K (25 °C),
temperature coefficient will increases from 0 to 0.010 times when temperature changes
from 25 °C to 25 ± 45 °C with respect to the RF resistance variation in fixed current
biasing condition.
2 .2 .3 .I.3 .
E x p e rim e n t A n d T est R e s u lts
Figure-2.2.10 shows the schematic circuit diagram of the circuit that we have used for
experiment. Isolated port of a 3-dB hybrid coupler is used as the output of the circuit,
two diodes are connected at the direct port and coupled port of the hybrid and other port
is used as the input port. R is load resistor to drive the circuit from a voltage source V.
Depending upon RF resistance of the diodes, determined by the bias current through the
diode, some portion of RF power will be dissipated within the diode and rest of the
power will be reflected from the diode. The reflected power from both the diodes will be
combined at the output port of the circuit. Thus, S21 of the circuit is given by:
•PHD Thesis
‘Keg. Oio.: 5024. 19,i January, 2004.
S C (Bera , Qujarat U niversity, India.
55
Fig.-2.2.10\ Schematic circuit diagram
S21 = (Zd - ZB)/(Zd + Za)
(2.2.16)
where, Zj is the RF impedance of the diode, i.e. parallel combination of RF resistance Rd
and capacitance Cd with series combination of series resistance Rs. The circuit is realized
in 25 mil alumina substrate (er = 9.9). Lange coupler is used for 3-dB coupler at the
centre frequency of 3.7 GHz and layout is made for beam lead diodes. The photograph of
the realized circuit is shown in Figure-2.2.11.
The circuit is used with two numbers of beam lead medium-barrier Schottky diode of
type MSS-40,148-B10B of Metalics for the experiment. The circuit parameter
Fig.-2.2.11: Photograph of the diode circuit.
•Ph'D Thesis
%ty. 9{p.: 5024. 19,k January, 2004.
S C 'Bera , Qujarat University, India.
56
Fig.-2.2.12: Measured Schottky diode V-I data and load lines
for different attenuation at Pm= - 30 dBm
‘attenuation’ is measured to verify the RF resistance variation of the diode with change
of temperature. At forward bias condition of the diode, the temperature effect of parasitic
capacitance Q and series resistance Rs can be neglected compared to the effect of RF
resistance Rj.
Figure-2.2.12 shows the plot of measured voltage and current data of the Schottky
diodes for 4, 8 and 12 dB attenuation at three different temperatures of +60, +25 and -10
°C. These data are taken at -30dBm RF power input to the circuit. It is clear that the V-I
data of each attenuation value for different temperatures lies in a straight line. Load lines
are drawn for the attenuation of 4, 8 and 12 dB. All the load lines are intersecting the
voltage axis at the same point of value nearly 0.72 Volts, which is nearly equal to the
barrier potential (0.69 Volts) of the used Schottky diode.
Figure-2.2.13 and Figure-2.2.14 shows the plot of the similar measurements data as
of Figure-2.2.12 but at different RF input power levels of -25 and -20 dBm respectively.
Theses figures show that the V-I data of each attenuation value for different temperatures
also lies in a straight line for both the RF power level but intercept point of the load line
to the voltage axis increases with the increase of RF power level. The voltage axis
intercept i.e. optimum open circuit voltages are 0.80 Volts and 1.05 Volts for input
power level of -25 dBm and -20 dBm respectively. Increase of optimum open circuit
voltage with the increase of RF power level is due to the rectification effect of the diode
that is not taken into account in the equation (2.2.2) of the diodes RF resistance. Thus,
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5024 19* January, 2004
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57
self-biasing effect increases the requirement of open circuit voltage of the bias network
with the increase of RJF power level.
Measurement shows that with optimum load line bias, with the open circuit voltage of
0.72, 0.80, and 1.05 Volts corresponds to the RF input power of -30, -25 and -20 dBm
respectively, the attenuation variation remains within ±0.2 dB for all the attenuation level
of 4, 8 and 12 dB, over the temperature range of -10 to +60 °C.
Fig.-2.2.13: Measured Schottky diode V-I data and load lines for
different attenuation at Pin = - 25 dBm
Fig.-2.2.14: Measured Schottky diode V-I data and load lines for different
attenuation at P;n = - 20 dBm
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58
Thus, the proposed optimum bias load line technique ideally compensates the
temperature variation of RF performance at small signal operating condition. The
proposed technique also provides temperature insensitive RF resistance of the diode at
large signal operating condition but requirement of optimum open circuit voltage is a
function of RF power level due to the self-biasing effect of the Schottky diodes
rectification effect.
Therefore, to achieve temperature compensated RF performance of the Schottky diode
based RF circuits, based on the RF resistance of the diode; it is required to change the
optimum voltage over the operating range of RF power level. For that, sampling and
detection of RF power level is required and to generate a control signal to control the
(ap) /izs/
Ol
Oi
optimum open circuit voltage of the bias circuit as a function of RF power level. Thus for
-20
-35
-25
-15
-5
RF Power Level (dBm)
-95
60
■3
-105
-115
,
-35
-25
-15
-5
RF Power Level (dBm)
Fig.-2.2.15: M easured S 21 variation over R F pow er level f o r
= 0.75 Volts
SU'D ‘Ifiesu
%tg. 9(p ■5024 19* January, 2004
Vopt
S C Hera, Qujarat University, India
59
perfect compensation, bias circuit requirement will be complex. We will investigate the
deviation of the RF performance, by using simple bias circuit which will provide
constant open circuit voltage for entire operating RF power level.
Figure-2.2.15 shows the measured temperature variation of magnitude and phase of
S21 of the circuit of Figure-2.2.11, for Von = 0.75 Volts. Measurement shows that,
amplitude variation remains within ±0.6dB and phase variation remains within ± 2 deg
over the temperature range of -10 to +60 °C and over the entire operating RF power level
of -30 to -10 dBm. It is also clear that, attenuation and phase variation over the
temperature range is negligible at the small signal RF level condition.
2 .2 .4 .
CO NCLUSIO N_________________________________
Here an unexplored optimum bias load line technique is discussed to achieve
temperature invariant RF resistance of Sehottky barrier diodes. The proposed optimum
load line technique compensates the temperature variation of RF performance at small
signal as well as at large signal operating condition. However, requirement of optimum
open circuit voltage is a function of operating RF power level. The dependence of Von
on RF power level is for self-biasing effect due to the rectification effect of the Sehottky
barrier diode at high RF power level. It is also shown, fixed optimum open-circuit
voltage may also be used over die entire RF power level, wherever circuit operates
within small RF power range or small performance variation is acceptable. Measurement
shows that this novel bias technique leads to temperature invariant RF resistance without
using any separate temperature sensor and compensation circuits as used in conventional
temperature-compensation circuits. In this case, circuit responds directly to the junction
temperature of the diode thus, there will be no compensation error due to temperature
gradient and self-heating of the diode under high power operation.
SfaD Thesis
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2.3
LIGHT EMITTING DIODE
The temperature compensation scheme used to compensate PIN and Schottky barrier
diode as discussed in the previous chapters is extended for temperature compensation of
output light intensity (brightness/iUumination) variation of Light Emitting Diode (LED).
Solid-state lighting using light emitting diode (LED), has many advantages over
traditional light sources, such as longer life, higher efficiencies, low voltage operation,
fully dimmable without color variation, dynamic color control, no heat or UV in the light
beam, instant on, robust, vibration proof, etc [42]. Elimination of use mercury in the
LEDs, unlike most fluorescent sources, will enable to meet new and future increasingly
stringent environment regulations. Thus, solid-state lighting technology using LED, not
only leads to energy and environment savings, but also it will change the way we think
about lighting. The unique features of LEDs make them very attractive to many
industries including automobiles, traffic signal lighting and artificial lighting, similar to
natural daylight.
However, one of the drawbacks of the LED lighting technology is that the brightness
of an LED, operated with a fixed current bias, is greatly affected by the junction
temperature of the LEDs. Light intensity of LED decreases with increase of diode
temperature at fixed current bias condition. Temperature dependency of light intensity of
the light emitting diode (LED) is the problem in using these devices in their many
applications where LEDs are exposed to the wide ambient temperature variation.
Nowadays, LEDs are used widely for signal lighting in automobile applications. During
hot summer day at noon, LEDs light becomes dimmest contrary with the requirement of
brightest intensity, and during cold winter day at night, LEDs light becomes brightest
contrary to the requirement of dimmest intensity. In color printer application, light
intensity fluctuation results in a change of color. In the measuring instrument and camera
calibration applications, where it is used as a reference intensity light source, intensity
variation creates measurement errors.
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9{o • S0Z4 19' * January, 2004
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There are various analog and digital techniques to overcome these problems. Firstly,
by detecting the diode temperature or ambient temperature using temperature sensor
controls the current so as to maintain the constant intensity of LED [44] - [46], this
requires complex circuits with temperature sensor and temperature gradient between
diode and sensor results in compensation error. Secondly, by sampling output light
intensity of the diode and using feedback technique controls current to maintain constant
brightness, however, this technique is complicated and requires more hardware to
implement. Thirdly, modifying the diode structure such as using an optical filter of
wavelength dependent transmittance [43] temperature coefficient can be minimized.
However, filter loss decreases the available light intensity from the diode leads to
decrease of overall efficiency of the LEDs.
The temperature compensation scheme used to compensate PIN and Schottky barrier
diode is extended for temperature compensation of output light intensity variation of
Light Emitting Diode (LED). In this chapter, it is investigated the temperature behavior
of the diode parameters mathematically and exploited the temperature variation of diode
current to improve the temperature dependency of the LED’s light intensity [47], [48].
Here it is shown that the equiintensity curves of the LED can be considered as nearly a
straight line over the operating range of diode temperature. Therefore, by selecting the
best fitted load line along the equiintensity curve leads to nearly temperature invariant
light intensity. Test results confirms the same and verified that by selecting optimum
load line, peak to peak light intensity variation decreases from 99% to 6% over the
temperature range of -20 °C to +80 °C. This temperature stability is achieved without
using a temperature sensor or compensating circuit. Diode currents will be modified
according to the junction temperature of the diode itself. Therefore, there will be no such
error due to temperature gradient between junction of the diode and temperature sensor
as happened in case where temperature sensors are used.
2 .3 .1 .
OPERATION OF LIGHT EMITTING DIODE
Light Emitting Diodes (LEDs) are semiconductor p-n junctions that under proper
forward-biased conditions can emit external spontaneous radiation in the ultraviolet,
visible and infrared regions of the electromagnetic spectrum [1]. When carriers are
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injected across a forward-biased junction, the current is usually accounted for by
recombination in the transition region and in the neutral region near the junction. In a
semiconductor with an indirect band gap, such as Si or Ge, the recombination releases
heat to the lattice. On the other hand, in a material characterized by direct recombination,
considerable light may be given off from the junction under forward bias. This effect,
called injection electroluminescence, provides an important application of diodes as
generators of light.
The effectiveness of light for stimulating the human eye is given by the relative eye
sensitivity V(X), which is a function of wavelength. For the maximum sensitivity of the
eye at 0.555 pm, V(0.555) = 1.0; the value of V (X ) falls to nearly zero at the extremes of
the visible spectrum at 0.39 and 0.77 pm.
Since the eye is only sensitive to light of energy h v > 1.8 eV (-0.7 pm),
semiconductors of interest must have energy bandgaps larger than this limit. Direct
bandgap semiconductors are particularly important for electroluminescent devices,
because the radiative recombination is a first-order transition process and the quantum
efficiency is expected to be much higher than that for an indirect bandgap
semiconductor, where a phonon is involved.
A good example of the variation in photon energy obtainable from the compound
semiconductors is the material Gallium-Arsenide-Phosphide (GaAsP), which is a
combination of GaAs and GaP. As the percentage of As is reduced and P is increased in
the material, the resulting band gap varies from the direct 1.43 eV gap of GaAs (infrared)
to the indirect 2.26 eV gap of GaP (green). The symbol for the resulting material is
written as GaAsi-xPx, where the x represents the fraction of P atoms. The band gap of the
resulting material varies almost linearly with x until the 0.44 composition is reached, and
electron-hole recombination is direct over this range.
2 .3.2.
TEMPERATURE CHARACTERISTICS
Solid-state lighting using LEDs is more efficient to generate visible light than many
filament type light sources. However, the heat energy developed during operation of
LEDs does not radiate away from the LED, but conducts back into the semiconductor
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and increase the junction temperature of the diodes. Ambient temperature also varies
from day to night and from one season to another season.
Thus, LEDs junction
temperature varies due to self-heating of the diode as well as due to variation of ambient
temperatures. We will investigate how brightness of LEDs changes with temperature.
For a given input excitation energy, the radiative recombination process is in direct
competition with the non-radiative process. The quantum efficiency T]q is the fraction of
the excited carriers that combine radiatively to the total recombination and may be
written in term of the lifetimes as [ 1 ]:
,
9
= R r _
*nr
R
*nr+ rr
(2.3.1)
Where, Tnr is the nonradiative lifetime and tr is the radiative lifetime, and Rr and R
are the radiative recombination rate and total recombination rate, respectively. The
external efficiency decreases with the increase of temperature. Thus, output light
intensity of LEDs decreases with the increase of temperature.
Output light intensity of a LED varies as the exponential function of temperature (7)
[43] and also varies linearly as a power p of the forward diode current (/<*):
L{ld,T)=z[ld{Vd,T)]pexp(TcT)
(2.3.2)
Where, z, Tc and p are the constants for a particular diode, for most of the cases p =1
and typical values of Tc lies between -0.007 to -0.01 / °C. Tc is the temperature
coefficient of brightness variation under constant current bias condition.
Forward diode current is the function of forward biased diode voltage Vd and diode
temperature T (neglecting a term corresponding to the reverse saturation current) [1],
[26]:
I d(Vd, T ) = A T " Qxp
q{Vd - I dRs - E b)
ijkT
(2.3.3)
A and n are the constants; q is the electron charge, k is Boltzman’s constant, Eb is the
band-gap potential of the semiconductor in volts,
77 is
the ideality factor and Rs is the
series resistance of the diode.
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64
Relative brightness
w.t.tat 30 °C
Fig.-2.3.1: Simulated brightness variation with temperature for
fixed voltage bias and fixed current bias condition.
Figure-2.3.1 shows the simulated brightness variation over the diodes’ junction
temperature of -20 to +80 °C under fixed current and fixed voltage bias condition, taking
temperature co-efficient Tc = -0.01/°C. Simulated result shows, under fixed current bias
condition, peak to peak brightness variation is nearly 104% with respect to the value at
30 °C. Whereas, under fixed voltage bias condition the brightness variation becomes
nearly 200%.
2 .3 .3 .
TEMPERATURE COMPENSATiON TECHNIQUE
In many cases, temperature stabilization of LED brightness can be realized by
maintaining the LED at constant temperature. This method cannot be used in cases
where LEDs are exposed to varying temperature environments. There are various analog
and digital techniques to compensate the temperature variation of LEDs brightness.
Firstly, by detecting the diode temperature or ambient temperature using temperature
sensor controls the current so as to maintain the constant brightness of LED [44] - [46],
this requires complex circuits with temperature sensor and temperature gradient between
diode and sensor results in compensation error.
Secondly, sampling the light intensity of the LEDs and controlling the current of the
LEDs to keep constant output light intensity. This is a closed-loop feedback control
system, which takes care the brightness variation due to temperature or by any other
I h D 'Thesis
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65
means.
Thirdly, modifying the diode structure such as using an optical filter of wavelength
dependent transmittance [43] temperature coefficient can be minimized. However, filter
loss decreases the available light intensity from the diode leads to decrease of overall
efficiency of the LED.
2 .3 .3 .I.
P ro p o s e d N o v e l T e m p e r a tu re C o m p e n s a tio n T e c h n iq u e
Simulation result shown in Figure-2.3.1 shows one interesting feature of the
brightness variation over the temperature is that “under fixed current bias condition
brightness of LEDs decreases with increase of temperature whereas it increases under
fixed voltage bias condition”. This feature suggests that, there should be an optimum
biasing condition, which may give temperature invariant brightness of LEDs. Combining
these equations (2.3.2) and (2.3.3), we can investigate the l-V relationship to achieve
temperature invariant light intensity.
To achieve temperature invariant light intensity of LED, bias network must meet
current requirement to achieve constant light intensity say L0 over the operating
temperature range. Therefore, combining equation (2.3.2) and (2.3.3) and putting L = L0,
we can write equation (2.3.4).
I d(vAT),T) = A T nexp
q{Vd{T )-Id{Vd{T\T)Rs-Eb)
rjkT
(2.3.4)
Where, Vd(T) is given by.
V„(T)-Eb + Rs
p
zexpfcr)
JjkT _
+ - — Ln
qp
zApT npexp(rcr )
(2.3.5)
This parametric equation is the equation of the ideal bias-point locus that will satisfy
L = L0 across the operating temperature range. We can call this locus the “Equiintenstiy
curve” [47] for the required intensity L„.
Figure-2.3.2 shows the plot of equation (4.4) over the temperature range of -20 to
+80 °C, taking, Eb = 2.1 Volts, T) = 2.0, n =2, fis = l f l , p = l and Tc = -0.01 / °C. This
constant light intensity curve shown in this figure can be considered as a straight line
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66
within this temperature range. Therefore, selecting the best fitted load line along the
equiintensity curve, over this operating temperature range, will provide the required
current to achieve nearly temperature invariant light intensity. Such a load line is shown
in this Figure-2.3.2. This load line intersect the voltage axis at a point marked as V opt
will be the open circuit voltage. The inverse slope of this load line will be the open
circuit resistance (in means of Thevenin’s theorem) of the bias network of LED.
Figure-2.3.3
shows the equiintensity curves for different intensity levels
corresponding to bias current of 20, 16 and 12 mA at 30 °C. Load lines correspond to
these three light intensity levels are also drawn. It shows a very interesting characteristic
of the load lines that “ all the load lines intersect the voltage axis nearly at the same
point” marked as V opt■This characteristic implies that open circuit voltage of the bias
network will be same for different light intensity levels.
Figure-2.3.4 shows the equiintensity curves for different values of temperature
coefficient Tc, and Figure-2.3.5 shows the same for different values of current exponent
p over the temperature range of -20 to +80 °C.
30
O
O
Id(mA)
20
Fig.-2.3.2: Equiintensity curve and load line
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67
20 mA -
Tc = - 0.0085/ °C
^
_____________________
16 mA —
—
P= 1
\
Id(mA)
12m A ---load lines
S
1-6
Vd(Volt)
V OPT
18
20
Fig.-2.3.3i Equiintensity curves and load lines for different
brightness levels.
Id(mA)
30
1 .7
1.8
1 .9
Vd(Volt)
Fig.-2.3.4: Equiintensity curves for different values of
temperature coefficient (Tc)
Id(mA)
30
10
1.7
1.8
Vd (Volt)
1.9
Fig.-2.3.5: Equiintensity curves for different values of p.
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R
LED
Fig.-2.3.6: A simple bias circuit of LED
From Figure-2.3.3, 2.3.4, 2.3.5 it is clear that for all practical values of the diode
parameters the equiintensity curves can be considered as a straight line over the
operating temperature range of the light emitting diodes.
Thus the load line of a simple bias circuit as shown in Figure-2.3.6 can be adjusted
along the equiintensity curves with minimum deviation within the temperature range of
interest and will then maintain the desired output light intensity with low temperature
coefficient. The value of the open circuit voltage source V will be V0pr, the load line
intercept point with voltage axis, and resistance value of resistor R will be Ro, the
inverse slope of the load line as shown in Figure-2.3.2.
2 .3 .3 .1 .1 .
Theoretical D eterm ination o f O ptim um Bias Load Line
Combining equation (2.3.4) and (2.3.5) and eliminating parameter ‘T we can write
another form of the equiintensity curve given by equation (2.3.6).
Vd = E b + I dRs +
(2.3.6)
The slope M0 of this equiintensity curve at the diode voltage V* and corresponding
current ho at temperature T0 is given by equation (2.3.7).
1
M
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o
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69
Therefore, equation of tangent of the equnntensity curve at point (Vdo, Ido)
corresponds to temperature Ta is given by equation (2.3.8).
R , p{Eb + I doRs -V do) , TJkTn f | np A
T
T ,
do
I cx o
TcT0Ido
[/„-/J
(2.3.8)
Taking this tangent to be our optimum load line, the voltage axis intercept of the load
line is given by equation (2.3.9).
V
= Vr do —1I d o lRy s — p { Eb + IdoRs-Vjo)
v OPT
TCT„
{
VkT0 1 + np
q \
TCT0 ^
(2.3.9)
This optimum voltage Vopt will be the open circuit voltage of the bias network of
Figure-2.3.6. Putting voltage drop across the junction of the diode VdJ0 =Vdo- IdoRs, the
optimum voltage will be given by (2.3.10) [47].
V
- V
v OPT
v djo
p(Eb ~ v J
TT„
ykT0 ( , np
TT
(2.3.10)
And resistance of the bias network will be given by (2.3.11). This resistor will also
be the current limiting resistor of the diode.
V
-V
R o = lPPT---- y_do_
(2 .3 .11)
Ido
Relative light intensity
w j.t at 30 °C.
Simulated light intensity variation of LED with temperature is shown in Figure-2.3.7
Fig.-2.3.7: Calculated light intensity variation with temperature
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for Tc = -0.01/ °C. Simulated result shows that, with fixed current bias, light intensity
will vary by 104 % with respect to intensity at +30 °C over the temperature range of -20
to +80 °C. Whereas under optimum load bias condition, this peak to peak variation
reduces to only 10%.
2.3.3.I.2.
Theoretical Accuracy of Optimum Bias Load Line
Technique Over Temperature
Now we will investigate the accuracy of the optimum bias load line technique over
the operating range of temperature. Temperature coefficient of output light intensity at
any type of bias condition, say Tc /°C, of light emitting diode can be written from
equation (2.3.2) as:
T_ + P d ld
r.c =~
A
=
l d dT
377
(2.3.12)
When diode is biased by a voltage source of value V and series resistance R, as shown
in Figure-2.3.6, we can write from this circuit.
/, =
V-V.4
R + R„
(2.3.13)
Here, V^is the voltage drop across the junction of the diode.
Combining equation (2.3.12) and (2.3.13), we can write:
T =T
P dv<u
V -V dj dT
(2.3.14)
Differentiating expression of Vdd from equation (2.3.5) with respect to temperature T
1
1
"S3
J*3
1
£
we can write:
dV<u _ TIc.C
dT
p
L
J]kT r
i+
q
np + 7 W I
IT J qTc CJ
(2.3.15)
C
Putting this expression to equation (2.3.14) we get the temperature coefficient of
brightness variation under simple bias condition of voltage source V and series resistance
R as:
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y .y
rc = tc
+ p(Eb ~ VJ + T}M(l + np^
*
TTC
q \
TTc j
TjkT
y - V Jj +
(2.3.16)
When, diodes are biased by constant current source, V-» °°, and T0 = TJ °C
Now, when diodes are biased by proposed optimum bias load line condition, then
putting V = Vqpt in equation (2.3.16) from (2.3.10), temperature coefficient will be:
K -v jK -v j-
pEb
P f ^djo
t
r ,
iE b -~Vdjo )
TT
)
(2.3.17)
T]k
- ( T ' - T h ’t f 9
<Pc
This expression shows that at T = T0, temperature coefficient of brightness becomes
zero. The temperature coefficient at other temperatures for LEDs can be determined from
this expression putting typical parameter values of the diodes.
2 .3 .3 .I.3 .
E x p e r im e n t A n d T e s t R e s u lts
The 1N6092 GaAsP on GaP red LED ( a MIL grade device for space applications)
has been tested over the temperature range of -2 0 to +80 °C to confirm the theoretical
prediction of temperature compensated light intensity by proper selection of load line.
Figure-2.3.8 shows the measured equiintensity curves of LED 1N6092 for three
different constant light intensity levels, corresponds to 16, 20 and 24 mA current at
temperature of +30 °C. Load line corresponds to these three intensity levels are also
drawn. All these load lines intersect the voltage axis nearly at 1.90 Volts. Putting £& =
2.1 Volts, n = 2, p = 1, Tc = -0.0085 / °C and Vdj0 = 1.76 Volts in equation (4.10)
optimum voltage Vopt = 1-88 Volts; which agrees with the value obtained from graphical
method (nearly 1.9 Volts in Figure-2.3.8).
Figure-2.3.9 shows the measured light intensity variation of diode 1N6092. It shows
that when diode is biased by a conventional fixed current source, peak to peak light
intensity variation is 99% with respect to at temperature +30 °C. But when this diode
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biased through a optimum voltage source of Vopt = 1-9 Volts, then peak to peak light
intensity variation decreases to only 6 % over the temperature range o f-20 to +80 °C.
Fig.-2.3.8: Measured equiintensity curves of the diode 1N6092
over the temperature of -20 to +80 °C.
Temperature °C
Fig.-2.3.9: Measured light intensity variation with temperature
of the diode 1N6092
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2 .3 3 .1 .4 .
LED Driver Circuit fo r S everal LEDs Connected in Series
a n d P aralle l Combination
To achieve sufficient brightness for many industries applications including
automobiles, traffic signal lighting and artificial lighting similar to natural daylight, it is
required to combine several LEDs in series and parallel combination. Such a
combination is shown in Figure-2.3.10. Here m numbers of LED connected in series and
n number of such series combined LEDs are connected in parallel. Driver circuit
contains a voltage source of open circuit voltage value V optm and individual series path
contains separate series resistors of value 1?;, R2
To achieve temperature invariant brightness of these combination of LEDs the
optimum voltage source V optm and series resistors are determined by:
V0PTM = m x V OPT
K = (V o P T M -V j/ldn
(2.3.18)
(2.3.19)
Where, Vqpt is given by (2.3.10), /„ is the cunent through n* series path mid Vn is the
total voltage drop across all the diodes in n* path.
V optm
Flg.-2.3.10: Series and parallel combination of several LEDs to
achieve sufficient brightness
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2 .3 .3 .1 .5 .
Effect o f Supply Voltage Variation
The proposed simple bias circuit, as shown in Figure-2.3.6, for temperature invariant
brightness, biases LEDs by a optimum voltage source of value given by equation
(2.3.10). The current is controlled by series resistor R given by equation (2.3.11). The
circuit provides more current to the diode with increase of temperature due to the
decrease of diode voltage with increase of temperature and maintains nearly temperature
invariant brightness of the LEDs.
In previous section, we have seen that, optimum voltage Vopt is slightly higher than
the voltage drop across the diode. Thus, series resistor value is small which leads to
efficient bias circuit but more sensitive to the supply voltage variation due to
temperature, line regulations or by any other means. From (2.3.2) and (2.3.13) we can
write the brightness sensitivity on the supply voltage variation as:
dL
L
—
=
dV
FV- V,
P
----------------------
(2.3.20)
Denominator of this equation is very small, thus very stable supply source should be
used or supply voltage should be derived from stable voltage reference source, e.g. band
gap reference may be used.
2.3.4.
CONCLUSION
This chapter discusses the brightness variation of LEDs and discusses different types
of compensation techniques. Here a simple LED driver circuit proposed based on the
proper selection of bias load line which will provide nearly temperature invariant output
light intensity. Mathematical expressions are given to derive the component values of the
driver circuit. Experimental result shows good agreement with theoretical prediction.
The optimum voltage (Vopt) is determined by the intrinsic junction property (£*, VdJO)
and temperature coefficient (Tc) of the diode. Thus, it is expected, the presented
temperature compensation scheme will be valid for long term operation of LEDs where
gradual decrease of external quantum efficiency and increase of series resistance (Rs)
takes place. This circuit uses no separate temperature sensor or compensating
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mechanism, but responds directly to the junction temperature of the diodes. This
prevents any error caused by temperature gradient, or by self-heating due to power
dissipation in the diode.
2 .4
CONCLUSION
The chapter demonstrates the proposed novel “optimum bias load line technology”
for diode based microwave circuits to provide temperature invariant RF performance.
Different diode based circuits are realized operating at different frequency bands and
demonstrate the experimental results under optimum bias load line condition. This
chapter also demonstrate that same optimum bias load line technique is also applicable to
temperature compensate the output light intensity of Light Emitting Diode. The proposed
temperature compensation technique does not use any temperature sensor or separate
compensation mechanism but responds directly to the junction temperature of the diodes.
Thus there will be no compensation error caused by temperature gradient, or by self­
heating due to power dissipation in the diode. It is expected that microwave engineers
will be encouraged to use diode based RF circuits more and more to provide temperature
invariant performance.
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C
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3.0
GaAs FET AND HEMT
The gallium arsenide (GaAs) metal semiconductor field effect transistor (MESFET),
hetero-structure FETs (HFETs) and pseudo-morphic high electron mobility transistors
(pHEMTs), play an important role in microwave applications. These devices are
becoming more prevalent as the demand grows for high-speed data transfer and internet
access. As a result, users are demanding smarter circuits with temperature insensitive
performance for the present systems.
In recent years, the performance of these GaAs devices has been improved
significantly and there are significant works has been reported related to temperature
behavior of these devices including temperature dependence of MESFET and HEMT
equivalent-circuit parameters(ECP’s) to model circuit performance over possible
operating temperature range [52], [54]. Various methods to achieve temperature
compensated RF performance of the transistor-based circuits are reported. However,
design details and optimization procedure of the compensation circuits are not available
in the open literatures.
Linear gain, saturated output power level and efficiency of GaAs-based FETs depend
strongly on temperature. At fixed drain current and voltage bias, transconductance as
well as FET unity current gain cut-off frequency if,) decreases with increasing
temperature. Therefore, in case of fixed drain current bias, small signal gain, saturated
output power level and efficiency of amplifier based on MESFET and HEMT decreases
with increasing temperature. It has been known from the equivalent circuit model and
verified by the experiments on various amplifier circuits that the linear gain of GaAsbased FETs and HEMTs decreases approximately by 0.015 dB/°C for each stage as the
channel temperature increases. Thus, FET gain tends to decrease greatly with
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temperature, even for FETs biased at a constant current bias. Such a decrease will
significantly degrade the system performance.
To achieve temperature invariant small signal gain and saturated output power of
MESFET and HEMT based amplifier, it is required to increase drain current or drain
voltage with increase of temperature. There are various digital and analog temperature
compensation schemes for FET based amplifiers. Generally, temperature sensitive
voltage controlled variable gain block used in the line up of the FET based amplifier to
compensate the overall small signal gain variation of the amplifier, whereas, temperature
compensation mechanism used to increase the drain voltage or drain current or both of
the final power device, to achieve temperature invariant saturated output power of the
amplifier.
3 .1
T E M P E R A T U R E B E H A V IO R __________________________
High channel temperature has well known detrimental effects on the performance of
GaAs MESFETs and HEMTs. Degraded transconductance and distortion of the DC I-V
characteristics represent typical consequence of substantial self-heating of the device.
Advances in thermal on wafer computerized automatic probing systems have drastically
reduced the complexity of deriving temperature dependent DC and S-parameter data for
devices, integrated circuits and MMICs. These parameter-extracting systems can
effectively measure devices such as FETs and pseudo-morphic high-electron-mobility
transistors (PHEMTs) for development of temperature-dependent equivalent-circuit
parameters (ECPs).
Over the temperature range of -50 °C to 125 °C, the most of the equivalent circuit
parameters (ECPs) depends linearly on temperature and fitted to an equation such as:
P(T) = P(T0)[1+B(T-T„)]
(3.1)
where B is the temperature coefficient (TC) that has units per degree, Ta is the reference
temperature which is usually 0 °C, and Pip) is the value of the parameters at the
reference temperature.
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The main factors responsible for the change in device performance with temperature
are:
1)
The temperature coefficient of the electron velocity, which is given by:
vsat(T) = vUTo)U+BvUT-T0)]
(3.2)
where Bvsmis between -1 and -2.5x10-3 /°C.
2)
The temperature dependency of electron mobility factor, which is given by [55]:
A t )= m{t0
(3.3)
Where s is the scaling factor, depends on the doping concentration, varying from 0 to 2.
3)
H ie changes in the Schottky barrier height and surface potential with temperature
V*(r)=v* (rj + m[Egap(:t ) - e „
(3.4)
(t 0)]
Where m is between 0 and 1. and band gap potential is given by:
f/r2
=
=
<3-5>
Where or and /? are constants. In GaAs the bandgap at room temperature is 1.42 eV and at
OK it becomes 1.52 eV.
4)
The temperature coefficient for the dielectric constant (X = £) and specific contact
resistance (X = pc)
x(r)=x(rJ[i+Bx(r-rJ ]
(3.6)
The major physical factors driving MESFET and HEMT device characteristics are
the temperature dependence of the electron saturation velocity, the electron mobility, and
the pinch-off voltage. The pinch-off voltage is influenced by the Shottky barrier height
shift with temperature, kT spreading in the electron profile and possibly piezo-electric
effects.
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Due to the temperature dependency of the above physical parameters, the
performance of the solid-state devices changes with temperature. The performance of the
devices and change of the parameters are modeled by the temperature dependent
equivalent circuit parameters (ECP). The temperature dependent DC performance as well
as RF performance is modeled by the equivalent circuit parameters which will be
discussed in the next sections.
3.1,1
TEMPERATURE EFFECTS IN DC CHARACTERISTICS
The major physical factors driving MESFET device characteristics are the
temperature dependence of the electron saturated velocity, the electron mobility, and the
pinchoff voltage. The pinchoff voltage is influence by the Schottky barrier height shift
with temperature and kT spreading in the electron profile [52].
Luca Selmi and Bruno Ricco' had studied thoroughly [53] the temperature effects in
the DC 1-V characteristics of GaAs MESFETs. They have done pulsed I-V measurement
at different temperatures keeping the device in a temperature controlled chamber. They
have shown the two main temperature effects as shown in Figure-3.1, these are:
Pulsed measurement at low temperature
Fig.-3.1: Typical 1-V characteristic of MESFET with change of temperature
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1) At high gate bias condition (high Vos,) the drain current decreases as the temperature
increases, this is mainly due to the reduction of mobility and saturation velocity with
the increase of temperature.
2) As Vos approaches the transistor threshold voltage (VV), the drain current increases
with the increase of temperature. This is mainly due to the decrease of threshold
voltage with the increase of temperature.
Thus, it can be concluded that, if the device is operated much above threshold in the
saturated current regime, negative temperature coefficients are observed for /* and gm.
Whereas, if the device is operated close to threshold, both the parameters measured at
fixed gate voltages give positive temperature coefficients.
3. 1.2
TEMPERATURE EFFECTS IN R F CHARACTERISTICS
The basic RF equivalent circuit of MESFET and HEMT is shown in Figure-3.2,
which is a standard one and is adequate for most of the applications. Where Rs, Ra and Rg
are the source, drain and gate parasitic resistances respectively. Ls, La and Lg are the
C pf
S
Fig.-3.2: The equivalent circuit model of MESFET and HEMT
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corresponding parasitic inductances. These parasitic resistances and inductances are
independent of bias. Similarly, the parasitic capacitances Cpd, Cpg and Cpf are also
independent of bias.
S21 changes the most with temperature, which affects the gain of the amplifier.
Measured gains obtained from the S parameter fall off by -0.0075dB/°C per amplifying
stage. Simulation shows [52] that gain of the amplifier drops with the temperature at the
rate of -0.0055dB/°C, considering only the temperature effect of three numbers of ECPs
namely trans-conductance (gm), gate to source capacitance (Cgs) and drain to source
resistance (Rds)- Thus, these three ECPs are the key parameters affecting MESFET
device and amplifier performance over temperature.
The temperature coefficient of trans-conductance of the MESFETs highly depends
upon the gate voltage. At high gate bias voltage, TC of gm is negative and dominated by
the saturated electron velocity and mobility of the carrier and high current condition also
determined by the self-heating of the device. At low gate voltage near pinch-off
condition, TC of the trans-conductance becomes positive.
Overall temperature coefficient (TC) of gate capacitance Cgs influenced by the three
temperature coefficients [3] a) TC of Schottky barrier height b) TC of dielectric constant
and c) kT spreading. Overall temperature coefficient of the gate capacitance is positive
but highly depends upon the gate bias voltage.
Overall temperature coefficient (TC) of source-drain resistance
determined by the
temperature coefficient of the a) TC of dielectric constant b) TC of Schottky barrier
height c) TC of mobility and d) TC of specific contact resistance. Rds, is highly depends
upon the gate bias voltage. At high gate bias voltages, the TC of Rds is positive and is
dominated by the temperature dependence of the mobility and Schottky barrier height. In
the pinch-off region, because the pinch-off voltage becomes more negative as the
temperature increases, the currents at fixed gate voltages increase with temperature,
therefore the TCs of the resistance becomes negative.
Although the /, values are not ECPs, it is very useful in characterizing FETs and
computing noise figure and gains. The value off, is given by:
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8m
/, = 27cCg
g
(3.7)
where Cgg is the sum of the capacitances Cgs and Cgd. Thus, at normal operating
condition of the MESFET at high gate voltage bias, /, has negative temperature
coefficient.
Thus in brief, if the device is operated much above threshold in the saturated current
condition, negative TCs are observed for Ids, gm and f
and positive temperature
coefficient for Cgs, Cgd and R&. If the device is operated close to threshold, almost all
parameters measured at fixed gate voltages give positive temperature coefficient.
3. 1.3
TEMPERATURE BEHAVIOR O F HEMTs
For very high frequency, low-noise applications, pseudomorphic HEMT (pHEMT)
and InP lattice matched HEMTs (LMHEMTs) with InGaAs channels have demonstrated
excellent performance and been widely deployed in circuits operating at all the
frequencies starting from microwave to mm wave frequencies. And also these devices
are popularly used for low noise amplifiers as well as high power amplifiers over the
entire microwave and mm wave frequency range. InP lattice matched HEMTs advantage
has higher mobility and electron saturation velocities in InGaAs. The standard equivalent
circuit model of HEMTs are same as of MESFET as shown in Figure-3.2 with different
values of the ECPs.
As in MESFETs, the changes with temperature in the ECPs and DC parameters in
HEMTs are also influenced by the temperature dependence of the Schottky barrier
height, the electron mobility, the electron saturation velocity, the dielectric constant and
the specific contact resistance. The temperature coefficient of the ECP values are
qualitatively similar to MESFET TCs. Temperature coefficient of/,, gm and Ids values are
negative at high gate voltages and positive near threshold, and the TCs of Cgs and Rds are
positive.
A wide variety of nonlinear temperature dependent behavior like pinchoff-voltage
steps, kink effects and I-V collapse have also been observed [52] in conventional
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HEMTs and pHEMTs which are associated with the Si-doped AlGaAs layers. These non
linear phenomenon are thoroughly discussed by R Anholt in [52].
The step changes in the pinchoff voltage and I-V collapse, both are the low
temperature phenomenon.
The other effect, kinks in the I-V curve is also a low
temperature phenomenon and observed in the high drain voltage bias condition. Due to
the presence of these non-linear temperature phenomenon in the conventional HEMTs, it
is very difficult to design high-frequency components meeting constant gain versus
temperature specifications. However, all these nonlinear temperature phenomenon are
not observed in InP LMMEMTs.
3 .2
TEMPERATURE COM PENSATION
As discussed in the previous sections that standard RF equivalent circuit of MESFET
and HEMTs are described by nearly thirteen numbers of equivalent circuit parameters
and all the parameters are changes more or less with the change of temperature. Among
all the ECPs, transconductance (gm), gate to source capacitance (CgJ) and drain to source
resistance (/?«&) are mainly contribute to the gain variation of the amplifiers based on the
MESFET and HEMT devices. Gain variation is the most common temperature
dependent factor affecting circuit design. For staged amplifier configurations,
temperature effects ate critical since gain variation is multiplicative as:
AGAIN T = y.AT,n
(3.8)
Where / i s the decrease of gain per degree Celsius per amplifier stage, AT is the
temperature difference (T-T0) and n is the number of amplifier stages.
One of the most important parameters determining gain of microwave and mm wave
frequency amplifiers is fi, which decreases steeply with temperature, no matter what
biasing scheme is used. Thus, with the increase of temperature, gain of the FET based
amplifiers are decreases whether the devices are biased with constant gate voltage or
constant drain current bias conditions. The dominant temperature effects are the decrease
in the magnitude of S21 and S22.
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The paper [60] proposed an active bias scheme for FET amplifier, which maintains
fixed drain current irrespective of the change of FETs internal parameters due to change
of temperature or any other means. The scheme uses drain resistor Rd with a PNP bipolar
transistor and supply voltage VDd to achieve constant drain current Id, and drain voltage
V d- In this scheme, total drain current of the FET passes through the resistor Rd Thus,
constant drain current achieved with the penalty of extra power consumption of (VDdV d) x ID, with total power consumption of V dd
x
Id, which decrease the power added
efficiency (PAE) of the amplifier. In practice, to achieve maximum efficiency of the
power amplifier, drain bias of the FET should be given directly by a voltage source, to
minimize the total power consumption to Y d x Id.
The paper [60], also demonstrated that the small-signal gain drift with temperature is
compensated by keeping fixed drain current. This contradicts the investigation and
conclusion done by Anholt and Swirhun [54] (IEEE Trans, on Electron Devices, -39,
(Sept 1992). 2029-2036). Anholt and Swirhun have thoroughly investigated [52], [54]
the temperature dependency of the MESFET and HEMTs and had this to say: “If F E T s
are biased at a constant current or a constant fraction of /* s (which itself changes with
temperature) some what different ECP coefficients can be obtained,..........One of the
most important parameters determining gain of high frequency amplifiers is f , which
decreases steeply with temperature no matter what biasing scheme is used.” They have
concluded that [54] “.... FET gain values to decrease greatly with temperature, even for
FET,s biased at a constant current”. Thus, it is clear that, keeping constant drain current
of the FETs over the change of temperature, it is not possible to achieve temperature
invariant small-signal gain of the FET amplifiers.
Thus, it is not true as assumed in [60] that “The small-signal gain parameter is known
to decrease with increasing temperature due to a drop in the drain current...”. Actually,
at fixed drain current bias also, transconductance as well as FET unity current gain cut­
off frequency (/>) decreases with increasing temperature. Therefore, in case of fixed drain
current bias also, gain of the amplifier decreases with increasing temperature.
We have designed and developed various amplifiers, based on MESFET and pHEMT
devices and characterize over the operating range of temperature. MIC assembly and
photograph of a C-band 3-satge amplifier, based on MESFET of type NE13783 of NEC
is shown in Figure -3.3 [57]. The amplifier is designed based on the distributed
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matching network technology. The devices at this frequency are conditionally stable
device and stability of the amplifier achieved by providing selective resistive load at the
matching network. [57]. The circuit is realized using inter-digital capacitor as the DC
blocking capacitor instead of conventionally used chip capacitor, which enhances the
performance of the circuit and improves the repeatability of performance.
Photograph of a Ku-band amplifier based on MESFET NE-67383 is shown in
Figure-3.4. MIC assembly drawing and photograph of another 3-stage amplifier circuit
based on pHEMT of type CFY67-08 of Symens, at operating frequency of Ku-band is
also shown in Figure-3.5 [ 83]. This is a very compact circuit using discrete devices.
More details of the circuit will be given at the next chapter. Other amplifier circuits
based on medium power MESFETs MGF2407, MGF2430 of Mitsubishi also designed
and tested over temperature.
Fig.-3.3: MIC assembly drawing and photograph of a 3-stage C-band amplifier of
MESFET (NE13783)
Fig.-3.5: Photograph of a 3-stage Ku-band amplifier of MESFET (NE 67383)
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era , (jujarat University, India.
Fig.-3.4: MIC assembly and Photograph of a 3-stage Ku-band amplifier of
pHEMT (CFY67-08)
Our Experiments on various MESFETs (NE-13783, NE-67383, MGF2407,
MGF2430) and HEMT (CFY67-08) agrees with the conclusion of Anholt and Swirhun.
Experiment shows that constant drain current bias decrease the small-signal-gain drift to
nearly 0.01 ldB/°C per stage, compared to 0.015 dB/°C, in case of conventional resistor
network biasing. It has been also seen that to achieve constant small-signal gain of the
FET amplifiers, it is required to increase drain current or drain voltage with the increase
of temperature.
Thus, it can be concluded that it is not true that by keeping drain current constant, by
the suggested active biasing scheme, it is possible to improve the PAE and temperature
compensation of the small-signal gain drift of the FET amplifier as demonstrated in [60].
Huang, et. al. proposed and demonstrated [58] the use of the GaAs feedback resistor
in the amplifier stage to reduce the gain variation of the amplifier with temperature. This
temperature-compensation design approach is similar to feedback-controlled technology.
They have used GaAs thin film resistor as the feedback resistor since its temperature
coefficient is much larger than the conventionally used tantalum Nitride (TaN) resistor.
Here temperature compensation of a four-stage pHEMT based amplifier has been
demonstrated. GaAs Feedback resistor is used only at the first stage amplifier for
temperature compensation of the total amplifier. The resistivity of the GaAs feedback
resistor increases as the operation temperature increases. This will reduce the feedback
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magnitude and increases the small-signal gain. Similarly, the feedback magnitude
increases as the operation temperature decreases, so as to result in a smaller signal gain.
The active biasing schemes increases the drain current and/or drain voltage of the
MESFET and HEMTS to achieve the temperature invariant gain, output power and other
RF parameters of the circuits. Sometimes amplifiers using active biasing scheme
sufferers from instability and reduced gain flatness over the frequency range. We have
proposed and demonstrated the use of separate temperature controlled gain/attenuation
block to compensate the gain variation of the multistage power amplifier and channel
amplifier. The technique is very much simple and does not affect the stability of the
amplifier. The circuit optimization is also very much simple. The details of the scheme
will be discussed in the next chapter.
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4.0
S O M E M IC R O W A V E SUBSYSTEM S
Compactness, reliability, efficiency and ability to work at varying temperature
environment are important parameters of the subsystems for air borne and space based
applications. Compact systems are designed by accommodating more and more
functional blocks in a single module, as an example, is the active integrated antenna [72],
[73]. In this case, active antenna is designed integrating active circuits such as amplifiers,
mixer and filters within the same module. Compactness of the circuit not only reduces
the size and weight of the system but also improves the system efficiency eliminating
interconnection losses and also improves reliability due to the elimination of various
interface cables and connections. To operate the complete system in the varying
temperature environment, keeping in mind the above important parameters, design
aspect of the different temperature compensation mechanism has to be explored. The
proper temperature compensation mechanism reduces the complex heat sink requirement
to maintain stringent temperature limits of the circuits and systems. The thesis work
discussed the temperature behavior and compensation mechanism of four spacecraft
subsystems, first is diode based vector modulator for beamforming network [70], [71]
second is diode-based linearizer [39], [40], another is, over-drive level controlled
microwave solid-state high power amplifier [80] and another is a Ku-band solid-state
channel amplifier with automatic level control system [83]. These subsystems contain
GaAs FET, HEMTs, PIN diode and Schottky barrier diode. Temperature compensation
schemes for these subsystems are proposed and designed. Mathematical and practical
design procedures are also developed and presented to determine the circuit component
values.
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4.1
A T e m p e ra tu re C o m p e n s a te d D io d e - B a s e d
V e c to r M o d u la to r fo r B e a m F o rm in g N e tw o rk
Phased-array antennas are gaining significance for on-board satellite applications
because of their ability to form multiple beams and to provide power sharing among
beams [62] - [71]. These phased array antennas offer improved operational flexibility by
providing independent beam reconfigurability and steerability, resulting in more efficient
use of satellite-power resources. Moreover, with the generation of narrow beams, higher
effective isotropic radiated power (EERP) can be achieved,
enabling direct
communication with small earth stations.
Important consideration in selecting the beamforming structure is the kind of array
and the beamforming function implementation. Orthogonal beamforming based on
crossobar arangements is well-known technique particularly efficient when the matrix
elements are designed to control the relative phase and amplitude by command [69].
These kinds of matrix elements require circuits providing both amplitude and phase
control. These control circuit must be of small size, lightweight, of low power
consumption. Also amplitude and phase characteristic of the network must be
temperature invariant to operate it wide temperature varying space environment.
In general beamforming network comprising different individual circuits to perform
the amplitude and phase-control tasks: variable phase shifters, variable gain/attenuation
stages and switches. This approach results in higher insertion losses and cumulative
errors in combined results, larger size, and more complicated housing than using a single
circuit.
In this section, an single circuit approach vector modulator using PIN diodes with
novel optimum load-line biasing scheme for temperature invariant performance is
presented [70], [71]. The circuit is realized in a balanced configuration with very good
input and output return loss over the operating frequency range of 2.5 to 2.7 GHz for the
entire amplitude (up to -22 dB) and phase variation (360°). Other two vector modulators
are also realized at satellites C-band and Ku-band down link frequency.
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4 .1 .1
D e s ig n A p p r o a c h
A schematic block diagram of the circuit is shown in Figure.-4.L1 [70]. Here input
signal is divided into two parts with equal amplitude but with phase difference of 90°
with the help of a 3-dB 90° power divider. Then both the signals are selectively
attenuated and phase shifted by an amount of 0° or 180° using voltage/current controled
variable attenuator and finally combined by a in-phase power combiner. With this design
approach, all possible amplitide and phase combination can be achieved as shown in the
phaser diagram as shown in the figure.
Fig.-4.L1: Block schematic of the vector modulator
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A ll the circuits are realized on 25mil alumina (8r = 9.9) substrate. The circuit is
simulated and analyzed using Series-IV software o f HP-BEsof.
4.1.2
D e t a ils o f C ir c u it D e s ig n
Input 3-dB 90° coupler is implemented by Lange coupler to achieve good amplitude
and phase performance over broad frequency band. Output 3-dB in-phase power
combiner is realized using Wilkinson type power divider.
The most important circuit module is the variable attenuator with all possible
attenuation range and with the provision o f 180° phase switching. The module is realized
using reflection type PIN diode attenuator. Schematic o f the attenuator circuit is shown
in Figure-4.1.2. 3-dB 90° Lange coupler is used to realize variable attenuator. Pair o f
Lange
Fig.-4.1.2: Schematic circuit o f variable attenuator with phase shift
PIN diodes are connected to the direct and coupled port o f the Lange coupler and
isolated port used as output port o f the variable attenuator. The basic working principle
o f the circuit is that S n o f the diode terminated port will be the S2 t o f the circuit. The
incomming signal at the input port is split equally between the direct and coupled ports,
with a 90° phase difference. A further 3-dB split and 90° phase difference is introduced
when the waves are reflected back through the coupler. When the two reflected signals
are superimposed at the input port they are in antiphase and thus cancel. At the isolated
port o f the coupler the two reflected signals are in-phase and thus reinforce each other to
form the output signal. Thus for Lange coupler o f characteristic impedance R0 (= 50Q),
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S21 of the attenuator circuit, which is the reflection coefficient (p) of the diode terminal,
is given by:
Where, Zd is diode impedance. Hence a pure variable resistance of diode would lead
to a variable S21 amplitude, achieving both positive and negative signs. Therefore, S21 = 0
when the impedance of the diode will be 50 SL Considering diode impedance as pure
Z d <50Q
50Q
Fig.-4.1.3: S21 variation on diode resistance
variable resistor, the amplitude and phase of S21 of the variable attenuator circuit is
shown in Figure-4.1.3.
Equivalent circuit of a forward biased PIN diode can be represented by a voltage
variable resistor Rd with parallel combination of a capacitor Q a s shown in Figure-4.1.4
[27]. Thus considering Lange coupler as a ideal 3-dB 90° coupler, S21 of the circuit will
be given by (4.1.2):
Fig.-4.1.4: RE equivalent circuit of forward biased PIN diode
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Thus amplitude and phase of the variable attenuator circuit is given by:
K
(Rd - R o f + (0)CdRdRo )2
(A lt)
4 S 2 i (*„) — Tan
(4.1.3)
(Rd + R 0 f + (a)CdRdR0 f
2 a>CdR / R 0
R / - R 20 - ( a C . R j R o f
(4.1.4)
The circuit is realized by using PEN diode of type HPND-4005 of HP, which is beam
/S2J
(dB)
ZS2i (degree)
lead type for high frequency operation. Capacitance value of the diode is 0.017 pF and
-40
-100
10
100
1000
10000
Diode Resistance (£2)
Fig.-4.L5: Simulated S2 1 plot of the variable attenuator
resistance valua varies from 2kft to 5ft with diode carrent of 0 to 10 mA. The simulated
amplitude and phase variation of the attenuator circuit with the change diode resistor is
shown in Figure-4.1.5 [70]. Plot shows that PIN diode based attenuator provides very
high level of attenuation up to 35 dB and also provides 180° phase shift, as required for
vector modulator, when diode resistance crosses 50 ft. Slight phase deviation near diode
resistance of 50ft is due to paracitic capacitance of of value 0.017 pF. The phase
deviation can be eliminated by implementing inductance in parallel with the diode. The
value of the inductance will be such that parallel resonance frequency will be the
operating frequency of the vector modulator.
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4.1.3
Realization o f Vector Modulator
Schematic circuit diagram of the vector modulator circuit is shown in Figure-4.1.6.
Here input signal at the input port is split equally between the direct and coupled ports,
Wilkinson
Power
combiner
IN
O
OUT
Fig.-4.1.6: Schematic circuit diagram of the proposed vector modulator
with a 90° phase difference. These two divided signals are selectively attenuated and
phase shifted by the variable phase shifter and finally added by the Wilkinson power
divider.
The amplitude IS2 /I and and phase ZS 21 of the vector modulator is given by (4.1.4)
and (4.1.5) respectively.
(4.1.4)
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Where, pi and p 2 are the insertion loss of the attenuator-1 and attenuator-2
respectively and given by (4.1.2). And, 9 is the phase angle between the two signals
combined at the output port of the vector modulator and given by (4.1.6)
9 = <p2-<pl -9 0 °
Where (pi and
(4.1.6)
ph are the insertion phase of the attenuator-1 and attenuator-2
respectively and given by (4.1.3).
In ideal case, minimum insertion loss is 3dB is due to the 90° phase difference
between the two signal combined at ourput. Simulated insertion loss is 4.5 dB. Simulated
S21 of the vector modulator is shown in Figure-4.1.7. It shows that by varying diode
resistances it is possible to achieve all phase and amplitudes.
Three different circuits are realized at three different frequency band of centre
frequencies 2.6 GHz, 3.9 GHz and 11.6 GHz. The photograph of the circuits are shown
in Figure-4.1.8. All the circuits are realized with MIC technology. RF circuits are
realized on 25mil alumina (£r = 9.9) subatrate. Beam lead PIN diode of type HPND4005
of HP is used for variable attenuator. The test results of the S-band vector modulator is
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shown in Figure-4.1.9. The plot ahows some sample of the readings. Here, PIN diodes
are biased by constant voltage source by varying series resistor. Experimental achieved
minimum insertion loss is 4.5dB, when there is no bias to the attenuators. 7.5 dB
attenuation is achieved when there is no bias to one attenuator and maximum attenuation
is given to another attenuation. Constellation of vectors, of different amplitude and
phase, as shown in the figure is achieved by giving different sets of bias to the two
variable attenuators. This plot shows that all possible phase over 360° can be achieved
with different resultant amplitude.
Ku-Band
S-Band
Fig.-4.1.8: Photograph of vector modulators
Fig.-4.1.9: Measured polar S21 plot of S-band vector modulator
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4 .1.4
Temperature Behavior and Compensation o f Vector
Modulator
Variable attenuator of the vector modulator is realized using PIN diodes. RF
resistance of PIN diode determines the amplitude and phase of the vector modulator. As
discussed in the previous chapter, RF resistance of PIN diode is function of temperature.
Thus, amplitude and phase of the vector modulator will also vary when the vector
modulator will operate under varying temperature environment of spacecraft. Amount of
variation depend upon the different amplitude and phase setting of the vector modulator.
Worst-case amplitude and phase variation is nearly ±3dB and ±7.5° over the temperature
range of -10 to +60 °C when the diodes are biased by conventional constant current bias
condition. Constant current source bias not only vary the amplitude and phase
performance of the vector modulator, but also it requires complicated circuit to realize
variable constant current source to achieve constellation of amplitude and phase.
As discussed in the previous chapter, proposed optimum bias load line technology
can provide temperature invariant attenuation of the PIN diode based attenuator. Same
technology is applied to achieve temperature invariant amplitude and phase performance
of the vector modulator.
Voltage
RF
OUT
Fig.-4.1.10:
2Vt® 'Thesis
Schematic circuit of the temperature
compensated analog vector modulator
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Figure-4.1.10 shows circuits diagram of an analog vector modulator with
temperature compensation biasing scheme [24]. Here, both the attenuators are biased
from the same optimum voltage source (Vopt) derived from single voltage source with
the help of potential divider. The optimum voltage Vopt is derived in previous chapter,
which depends upon the PIN diode parameters and recalled here as:
(4.1.7)
For PIN diode of type HPND4005,
V opt
= 1.2 volts. Variable resistors Ri and Rq are
used to vary the currents of the attenuators continuously to vary amplitude and phase of
the vector modulator continuously. With this optimum voltage source bias, amplitude
and phase variation reduces to ±0.5dB and ±1.4° respectively over the temperature range
of -10 to +60 °C. Thus, the proposed temperature compensated circuit provides stable
constellation of amplitude and phase varying vector over very wide range of varying
temperature environment.
For digital beamforming network, amplitude and phase of the vector modulator is to
be changed digitally. Figure-4.1.11 shows the schematic circuit diagram of a digitally
controlled temperature compensated vector modulator. Here electronic analog switch
with bunch of selectable resistors are used to realize PIN diode based step attenuator.
Depending upon data to the analog switch, only one resistor at a time will be connected
to the input optimum voltage
V o pt -
The optimum voltage is determined by (4.1.7) by
putting diode parameter, also can be determined graphically by collecting V-I data
experimentally as discussed in the previous section. PROM is used to provide
predetermined data to the analog switches to select any combination of resistors from
both the switches, which will provide different combinations of current to the PIN diodes
and produce different combination of temperature invariant amplitude and phase of the
vector. Analog switch 74HC4051 is used to implement the digitally controlled vector
modulator.
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Analog \sw
\Switch
it
Vs
p
R
O
M
RF
OUT
Address
bus
Fig.-4.1.11: Schematic circuit of the temperature compensated
digitally controlled vector modulator
4 .1 .5
C o n c lu s io n
This section discussed a single circuit approach vector modulator using PIN diodes
with novel optimum load-line biasing scheme for temperature invariant performance.
Different circuits are realized in a balanced configuration with very good input and
output return loss over different frequency of operation at 2.6 GHz, 3.9 GHz and 11.6
GHz. Due to its good input, output return losses, the circuit can be easily integrated with
other circuits for beamforming network. The proposed circuit is very simple and
provides temperature compensated amplitude and phase performance.
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4 .2
Tem perature C om pensated Diode B ased
Linearizer
Today’s satellite communication transponder needs high power microwave
amplifiers such as TWTAs and SSPAs to meet high EIRP requirements. Power
amplifiers are require to operate at back off condition to meet required linearity during
multi-carrier operation and high data rate digitally modulated traffic, which degrade DC
to RF "efficiency of the system. Linearizing is the technique used to improve non-linear
performance of communication systems leads the high power amplifiers to operate at
near saturation or at less back off condition. Thus, trade off between efficiency and
linearity can be avoided by the use of linearizer.
Among the various types of linearizer [28] - [38], diode based pre-distortion
linearizers are suitable for space applications for its compactness and less power
consumption. There are numerous reported diode based pre-distortion linearizers [28] [36] with their own merits and demerits. Pre-distortion linearizer generates amplitude
and phase characteristic opposite to the characteristic of power amplifiers. Thus,
cascading linearizer with power amplifier, decreases overall amplitude and phase non­
linearity leads to overall improved linearity.
— *■
o
(_ j
O
l
ISD(mA)
Load line for series
diode linearizer
Load line for
parallel diode
linearize?''^
0.0
'°-5
VsD(Volt)
°-°
°-5
Fig.-4.2.h Measured I-V characteristics of Schottky diode in presence of RF power
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Measured I-V characteristic of a Schottky barrier diode (beam lead of type HSCH
5315) in presence of RF power is shown in Figure-4.2.1. Two types of load line, one for
nearly constant voltage bias condition and another for nearly constant current bias
condition are also shown in the plot. It is clear from the plot that RF resistance (dynamic
resistance) of diode decreases with increase of RF power level when the diodes are
biased with nearly constant voltage (small series resistance) bias condition. Whereas, RF
resistance (dynamic resistance) of diode increases with increase of RF power level when
diodes are biased with nearly constant current bias (high series resistance) condition.
Depending on the load line effects, some linearizers are parallel diode based, biased by
nearly constant current bias condition and some are series diode based, biased by nearly
constant voltage bias condition. Some linearizers [36], [39], [40] are also uses PIN diode
based attenuator with Schottky diode based distortion generator to generate adjustable
amplitude and phase non-linearity characteristic to linearize various power amplifiers
with different types of amplitude and phase non-linearity. Under conventional constant
voltage or constant current bias condition, diode based linearizer’s performance changes
with temperature since RF resistance of the diode is function of temperature as shown in
Fig.-6.2.2: Simulated RF resistance variation with temperature
Figure-4.2.2. It shows that, series diode based linearizer, with nearly constant voltage
bias condition will suffer severely with change of temperature compared to parallel diode
based linearizer with nearly constant current bias condition.
In conventional approach, digital or analog circuits with temperature sensors can be
used for temperature compensation of diode based linearizer, which lead to more part
count and complex circuit, resulting less reliability. In previous chapters, we have shown
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that by selecting proper bias load line it is possible to achieve temperature invariant RF
resistance of Schottky barrier diode and PIN diode.
This chapter discusses the design, development and test results of a novel diode
based linearizer based on Schottky barrier diode and PIN diode. Temperature
compensation of the proposed linearizer circuit, using novel diode biasing technique as
discussed in the previous chapters, is also discussed.
4.2.1
D e s ig n o f a N o v e l D io d e B a s e d L in e a r iz e r
Schematic circuit diagram of the proposed pre-distortion diode linearizer [39], [40] is
shown in Figure~4.2.3. To achieve different combination of magnitude and phase
characteristic, voltage variable PIN diode based (D1 and D2) attenuator circuit (labeled
as linear arm) utilized in combination with distortion generator circuit (labeled as non­
linear arm) using Schottky barrier diode (SD1 and SD2). These two arms are connected
with a 3-dB 90° input power divider and a 3-dB in-phase output power combiner.
Fig.-4.2.3: Schematic of the proposed diode based linearizer
Due to inherent property of 3-dB 90° hybrid at input and 3-dB in-phase output power
combiner, input and output return loss will be always good for all values of the diode
resistance as long as the diode pairs are matched. Thus, the proposed linearizer topology
eliminates the use of isolators at both input and output ports, as required for the
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previously reported many diode based linearizers [28] - [31]. Therefore, the proposed
diode based linearizer leads to very compact Iinearizer circuit with very good
input/output matching without using any isolators.
The equivalent circuits of a beam lead PIN diode and Schottky diode are consisting
of an equivalent resistance Rd with a parallel capacitance Q as shown in Figure-4.2.4.
Therefore, insertion amplitude p and phase (j>of the 3-dB 90° coupler with two numbers
of matched diode connected as shown in Figure-4.2.3, are given by equation (4.2.1) and
(4.2.2) respectively.
(Rd - R o y + (a>CdRdRo y
P =
*
(Rd + R 0 y + (mCdRdR0 ) \
0 = -Tan
2<oCdRd2R0
Rdz-Rl-{a>CdRdRoy
R,
(4.2.1)
(4.2.2)
Cd
Fig.-4.2.4: RF equivalent circuit of forward biased junction diode
Where p =p i and <]>- <f>ifor Rd = Rpa and Cd = Cpd in case of coupler with PIN diode
of linear arm.
And p = p 2 and $ = (jh. for Rd = RSd and Cd - Csd in case of coupler with Schottky
diode of non-linear arm.
The amplitude \S21\ and insertion phase ZS21 of the linearizer is given by (4.2.3) and
(4.2.4) respectively.
521
•PAD TAesu
;A
■*"f 2 A 1
j PiP-fi05®
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(4.2.3)
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106
rw. -i
pjSinO
Z S 2 1 = T a n 1------ --------/?, + p 2Cos 0
(4.2.4)
Where 0 is the phase angle between the two signals combined at the output port of
the linearizer and given by (4.1.5).
e=<t>r -<t>x-9 0 °
(4.2.5)
Figure-4.2.5 shows the plot of simulated amplitude and phase characteristic from
equation (4.2.3) and (4.2.4). The plot shows the amplitude and phase of the linearizer
Jo
fs
with respect to Schottky diodes resistance for different setting of PIN diodes RF
resistance at frequency of 11.55 GHz, taking Cpd = 0.01 pF and O = 0.15 pF. The plot
shows that, the linearizer shows positive amplitude and negative phase deviation when
diodes are biased by nearly constant current bias condition (i.e. RF resistance increases
with increase of RF power level) and initial bias of the diodes are corresponds to RF
resistance more than or equal to Ro- This amplitude and phase characteristics (increase of
amplitude and decrease of phase with increase of RF power level) is required to
compensate the amplitude and phase non-linearity of solid state power amplifier (SSPAs)
and traveling wave tube amplifiers (TWTAs). Other combinations of amplitude and
phase characteristics can be obtained by different biasing conditions of the PIN and
Schottky barrier diodes [39], [40].
A compact linearizer has been designed and developed [39] at the communication
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5
Fig.-4.2.6: MIC layout and assembly diagram of Ku-band lineanzer
satellite’s Ku-band down-link frequency of 11.4 to 11.7 GHz. MIC layout of the
complete linearizer circuit is realized very compactly of size 0.75 inch x 0.5 inch in a
single 25 mil thick alumina substrate of er = 9.9. The MIC assembly is shown in Figure4.2.6. A 3-dB Lange coupler with two numbers of diode connected at the direct port and
coupled port realizes linear and non-linear arms in a reflective type configuration.
Isolated port of the coupler used as the output port of the linear and non-linear arm. For
non-linear arm beam-lead Schottky barrier diode HSCH-5315 and for linear arm beamlead PIN diode HPND-4005 has been used. Input power divider and output power
combiner is realized by 3-dB Lange coupled and Willkinson power combiner
respectively. R1 and R2 are the biasing resistors for liner and non-linear arms
respectively. The complete circuit is analyzed using HPs Series-IV Harmonic- Balance
simulator.
Putting bias resistor R2 of the non-linear arm before the high capacitance value linefilter LF2, minimizes the memory effect of the linearizer. Therefore, change of voltage
and current of the bias resistor can follow the envelope of the multi-carrier signal. Thus,
linearizer can be considered as a quasi-memory-less system [76], [77] and linearizer can
be modeled only by their Pin/Pout characteristics (AM/AM conversion) and their input
power dependent phase shift characteristics (AM/PM conversion).The photograph of the
realized Ku-band linearizer is shown in Figure-4.2.7.
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Fig.-4.2.7: Photograph of Ku-band linearizer
4 .2 .2
T e s te d R e s u lts o f a K u -B a n d L in e a r iz e r
Tested amplitude and phase performance of the Ku-Band linearizer is shown in
Figure-4.2.8 at the frequency of 11.55 GHz. Here different combination of amplitude
(AM/AM) and phase (AM/PM) characteristics is obtained for different bias conditions of
the distortion generator and variable attenuator. Here Ug is the bias current of the
distortion generator at no RF condition and Ia„ is the bias current of the PIN diode
attenuator. This figure shows that different combination of amplitude expansion up to
7dB and phase change up to 44 degree is achievable. Plot (A) for Schottky diodes current
of 0.7 mA and PIN diodes current of 0.4 mA, generates gain expansion of 3 dB and
phase decrease of nearly 20 deg. That is required to linearize SSPAs non-linearity. Plot
(B) corresponds to, Schottky diodes current of 1.4 mA and PIN diodes current of 0.5
mA, generates gain expansion of 5 dB and phase decrease of nearly 35 deg. Whereas,
plot (C) corresponds to Schottky diodes current of 1.5 mA and PIN diodes current of 0.6
mA, generates gain expansion of nearly 6.5 dB and phase decrease of nearly 46 degree.
Both the combinations are suitable to linearize TWTAs to improve transmitter’s
linearity, which uses high power tube amplifier.
<
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Fig.-4.2.8: AM/AM & AM/PM characteristics of the linearizer for
different bias conditions
4 .2 .3
E v a lu a tio n o f a S -B a n d D io d e L in e a r iz e r
An S-band diode based linearizer also realized [40] based on the same principle of
operation as Ku-band linearizer discussed in the previous sections. The photograph of the
realized S-band linearizer circuit is shown in Figure-4.2.9. The S-band linearizer
designed at operating frequency of 2.55 to 2.63 GHz. 25 mil thick alumina (AI2O3)
substrate of £r - 9.9 is used to realize all the MIC circuits. Linear and non-linear arms (to
generate distortion) are realized by a 3-dB Lange coupler in a reflective type
configuration. For non-linear arm beam-lead medium barrier batch matched Schottky
barrier diode HSCH-5315 and for linear arm beam-lead PIN diode HPND-4005 has been
used. Input power divider and output power combiner is realized by 3-dB Lange coupled
and Willkinson power combiner respectively. All the individual circuits and integrated
complete linearizer circuit is analyzed using HPs series IV Libra simulator (harmonic
balance simulator).
VhT) ‘Thesis
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IN
OUT
Fig.-4.2.9: Photograph of S band linearizer
All the MIC cards are brazed on the gold plated kovar carrier plates to minimize
thermal stress of the alumina substrate and carrier plates are mounted in the aluminium
box of size nearly 2 inch x 2 inch.
HPs Vector Network Analyzer (VNA) HP8510A is used for amplitude and phase
characterization of the linearizer. The linearizer was evaluated with a 2-Watt S-band
SSPA. The improvement of gain compression and total phase shift is shown in Figure4.2.10. Gain compression decreases from 3.2 dB to 0.8 dB at nominal output power
Fig.-4.2.10: Amplitude and phase characteristic of SSPA with linearizer.
•H D Thesis
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level. In addition, maximum total phase shift decreases from 6.2 degree to only 1 degree.
Figure-4.2.11 shows the improvement of third order inter-modulation (IM3) product at
different output power back off. It shows that 3.8 dB IM3 improvement is at ldB O/P
back off and 8.7dB IM3 improvement is at 2 dB O/P back off.
Fig.-4.2.11: IMD of SSPA with and without linearizer
4 .2 .4
T em p eratu re C om pensation
As discussed in the previous chapters that, RF resistance of Schottky barrier diode and
PIN diode changes with temperature when diodes are biased by constant current or
constant voltage bias conditions. Therefore, amplitude and phase characteristic of the
linearizer will also vary with change of temperature and degrade the performance of the
overall system.
Figure-4.2.12 shows the measured amplitude variations and Figure-4.2.13 shows the
measured phase variation of the distortion generator circuit over different input power
levels at three different temperatures of -10, +25 and +60 °C. Amplitude variation is
nearly 5-dB at RF input level of -30 dBm and nearly 2-dB at -17 dBm. Whereas, drastic
phase variation is observed at fixed current bias condition, over the entire RF power
level, with the variation of diode temperature. Experimentally it is also observed, that to
achieve temperature invariant performance of the distortion generator, requirement of
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o
4
00
i
*4
t
M.
<
**
(8P) flZSf
-22 -I------------ ■
------------ ,------------ .------------35
-30
-25
-20
-15
RF Power Level (dBm)
Fig.-4.2.12: Measured |S 2 l| variation of distortion generator circuit
over RF power level under constant current bias condition.
voltages/current is not only function of temperature but also it is function of input RF
power level. Thus, for temperature compensation of the linearizer circuit for entire RF
power levels, it is required to vary the voltage/cuirent by sensing temperature as well as
RF power level of the Schottky diode.
Here we have proposed a veiy simple technique [41], [21] - [24], which will ensure
temperature invariant RF resistance of the individual diode leads to temperature invariant
performance of the linearizer. Where for both PIN diode and Schottky barrier diodes are
Fig.-4.2.13: Measured S21 variation of distortion generator
circuit over RF power level for constant current
bias condition.
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biased by the optimum load line biasing scheme as discussed in chapter II.
Figure-4.2.14 shows the measured amplitude and phase characteristic of the Schottky
barrier diode based distortion generation circuit under optimum bias voltage of 0.75
Volts under different temperatures of -10, +30 and +60 °C. The measurement shows that
practically there is negligible amplitude and phase variation with change of temperature
at small signal condition. However, with the increase of RF power level, it shows slight
amplitude and phase variation with change of temperature. At higher power level, this
deviation is due to the self-bias effect of the Schottky diode. The measured plot shows
that due to temperature change of -10 to +60 °C, over the RF power level of -30 to -10
dBm the amplitude variation remains within ±0.6 dB and phase variation remains within
±1.5 degree.
PIN diode based attenuator also separately tested with the proposed optimum load
m ) jlZ S j
Input RF Power Level (dBm)
Fig.-4.2.14: Measured S21 variation with temperature for
optimum load biasing
line biasing technique [21] - [24] of optimum open circuit voltage of 1.19 Volts. The
attenuation variation remains within ±0.2 dB over the temperature range of -10 to +60
°C.
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114
Complete linearizer circuit is also tested over the temperature range of -10 to +60 °C.
Measurement shows that the amplitude variation remains within ± 0.5 dB and phase
variation remains within ±1.2 degree over the entire operating RF power level of -30 to 10 dBm.
4 .2 .5
C o n c lu s io n
This chapter discussed a diode based novel linearizer for high power TWTAs and
SSPAs. The proposed circuit is capable very wide range of amplitude and phase
nonlinearity of TWTAs and SSPAs. Temperature behavior of the linearizer is also
shown. Temperature compensation of the linearizer, using proposed optimum load line
biasing technique shows that very simple bias circuit can provide temperature
compensated RF performance over a very wide range of amplitude and phase variation.
The compensation circuit is also applicable for wide range of RF power level over a
broad temperature range.
c h e s ts
fAto
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S C (Bera, Qujarat U niversity, India
115
4 *3
A T e m p e r a t u r e C o m p e n s a t e d O v e r -D r iv e
L e v e l C o n t r o lle r F o r S S P A s
Reliability of GaAs power FETs used in SSPAs is affected adversely when operated
under overdrive (excess input power) condition for a long time due to excess gate
current. Manufacturer of SSPAs uses different schemes to protect FET power devices
from over drive condition. One method for over-drive level control is by distributing
compression in device line-up and another is by using limiter circuit. However, in both
the cases it is very difficult to fully protect the devices from overdrive condition, which
becomes worse when SSPAs are operated at varying temperature environment. Some
manufacturer also uses feedback control loop for over drive control but its limiting
power level is highly sensitive to temperature. Temperature compensated closed loop
overdrive control circuit can eliminate all the above problems.
Here detailed description with mathematical expressions and test results of a
temperature compensated closed loop overdrive level control (OLC) circuit is given [80],
The proposed scheme can properly protect FET power devices from over drive condition
irrespective of change of temperature and given mathematical expressions helps to
design the temperature compensation network without trial and error.
4.3.1
D iffe re n t OLC S c h e m e
Manufacturer of SSPAs uses different types of schemes to protect GaAs power FET
devices from over drive conditions.
In some cases, proper device line-up selected to distribute overdrive power among the
devices in SSPA line up, so that power devices cannot be compressed heavily and gate
current of GaAs FET devices remains within limit. However, in this case, device
selection in the line up is not simple. Also precise overdrive protection of the device is
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not possible without compromising the non-linearity of SSPAs, which becomes worse
when SSPAs are operated at varying temperature environments.
Another approach to protect FET devices from overdrive power is use of Schottky
barrier diode (SBD) based or GaAs FET based limiters [78], [79] at microwave
frequency. However, the limiter can operate only at low power level range. Therefore,
gain variation over temperature of the intermediate stages used in between power device
and limiter circuit will change the limiting power level. Thus, in practice limiter circuits
are also not suitable to properly protect the FET power devices used in SSPAs under
over drive condition.
Some manufacturer of SSPAs uses closed loop power control systems in SSPAs for
this purpose. However, due to high closed loop gain, the limiting power level is very
much sensitive to variation of the detector diodes behavior over the temperature.
Temperature compensated closed loop over drive control circuit can eliminate all
these disadvantages.
4 .3 .2
O p e ra tio n o f C lo sed Loop OLC
A simplified schematic block diagram of a closed loop overdrive level control system
is shown in Figure-4.3.1. This scheme is capable to protect high power FET devices in
power amplifier as well as small signal and medium power FET devices in driver
Variable
Attenuator
FET
Vc
Fig.-4.3J: Block diagram of proposed OLC
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amplifier from overdrive conditions. Since power sampler is used to sample and detect
the power level, the scheme can be used to control any level of high power range.
Photograph of the C-Band SSPA based on several small signal MESFETs (MGF1423)
and power MESFETs (MGF2407 & MGF2430) of Mitsubishi is shown in Figure-4.3.2,
which is used to demonstrate the proposed temperature compensated over-drive level
control system.
Mainly Schottky diode detector is used at microwave frequency to detect microwave
power. Control voltage Vc is applied to the detector diode to finely adjust the limiting
power level. When coupled power exceeds a certain specified level determined by the
barrier potential (</%) of Schottky barrier diode and given control bias Vc, then there will
be detected output voltage. This detected voltage is amplified by DC amplifier and is fed
to a voltage variable attenuator. Commonly PIN diode-based attenuators are used as a
voltage variable attenuator. Depending upon required amount of over drive level
protection, single or multi-stage attenuators can be used.
Test response of a closed loop OLC system is shown in Figure-4.3.3, where Pi_N
and Po_N are the normalized input and output power [80], Here different limiting power
level setting is achieved by controlling bias voltage (Vc) of the Schottky barrier detector
diode.
Fig.-4.3.2: C-Band SSPA to demonstrate OLC Scheme
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Fig.-4.3.3: Closed loop OLC response under different control bias
voltages
4 .3 .3
T e m p e r a t u r e B e h a v io r o f C lo s e d L o o p O L C
Behavior of PIN diode attenuator and driver amplifier circuit of OLC system changes
with temperature. RF resistance of PIN diode of variable attenuator circuit changes with
temperature causing variation of attenuation of PIN diode attenuator. Gain of FET
devices used in driver amplifier circuit also changes with temperature. Due to variation
of attenuation of attenuator and gain of driver amplifier, the input power of the detector
will vary with variation of temperature.
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Also I_V characteristic of detector diode changes with temperature as shown in the
Figure-4.3.4. In case of hard limitation of power level, gain of DC amplifier is very
high. Therefore, small change of detector diode voltage causes significant change of
limiting power level. Tested results of variation of limiting power level of OLC system
due to temperature variation of only the detector diode is shown in Figure-4.3.5. The
limiting power level variation observed near about 3 dB peak to peak for the temperature
variation o f -10 to +60 °C.
4 .3 .4
T e m p e r a t u r e C o m p e n s a t io n o f O L C S y s te m
Variation of limiting power level due to attenuation variation of attenuator and gain
variation of driver amplifier are eliminated inherently due to the closed loop negative
feedback behavior of the OLC system. Since gain of the DC amplifier is very high, when
power level at the detector input changes due to the temperature variation then detector
output changes automatically to adjust the attenuation of the attenuator to keep the input
of the detector fixed.
Fig.-4.3.5: OLC response due to the variation of temperature of the detector diode
Therefore, as long as the detector diodes behavior is unchanged, power variation at
the input of the detector, by any means, will be compensated by automatically changing
the detected output voltage, which leads the change of attenuation of the attenuator such
that power level at the input of the detector remains constant. Thus, attenuation and Gain
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RF
Temperature Dependent Supply
RF
Fig.-4.3.6: Block diagram of OLC circuit
variation of the attenuator and FET devices of the driver amplifiers respectively will not
affect the limiting power level of the OLC circuit if the voltage variation of the detector
diode can be compensated over the temperature range of interest.
A temperature compensated OLC scheme is shown in Figure-4.3.6 [80]. The
temperature dependent control voltage Vc biases the detector diode Ds through a bias
network containing resistor Rb and a RF decoupling network containing 7J4 transmission
line and a capacitor Cb.
Output of the detector diode filtered by a low pass filter is then connected to the one
input of the differential DC amplifier. Other input of the differential amplifier is
connected to the control voltage Vc. Output of the differential amplifier is connected to
the control terminal of the PIN diode attenuator circuit.
When RF power is below the limiting power level (threshold power level) then there
will be no detector diode current, therefore, voltage at the two inputs of the differential
amplifier will be same; it leads no output voltage and no attenuation of the attenuator.
When the RF power amplitude crosses the threshold power level then detector diode
starts to conduct. Therefore, there will be voltage difference between the inputs of the
differential amplifier, which leads to a finite output voltage at the output of the
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differential amplifier. This output voltage sets the attenuation of the attenuator so that the
RF power will be limited to the predetermined limiting power level. Changing the
control voltage (Vc) to the diode can compensate for the variation of limiting power level
due to variation of detector diodes voltage (V5bd) over the temperature. For this,
combination of detector diodes voltage and control voltage that is (VW-Vc) should
remain invariant over the temperature range of interest. Therefore, the condition for
temperature compensation is:
dV„
dVsbd
dT
dT
(4.3.1)
I-V characteristic over the temperature, for a p-n junction diode and Schottky barrier
diode are similar in nature. Therefore, temperature dependent voltage source Vc
generated by using temperature dependent voltage drop of a p-n junction diode D l.
In the circuit as shown in Figure-4.3.6, the control voltage Vc is given by:
Vc = r x - ^ - ( l - r y ]
(4.3.2)
where, G = l + R4/R 3 , r - R 2/ R l and Vpnd is the voltage of the p-n junction diode D 1.
Therefore, combining equation (4.3.1) and (4.3.2), the condition for compensation
becomes:
^pnd
^
1+ r ’ dT
_ d V sbd
g
dT
K' ' }
Now, the temperature dependent I-V relationship of the diode is given by [1]:
I d = A.TmExp (-
tjJcT
Exp <lVd
J].kT
x
(4.3.4)
where, Vj is diode voltage, A is the area factor of the diode independent on temperature,
1i
is diodes ideality factor, Eb is band gap potential of the semiconductor in case of p-n-
junction diode and barrier potential
(</>b)
in case of Schottky barrier diode,
m
is the
temperature exponent of the diode, k is the Boltzman’s constant and q is the electron
charge.
And load line equation is given by:
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Vs =Vd + I d x R b
(4.3.5)
where, Vd is the diode voltage at current Id when biased by a voltage source Vs through a
bias resistor /?*,. For forward biased diode where the reverse saturation current is
negligible compared to the total forward current, the rate of change of diode voltage with
respect to the temperature is given by [from expression (4.3.4) and (4.3.5)]:
dVd
dT
Eb 4-mJ}.VT - V A
- I d.R„
{ld.Rb +T}VT)
(4.3.6)
where, VT = k T / q . This expression (4.3.6) is valid for both the p-n junction and
Schottky barrier diode. Thus, knowing the diode parameters present in the expression
(4.3.6), the circuit component values for the temperature dependent voltage source can
be derived from expressions (4.3.2), (4.3.3) and (4.3.6).
4 . 3 .5
C ir c u it R e a liz a t io n a n d T e s t R e s u lt
The schematic circuit as shown in Figure-4.3.6 has been implemented for a C-Band
solid-state power amplifier. Voltage variable attenuator, using PIN diode HPND 4005,
has been realized in alumina substrate (er = 9.9) by using 3-dB Lange coupler in a micro­
strip line configuration. Power sampler circuit and detector circuit using Schottky barrier
diode HSCH 5315 also realized on the Alumina substrate. To generate the temperature
dependent control voltage p-n junction diode 1N5806 used with supply voltage of ±5
Volts. High gain differential DC amplifier realized using operational amplifier (jxA741).
Typical parameters for Schottky barrier diode HSCH-5315 are: Eb = 0.69 Volts, T] =
1.1, m = 3 and Vsm = 0.28V. Detector diode operates under very high bias resistor
therefore,
tj.V t
« I d.Rb■ Putting all these values dVSbd!dT = 1.6 mV/ °C at room
temperature. And typical parameter for p-n junction diode IN-5806 are: Eb - 1.16 Volts,
7 = 1.5, to = 1.5 and Vpnd = 0.6 Volts. Therefore, dVprJ d T = 2 . 1 mV/ °C at room
temperature. Putting these values to expression (6.2.2) and (6.2.3) and considering the pn junction diode operates at current of 10 mA, and control voltage (Vc) requirement at
room temperature is 0V, the circuit parameters becomes: RJ = 5000, R2=440£l, R3=]M1
and R4 = 4330.
PhD 'Thesis
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123
The test result of the temperature compensated OLC circuit is shown in Figure-4.3.7.
From this figure, it is clear that limiting power level remains nearly invariant over the
temperature range o f-10 to +60 °C.
6 0 deg.
a
♦
•
2 5 deg.
-1 0 deg.
•
W /O A L C
l
a
i
Po_N (dB)
L
/
V*'
__ 4
i
* > - * • —*■
CO
•
*
__ { i __
t
L __
5
-3
-
;
7
<
11
13
15
17
19
21
23
25
Pi_N (dB
Fig.-4.3.7\ Complete OLC circuit response over the temperature variation
4 .3 .6
C onclusion
Here detailed description with mathematical expressions and test results of a
temperature compensated closed loop overdrive level control (OLC) circuit is given. The
proposed scheme can properly protect FET power devices from over drive condition
irrespective of change of temperature. Thus, the circuit will be very much useful to
protect sensitive MESFET, HFET, HFET microwave power devices from overdrive
power conditions and will increase life of SSPAs. The given mathematical expressions
help to design the temperature compensation network without trial and error.
Temperature compensated closed loop control circuit scheme presented here can be
used not only for SSPAs but can be used in any microwave systems for power level
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control. These types of microwave power limiters are also suitable for signal processing
for example to suppress AM components before demodulation of a microwave FM
signal since PIN diode attenuator offers very low phase variation over the attenuation
range.
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125
4 .4
>1 Tem perature C om pensated C hannel
A m p lifie r W ith A u to m atic Level Control For
S a te llite Transponder
Here a design, development and test results of a Ku-band channel amplifier with
automatic level control (ALC) system for satellite transponder is discussed. A systematic
temperature characterization and optimization procedure depending upon measurement
data of the channel amplifier is presented. This procedure takes into account the effect of
parameters variation from one unit to another and eliminates conventional trial and error
method to determine the optimum component values of the temperature compensation
circuits.
Satellite communication link at Ku-band frequency in tropical region faces excess
up-link and down-link path loss due to rainfall. The up-link rain attenuation causes
decrease in signal level received at satellite receiver which leads to decrease of satellite
transmitter power. The signal level is further reduced by down link rain attenuation. This
may cause signal level to fall below threshold level of the ground receiver for specific
BER performance. Adaptive power control system at satellite transmitter [81], [82] and
ground receiver can solve this problem. Here design, development and characterization
of a Ku-band channel amplifier with automatic level control (ALC) system is discussed
for spacecraft application to control input of the final power amplifier (TWTA or SSPA)
according to the signal level arriving at the channel amplifiers input [83]. Thus, ALC
system also protects the final power amplifier against any accidental high power from
up-link as discussed in the previous section in over-drive-level control system.
The presented channel amplifier [83] can operate in ALC mode as well as in fixed
gain mode (FGM). In fixed gain mode, gain will be 44 dB and in ALC mode, gain will
vary automatically from 39 dB to 59 dB depending upon the input power level. This
amplifier has a commandable gain control (22 dB) system to operate the final power
amplifier in different back-off conditions in both modes of operation.
(PhU Thesis
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126
The required gain of the amplifier is achieved by using three numbers of amplifier
modules using HEMT devices (CFY67-08). In ALC mode, the channel amplifier
operates as a closed loop feedback system. The amplifier contains a Schottky diode
detector to detect the sampled RF power. The detected voltage amplified by a differential
DC amplifier and applied to the control input of a variable PIN diode attenuator.
Attenuation of the attenuator will vary according to the input power level so as to
maintain the constant output power level of the channel amplifier. Another PIN diode
attenuator is used for commandable step gain control.
It is known that gain of the HEMT based amplifier module, attenuation of the PIN
diode attenuator and detected power level of the Schottky diode detector, all are function
of temperature. Thus, suitable compensation circuits are included to compensate the
temperature variation of the channel amplifiers performance over the qualification
temperature range for satellite applications. A practical systematic procedure based on
the measurement data instead of conventional trial and error method is presented to
determine the component values of the compensation networks.
4. 4.1
B lo c k S c h e m a tic o f th e C h a n n e l A m p lifie r
The basic block schematic of the channel amplifier with ALC is shown in Figure4.4.1. The amplifier module A l, A2 and A3 are used to meet the total gain requirement
of the channel amplifier. The voltage variable attenuator ATI is for automatic gain
control of the amplifier depending on the input power level. The range of gain control by
RF
OUT
RF
IN
Fig.-4.4.1: Basic block diagram of the channel amplifier with ALC
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127
this attenuator is +15 dB to -5 dB over the nominal gain of 44 dB. Attenuator AT2 is for
the commandable gain setting of the amplifier up to -22 dB, in steps of -2 dB.
For ALC function, RF power is sampled after amplifier A2 and detected by the
detector diode. This detected voltage is applied to one of the input of the differential DC
amplifier. Other input of the differential amplifier is connected to the reference voltage
Vr2 - Voltage Vr2 will determine the output power level of the channel amplifier. Output
of the DC amplifier connected to the ALC attenuator ATI, through an analog switch.
Switch position will determine the mode of operation of the channel amplifier. When the
switch connects attenuator ATI to the other reference voltage Vri, then the operation of
the channel amplifier will be in fixed gain mode (FGM). Reference voltage Vri will
determine the gain of the channel amplifier in fixed gain mode operation.
The gain of the microwave amplifier modules, attenuation of the PIN diode
attenuators and detected power level of the Schottky diode detector are all functions of
the temperature. Thus, suitable compensation circuits are required to compensate the
temperature variation of the channel amplifiers performance over the qualification
temperature range for satellite transponder.
The schematic circuit diagram of the different temperature dependent control signals
is shown in Figure-4.4.2. Here, temperature controlled reference voltage Vri is generated
by using thermistor to compensate the gain in FGM operation. Whereas, temperature
controlled reference voltage Vrz is generated by using P-N junction diode to compensate
the output power level variation in ALC mode operation. To achieve temperature
invariant accurate step attenuation, proposed optimum bias load line technique used with
analog switch CD4051 and bank of resistors, as discussed in chapter-II for PIN diode
attenuator.
4.
4.2
R e a liz a tio n o f In d iv id u a l M o d u le s
Amplifier modules A l, A2 and A3 are used to achieve required channel amplifier
gain. A l & A2 are the three-stage amplifier and A3 is a two-stage amplifier. All the
amplifier modules are realized using pHEMT device of type CFY67-08 for Ku-Band
*Pk(D ‘Thesis
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. $024 19*6 January, 2004 .
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128
/
11
MS §oibuV
‘K es "Hp. 5024.19“ January, 2004.
/
ThV Thesis
A
HA/VO > ~ |
a
fn,
hq
" 1
Si
►
Fig.-4.4.2: Block diagram with control signals.
V)
-V2
±n±
i, *
s C ‘B ern, gujarat University, India.
Step G ain Set.
C ontrol
DATA
j
A LC M ode Tem p.
C ontrol
*------ o > ■
FG M Temp.
Control
i £
Mode
Selection
3
$2
> 0 ------- VAr— W r l
---------
+/
downlink frequency of 11.45 to 11.7 GHz. MIC assembly drawing of the three stage
amplifier is shown in Figure-4.4.3.
Reactive matching networks are used for input and output matching of the devices.
Resistive loading with open circuit stubs are used at input and output cards to achieve
high stable amplifier. Drain and gate bias resistors with appropriate capacitors are
accommodated within the MIC cards, which leads to very compact circuit with improved
out-of band stability.
Fig.-4.4.4: MIC assembly of PIN diode attenuator
All the matching networks of amplifiers are realized in a 25 mil alumina (AI2O3)
substrate (£,■= 9.9) of size 0.25" x 0.5", with the accommodation of all the bias resistors
in the alumina substrate at the RF tray. Two-stage amplifier A3 is realized with the same
MIC cards used in the three-stage amplifier, by eliminating one inter-stage matching
■P/t© Thesis
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130-
network of three-stage amplifier. These circuits are simulated and analyzed by series IV
circuit simulator of HP (EEsof).
To adjust overall gain of the channel amplifier the resistors Rdi, Rd: & R d3 are used
in the +V1 supply lines o f the amplifier A l, A2 & A3 respectively.
Attenuator for step gain control (22 dB) and automatic level control (20 dB) are both
the two-stage PIN diode based voltage variable analog attenuator. MIC assembly
drawing o f the attenuator is shown in Figure -4.4.4. D l, D2, D3 & D4 are the beam lead
PIN diodes (MPND-4005) mounted in a 3-dB Lange coupler designed at Ku-band
frequency. In this configuration, the circuit will provide maximum attenuation when PIN
diode resistance will be 5 0 0 which will be determined by current through the diode.
The photograph o f the integrated RF tray is shown in Figure-4.4.5.
Fig.-4.4.5: Ku-band channel amplifier with ALC Scheme
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131-
4 . 4.3
T e m p e ra tu re C o m p e n s a tio n o f th e C h a n n e l
A m p lifie r
It is known that gain o f HEMT based amplifier module, attenuation o f PIN diode
attenuator and detected power level o f Schottky diode detector are function o f
temperature. Moreover, temperature dependency o f on resistance o f analog switch
(CD4051) is influence the temperature characteristic o f the channel amplifier. Thus
suitable temperature compensation circuits, with suitable method to determine the
component values, are required to compensate the temperature variation o f the channel
amplifiers performance over the qualification temperature range for satellite applications.
There are various temperature compensation schemes for ALC loop compensations [30],
however, here a novel techniques are used for this purpose [83]. There are two
temperature- dependent control signals V ri and V r2 generated to compensate the
temperature variation o f the channel amplifiers gain at FGM and ALC mode
respectively. Another control signal V r3 ( Figure-4.4.2) is generated to achieve
temperature invariant step gain setting. The practical procedures to determine the
component values o f the control circuits in different mode o f operations are discussed in
the following sections.
4.4.3.1 T e m p e ra tu re C o m p e n s a tio n fo r F ix ed G ain M o d e
In fixed gain mode it is to be ensured that, gain o f the channel amplifier will be
within the specified limit over the operating temperature range. The function o f the
control signal V ri is (a) to set the gain o f the amplifier 44 dB and (b) this gain should
remain within the specified limit (peak to peak <0.8 dB) over the temperature range o f 10 t o +60 °C.
To generate temperature controlled reference signal V ri, thermistor Rth used with
the resistors R fi, R f2 & Rra as shown in Figure-4.4.2.
Equivalent circuit o f the V ri signal generator with attenuator and analog switch is
shown in Figure-4.4.6. Here, reference point taken before analog switch to include its
(resistance o f analog switch) temperature variation. Where VFo(T) and R fo(T) are the
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function of temperature due to dependency on
R th (T )
and given by the equations (4.4.1)
and (4.4.2) respectively.
Vt[RF2 l l ( * „ + * 7H ( r ) )]
(4.4.1)
R fo (T ) = RFi II [RF2 II ( * „ + R th (7*))]
(4.4.2)
Suppose at temperature T, voltage at the reference point and current to the attenuator (as
shown in Figure-4.4.2) is VRi(T) and Iri(T)] to achieve required gain (44 dB) setting of
the channel amplifier, then from the circuit of Figure-4.4.6 we can write the equation
(4.4.3).
CO - VRl ( r ) = R fo {T)x l n (T)
To determine the circuit component values (R f i ,
R f2
(4.4.3)
& R f 3) of this network it is required
to take three sets of voltage and current readings [{Vri(Ta), Iri(Ta)}; (Vri(Tc), Ir(Tc)} ;
(Vri(Th), Iri(Th)} ] at three different temperatures (TA, TC,TH) by setting the amplifiers
gain at the required value (44 dB). The optimum value of Rfi, Rfi & Rf3 will be the one
which
satisfies the following three equations (6.4.4A),
(6.4.4B)
& (4.4.4C)
simultaneously.
vn <Ta ) -
V * (Ta) - R fo{Ta ) x /„ ( T a)
(4.4.4A)
VFo<Jc)~
V*,(Tc ) = R
(4.4.45)
M x I* (Tc )
<Th ) - V R]{Th ) = R fo{Th ) x I n (Th )
Vpo(T)
(4.4.4C)
Rfo(T)
(VR1(T),IRI(T))
Fig.-6.4.6: Equivalent circuit for V ri
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Based on these three equations the characteristic of the temperature-compensation
circuit, in fixed gain mode, is modeled using a spreadsheet. The three unknown
parameters R fi , R iv and R& were varied interactively until all these three equations
(4.4.4A, B, C) satisfied simultaneously. The particular values of R f i , R f2 and R f3 for
which all these three equations satisfied simultaneously with minimum error, will be the
optimum value of these resistors to provide temperature compensated gain in fixed gain
mode operation of the channel amplifier.
For utmost accuracy the resistance of the actual thermistor, ultimately to be used in
the compensation circuit, should also be recorded at three different temperatures. The
three temperatures should be stabilized before taking the measurements to minimize the
thermal gradient between thermistor and the circuit components.
4 A .3 .2
T e m p e ra tu re C o m p e n s a tio n fo r A LC M o d e
In ALC mode, the detector circuit will detect RF power, and there will be voltage
drop across the load resistor R aw - This voltage will be amplified by the differential
amplifier and fed to the attenuator ATI through analog switch. The reference voltage Vfe
will determine output of the channel amplifier in ALC mode. Variation of the output
power over the range of the input power variation will be determined by the overall gain
of the ALC loop. Loop gain is determined by the resistor R ah, R a% Raio & R a h - Offset
current provides through the resistor Ra? will ensure the output of the differential
amplifier at negative potential at very low RF power level.
Characteristic of the detector, amplifier modules and attenuators are changes with
temperature. Thus, reference voltage Vfe should be a temperature dependent voltage to
achieve required fixed output power level within the specified range over the operating
temperature range.
To generate temperature controlled reference voltage V®, p-n junction diodes are
used as shown in the Figure-4.4.2. The equation (4.4.5) and (4.4.6) will determine
current (/<*) through the diodes and voltage (Vd) across the all diodes.
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M n
, m
=
f(y, (r))
(4 .4 .5 )
V,-Vn(T)-lAT)*R,2+V1 vA T )+ lA T )x R ,z-V z
*,
4
(446)
* ,,+ * .,4
' ,s l ( r ) x [ « „ + * 4 4 ] = K ( J ’) + / I, a ’) x R ,J - F 2] x « „
(4.4.7)
We can assume that there is no current drawn by the detector diode & differential
amplifier from the reference supply Via. Therefore, at a particular temperature T, if the
reference voltage requirement is Vja(T) then from Figure-4.4.2 one can write equation
(4.4.7).
The circuit component values (Rai, Ra2, Ra3
& Ra4) of this network can be
determined by taking three sets of voltage readings [Vjk(ZaA Vm(Tc), VR2(Th)} at three
different temperatures (Ta, Tq Th) by setting the amplifiers output power level at the
required value ( say 0-dBm). Then the optimum value of Rai, Ra2, Ra3 &Ra4 will be the
one which satisfies the following three equations (4.4.8P), (4.4.8Q), and (4.4.8R)
simultaneously.
Vr2(Ta ) x [RA3 + r J = [ vd (Ta ) + I d (Ta ) x R a2 - V 2] x R m
(4.4.8 P)
VrA T c ) x [Ra3 + R a4]=I v d (Tc ) + I d (Tc ) x RA2- V 2]x R m
(4.4.8 Q)
VR2 {TH) x [ R A2 + RM} = ty D{TH) + l D{TH) x R A2- V 2} x R M
(4.4.8P)
Based on these equations 4.4.5, 4.4.6 and 4.4.8P, Q, R, the characteristic of the
temperature-compensation circuit, in ALC mode, are modeled using a spreadsheet. The
four unknown parameters R a i , R a2, R a3 and R a4 were varied interactively until all these
three equations (4.4.8P, Q, R) satisfied simultaneously. The particular values of Rai, Ra2 ,
Ra3 and Ra4 for which all these three equations satisfied simultaneously with minimum
error, will be the optimum value of these resistors to provide temperature compensated
output power level in ALC mode operation of the channel amplifier.
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4 A .3 .3
T e m p e ra tu re C o m p e n s a te d S te p G ain C o n tro l
Step attenuator AT2 is for the commandable gain setting o f the channel amplifier.
Proper load line selection technique [21] - [24], [83] is used to achieve temperature
invariant attenuation o f the PIN diode based attenuator AT2. Analog switch (CD4051)
with a bank o f resistors Rsi to Rsn and reference voltage Vm is used for the control signal
o f the PIN diode attenuator circuit.
Reference voltage Vr3 is the critical parameter to achieve temperature invariant
attenuation. The resistor R ri and R r2 determine this reference voltage. To determine R ri
and R r2 , it is required to take three sets o f voltage and current readings [(Vat2 (Ta),
Iati(Ta), Vsw(Ta)}; {Vxn(Tc), h n (T c), Vsw(Tc)}-, {VAn(TH), IAn(TH), VsMT h)}] at three
different temperature (Ta, Tc, Th) by setting the attenuation o f the attenuator preferably
at the maximum required value (say 22-dB). Then plotting these data, as shown in
Figure-4.4.7, one can get reference voltage Vr3 . Where, Vrj is the intercept point o f the
load line with the voltage (VAt2 +Vsw) axis. Then R ri and Rr2 will be determined by
satisfying the equation (4.4.9) and putting the constraint on current drawn from the
source Vj.
_
R3
x RR2
(4.4.9)
Rm + Rtn
The resistance values Rsi to Rsn can be determined by setting the reference voltage V r3
and then adjusting these resistor values to achieve different required step attenuation (2,
4, 6,— dB). This can be done at any temperature.
(JAn)
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In this method, temperature variation of the ‘analog switch resistance’ has been taken
into account by plotting the load line with respect to voltage {Vat2 + Vsw)- Therefore, this
procedure eliminates the separate temperature characterization of the ‘analog switch
resistance’ over the temperature.
4 .4.4
M e a s u re m e n t a n d Test R esults
The integrated channel amplifier has been fabricated and tested over the Ku-band
down-link frequency of 11.45 to 11.7 GHz. Measured 1-0 characteristics of the channel
amplifier in ALC mode, without temperature compensation, is shown in Figure-4.4.8.
Measured temperature compensated 1-0 characteristic in ALC mode is shown in Figure4.4.9. The measured output power variation is within 1-dB over the input power
variation of -59 to -39 dB.
As shown in Figure-4.4.8, without temperature compensation circuit, the output
power variation in the ALC mode over the temperature range of -10 °C (cold) to +60 °C
(hot) is nearly 4.5 dB. The required reference voltage (Vr2) is measured at cold, ambient
Fig.-4.4.8:1-0 characteristic without temperature compensation.
and hot temperature to achieve same output power (0-dBm) of the channel amplifier.
The optimum component values RA/, RA2, Ra3 & RA4 are determined by satisfying the
P(x<D Thesis
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equations (4.4.8P), ( 4 .4 .8 0 & (4.4.8/?) simultaneously. 1-0 characteristic o f the channel
amplifier with the optimum components value o f temperature compensation circuit in the
ALC mode is shown in Figure-4.4.9 at three different temperatures. With temperaturecompensation circuit, the measured variation reduces to less than 0.4 dB over the
temperature range of -10 to +60 °C.
In fixed gain mode, without compensation circuit the gain variation o f the channel
amplifier is nearly 5-dB over the temperature range of -10 to +60 °C. The three sets of
voltage and current readings ( Vri, //?/) at cold, ambient & hot temperatures are measured
to get same gain of 44 dB. The optimum values o f the resistors Rfi, Rf2 , Rf3 o f the
compensation network are determined for 5 k£2 thermistor by satisfying equations
(4.4.4A), (4.4.4 B) & (4.4.4C) simultaneously. With the optimum components value of
the compensation network, the measured gain variation over the temperature range
becomes 0.4 dB p-p.
The optimum reference voltage Vrj is determined by the optimum load line selection
technique. The three sets o f voltage and current [(V)i7 2 +Vs), (IAn)] readings at cold,
ambient & hot temperatures are measured to get same attenuation of 22 dB. Plotting
these data, the optimum voltage (V/u) is 0.963 Volts. For this optimum reference voltage,
Fig.-4.4.9 :1 -0 characteristic with temperature compensation.
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the resistor values Rri, Rr2, and Rsi to Rsn are determined. With this reference voltage,
the achieved step attenuation accuracy is within ±0.4 dB for all the steps (-2, -4, ...,-22
dB) over the temperature range of -10 to +60 °C.
4 . 4 .5
C onclusion
This paper discusses the design, temperature characterization and test results of a Kuband channel amplifier with automatic level control (ALC) system for spacecraft
application. The presented practical procedure to determine the component values of the
compensation circuits will be very much useful to optimize the performance of the
channel amplifier without characterizing the individual modules. This systematic
procedure takes into account the effect of parameter variation from one unit to another
and eliminates conventional trial and error method leads to minimize man hour time.
th esis
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4.5
CONCLUSION
This chapter discussed the temperature behavior and compensation mechanism of
different microwave subsystems. Mathematical and practical design procedures are also
developed and discussed to determine the circuit component values. Different devices
are temperature compensated by different techniques depending upon the function of the
devices. For example, in case of PIN diode and Schottky barrier diode, optimum load
line technique is used in case of vector modulator and linearizer application, in which,
temperature insensitive RF resistance is required. Whereas, P-N junction diode is used
for OLC and ALC application, in which it is used as RF detector. Overall gain of the
channel amplifier based on MESFET and HEMTs are stabilized using temperature
dependent variable PIN diode based attenuators. In all the cases temperature invariant
attenuation is achieved with the use of PIN diode attenuator with proposed optimum load
line biasing scheme.
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TktD *Tfosts
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5.0
S U M M A R Y CO N C LU SIO N A N D FUTURE S C O P E
The thesis work addresses temperature behavior and compensation technique
of PIN diode, Schottky barrier diode and EET based microwave circuits. The
research
work proposed and successfully demonstrates a novel temperature-
compensation technique, based on optimum bias load line technology, which
compensates temperature behavior of PIN and Schottky barrier diode based
microwave circuits, which is more accurate, simpler and more
reliable than
previous reported circuits. The work shows that the same technology is also
applicable for temperature compensation of light emitting diode. The thesis work
also presents and demonstrates practical and theoretical design details of the
temperature compensation mechanisms of various microwave circuits and sub
systems based on MESFET, HEMT, PIN diode and Schottky barrier diode.
5 .1
S U M M A R Y a n d CO NCLUSIO N__________________
A hitherto unexplored mathematical property of PIN diode, Schottky barrier
diode and Light Emitting Diode has been discussed. This property is exploited in
a simple and easily adjusted control circuit that provides excellent setting
accuracy and temperature stability. It is shown that this approach can be used for a
wide range of practically available diodes. This circuit uses no separate
temperature sensor or compensating mechanism, but responds directly to the
junction temperature of the diodes. This prevents any errors caused by
temperature gradients, or by self-heating of the diodes due to high RF levels. The
proposed novel technology provides setting and temperature accuracy, which is
Vh(D ‘t hesis
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better than accuracy available from MMIC based passive digital attenuator
circuits.
5. 1.1
T e m p e ra tu re D e p e n d e n c y o f B a n d g a p
P o te n tia l
Temperature compensation technique of PIN diode and light emitting diodes
based on the optimum bias load line technology, where open circuit voltage of the
control circuit depends upon bandgap potential of the semiconductors. All the
above discussion assumes that band gap potential is independent on temperature
or
any
other
environment
condition.
However,
bandgap
potential
of
semiconductor is a function of temperature and pressure. At room temperature and
under normal atmospheric pressure, the values of the bandgap are 1.12 eV for Si,
and 1.42 eV for GaAs. These values are for high-purity materials. For highly
doped materials, the bandgaps become smaller. Experimental result shows [1] that
the bandgaps of most semiconductors decreases with increasing temperature. The
bandgap approaches 1.17 eV, and 1.519 eV, respectively for Si and GaAs
semiconductor at 0 K. The variation of bandgap with temperature can be
expressed approximately by a universal function:
r/r2
Eg(T) = Eg(0 ) - —
(5 . 1)
where Eg{0), a and (3 are 1.17 eV, 4.73x104 and 636 K for Si and 1.519 eV,
5.405x1(l4 and 204 K for GaAs respectively.
Near room temperature, the bandgap of Si decreases with pressure, dEJdP - 2.4 xlO"6 eV/(kg/cm2). The bandgap of GaAs increases with pressure, dEJdP = 12.6 xlO'6 eV/(kg/cm2).
J ’fi©
'Thesis
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C Sera, Qujarat University, India.
5 .1 .2
T e m p e ra tu re D e p e n d e n c y o f S c h o ttk y B a rrie r
H e ig h t
Temperature compensation technique of Schottky barrier diodes based on the
optimum bias load line technology, where open circuit voltage of the control
circuit depends upon Schottky barrier height ($g) of the metal-semiconductors
junction of the diode. The change in the Schottky barrier height [52] with
temperature is given by:
0B{T) = </>B(To)+m[Eg{ T ) - E g(To)]
(5.2)
where m is in between 0 to 1, depending upon the surface state energy level.
Thus, Schottky barrier height decreases with the increase of temperature as of
bandgap potential on the semiconductor.
5 .1 .3
T e m p e ra tu re D e p e n d e n c y o f B ias R e s is to r
Temperature coefficient of bias resistors contribute to the temperature
coefficient of RF performance of the diode based circuits. Moreover, on-
L
F ig.S.l : Measurement to consider temperature dependency of
analog-switch, resistor, Eg,
resistance of the analog switches used in the driver circuit for stepwise control of
the RF performance has also some temperature coefficient. Figure-5.1 shows the
‘T&tsis
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S C tBera, Qajarat ‘University, India.
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schematic block diagram of a diode based circuit. The driver circuit contains
analog switch and series resistor, both have finite temperature coefficient. To
determine the optimum open circuit voltage of the circuit, the voltage and current
of the circuit is measured at point ‘C ’ instead at the diode terminal point ‘A ’.
Value of the resistor R is kept approximately equal to the actual value required for
the circuit. The diode (PIN, Schottky or LED) based circuit, analog switch and the
biasing resistor are all kept in the temperature controlled thermal chamber as
shown in the figure. Three sets of voltage and currents are measured at three
different temperatures (two extreme operating temperatures and another at the
middle of the operating temperature) of the chamber, adjusting voltage/current to
the circuit for same performance, and are plotted as shown in the Figure-5.2. The
load line will be the best fitted straight line passing all these three points. The
intercept of the load line with the voltage axis will be the optimum open circuit
voltage of the bias network and inverse slope will give the extra resistor value
required which will be algebraically added to the resistor R.
The presented practical procedure takes into account the temperature
coefficient of the analog switch as well as bias resistor since measurement
I REF (mA)
reference point taken after these components.
Fig.-5.2i Plot of load line
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Today, very precession voltage source, based on the bandgap reference are
available with excellent temperature stability. Thus, the above procedure with the
temperature stable voltage source provides highly accurate temperature stable
performance of the diode (RF performance of PIN and Schottky barrier diode and
illumination of LED) based circuits.
The proposed optimum bias load line technology for PIN and Schottky barrier
diode definitely solve the problems of temperature variation of RF performance of
the diode based circuits. RF circuit manufactures definitely encouraged to use PIN
diode and Schottky barrier diode for dynamically RF signal control circuit such as
attenuator, phase shifter linearizer, etc. Similarly, using optimum bias load line
technology, lighting industry will provide temperature compensated LED
illumination without increasing circuit complexity.
5. 1.4
S e lf-H e a tin g a n d T h e rm a l R u n a w a y
The proposed driver circuits for diode based circuit uses no separate
temperature sensor or compensating mechanism, but responds directly to the
junction temperature of the diodes. This prevents any errors caused by
temperature gradients, or by self-heating of the diodes due to high power
dissipation of the diode.
In case of fixed current bias condition, the device current remains constant
irrespective of the diode temperature. However, when diodes are operated under
temperature compensation condition, the diode current will increase with the
increase of diode temperature this will further increase the diode temperature and
diode current. This effect may cause thermal runaway of the device. Though, the
series resistance of the bias network limit the current to the diode, for more
precaution, it should be ensured that the device operation is within the limited
PAW ‘Thesis
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temperature variation environment to keep the device current and power
dissipation within the allowable limit.
5 .2
FU TU R E S C O P E O F THE W O R K
The thesis work invented and successfully demonstrated the “optimum bias
load line technology” for achieving temperature invariant performance of the
junction diode-based circuits. Although optimum bias load line technology solve
the temperature variation problem of the diode-based circuits, there is no such
type of technology evolved for microwave transistor based circuits. As discussed
in the previous chapters that many ECPs are function of temperature and these are
also vary from one device to another device. Thus, in the design phase it is nearly
impossible to incorporate accurate temperature compensation circuit to meet the
performance requirement over the broad operating temperature environment.
Though, the thesis work proposed and demonstrate the use of separate gain
variation block to compensate the overall gain variation of the amplifier it is not
suitable to compensate the output power variation of the power amplifier.
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