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Low-cost microwave radiometry for remote sensing of soil moisture

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LOW-COST MICROWAVE RADIOMETRY FOR
REMOTE SENSING OF SOIL MOISTURE
by
Eric Ndjoukwe Chikando
A Dissertation Submitted in Partial Fulfillment
Of the Requirements for the Degree
Doctor of Engineering
MORGAN STATE UNIVERSITY
March 2007
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UMI N um ber: 3258433
Copyright 2007 by
Chikando, Eric Ndjoukwe
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ABSTRACT
Title of Dissertation:
LOW-COST MICROWAVE RADIOMETRY FOR
REMOTE SENSING OF SOIL MOISTURE
Eric N. Chikando, Doctor of Engineering, May 2007
Dissertation chaired by:
Carl White, Ph.D.
Department of Electrical and Computer Engineering
Remote sensing is now widely regarded as a dominant means of studying
the Earth and its surrounding atmosphere. This science is based on blackbody
theory, which states that all objects emit broadband electromagnetic radiation
proportional to their temperature.
This thermal emission is detectable by
radiometers - highly sensitive receivers capable of measuring extremely low
power radiation across a continuum of frequencies. In the particular case of a
soil surface, one important parameter affecting the emitted radiation is the
amount of water content or, soil moisture. A high degree of precision is required
when estimating soil moisture in order to yield accurate forecasting of
precipitations and short-term climate variability such as storms and hurricanes.
Rapid progress within the remote sensing community in tackling current
limitations necessitates an awareness of the general public towards the benefits
of the science.
Information about remote sensing instrumentation and
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techniques remain inaccessible to many higher-education institutions due to the
high cost of instrumentation and the current general inaccessibility of the science.
In an effort to draw more talent within the field, more affordable and reliable
scientific instrumentation are needed.
This dissertation
introduces the first low-cost handheld microwave
instrumentation fully capable of surface soil moisture studies. The framework of
this research is two-fold. First, the development of a low-cost handheld
microwave radiometer using the well-known Dicke configuration is examined.
The instrument features a super-heterodyne architecture and is designed
following a microwave integrated circuit (MIC) system approach. Validation of
the instrument is performed by applying it to various soil targets and comparing
measurement results to gravimetric technique measured data; a proven scientific
method for determining volumetric soil moisture content.
Second, the
development of a fully functional receiver RF front-end is presented.
This
receiver module is designed in support to a digital radiometer effort under
development by the Center of Microwave Satellite and RF Engineering
(COMSARE) at Morgan State University. The topology of the receiver includes a
low-noise amplifier, bandpass filters and a three-stage gain amplifier.
Design,
characterization and evaluation of these system blocks are detailed within the
framework of this dissertation.
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LOW-COST MICROWAVE RADIOMETRY FOR
REMOTE SENSING OF SOIL MOISTURE
by
Eric Ndjoukwe Chikando
Has been approved
March 2007
DISSERTATION COMMITTEE APPROVAL:
Chair
Carl White, Ph. D.
-JJtk
Jd tn ^R /P ie p m e ie r, Ph.D.
James Whitney, Ph.D.
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To my family and especially my parents;
And to my wife Lilian and son Eric Jr.;
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A ckno w ledg em ents
It’s been a long journey indeed, towards the completion of this work!
Many people, much more than I can mention within these few lines, played an
active role in shaping me into the person I am today. I am forever indebted to
those individuals.
First and foremost, I am grateful to Dr. Carl White not only for being my
advisor but for believing in my abilities as a student. You have taught me the
essence of conducting technical research and under your leadership, I have
attained new heights. To Dr. Jeffrey Piepmeier at NASA Goddard Space Flight
Center, thanks for your support, guidance and for challenging me in becoming a
better engineer. I will always cherish the memories of our many “lunches” and
lengthy conversations from which, spurred my interest in remote sensing. You
have been truly instrumental in the completion of this work. I would like to thank
Dr. James Whitney for agreeing to serve on my committee. Your timely inputs
were of great value during the course of this work.
I’m thankful to my parents and family for their everlasting support and
encouragement through all my endeavors over the years.
Your empowering
advices and continuous prayers have entrusted me with strength and courage for
always pursuing new horizons. To my wife Lilian, thanks for your tender love and
words of inspiration especially during those “down” periods. Love you always.
iv
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The greater part of the journey leading to the completion of this research
was spent at the Microwave Instrument and Technology Branch (M.I.T.B.) of
NASA Goddard Space Flight Center. There, I was privileged to be surrounded
by a huge pool of highly talented individuals.
I am grateful to Mrs. Catherine
Long and Mr. Terrence Doiron, branch-head and assistant branch-head
respectively, for making my tenure there both, enjoyable and memorable.
Special thanks go to the many acquaintances resulting from that tenure
especially to Mr. Joseph Knubble, Mr. Kevin Horgan, Dr. Kongpop U-Yen, Dr.
Paul Racette, Mr. Fernando Pellerano and Dr. Elissa Levine.
To my fellow peers and close friends at Morgan State University including
Mr. Duane Harvey, Mr. Ronald Green, Ms. Caroline Karangu, I will always hold
dear the memories of our late night meetings at school working to complete
either assignments or research related projects. I wish you all the very best in
your professional careers.
And to other selected individuals who have impacted my experience in
many positive ways, notably, Dr. Telesphor Kamgaing at Intel Corporation, Dr.
Jules Kouatchou, Dr. Nathan Richardson, Dr. Willie Thompson II, Dr. Michel
Reece, Dr. Carey Johnson, Dr. Robert Johnson, Dr. Otsebele Nare, Mr.
Lawrence Walker, and Mrs. Noukeutchesie (10th grade physics instructor, L.B.A.Cameroon), Thank you !
v
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T a b le o f C o n t e n t s
L is t
of
T a b l e s ............................................................................................................................... ix
L is t
of
F i g u r e s .............................................................................
L is t
o f a b b r e v ia t io n s ............................................................................................................... x v
C h a p te r 1 .............................................................................................................................................. 1
1.1 M o t iv a t io n ...............................................................................................................................1
1.2 P r e v io u s W
o r k ..................................................................................................................... 4
1.3 R e s e a r c h G o a l s A n d C o n t r ib u t io n s .......................................................................11
1.4 O r g a n iz a t io n
of
D is s e r t a t io n ..................................................................................... 12
C h a p te r 2 ............................................................................................................................................14
2.1 I n t r o d u c t io n ........................................................................................................................ 14
2 .2 F u n d a m e n t a l s
of
M ic r o w a v e R a d io m e t r y ............................................................ 15
2 .3 R a d io m e t e r S y s t e m s C o n f ig u r a t io n s .................................................................... 18
2 .4 C o s t C o n s id e r a t io n s ......................................................................................................22
2 .5 S o il M o is t u r e (a n
a p p l ic a t io n ) ....................................................................................2 5
2.5.1
Soil M oisture d e fin e d ..............................................................................................2 5
2 .5 .2
C onventional techniques of estim ating soil m o is tu re ................................ 2 6
2.5.3
Remote sensing retrieval of soil m oisture............................................ 28
C h a p te r 3 ........................................................................................................................................... 3 0
3.1
I n t r o d u c t io n ...................................................................................................................... 3 0
vi
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3.2 S y s t e m D e s c r ip t io n ....................................................................................... 31
3.2.1
Antenna...................................................................................................32
3.2.2 Receiver RF Front-End Unit................................................................. 40
3.2.3 Back-End Processing U n it....................................................................42
3 .3 P a r t s S e l e c t io n ................................................................................................................4 5
3 .4 V o l t a g e R e g u l a t io n U n i t .............................................................................................. 52
3 .5 A n t e n n a P a t t e r n C o r r e c t io n A n a l y s is ................................................................. 5 6
3 .6 C a l ib r a t io n
of
I n s t r u m e n t ........................................................................................... 5 8
3 .7 S u m m a ry .................................................................................................................................6 5
Chapter 4 ...................................................................................................................67
4.1 I n t r o d u c t io n ........................................................................................................................67
4.2 D e s c r ip t io n
of
DSDR S y s t e m ...................................................................... 68
4.3 W o r k O b j e c t iv e s ........................................................................................... 69
4 .4 B a n d -P a s s F il t e r D e s ig n ............................................................................................... 7 0
4.4.1
Filter Design Considerations.................................................................71
4.4.2 Design Methodology............................................................................. 74
4.4.3 Discussion of Measured Results.......................................................... 79
4 .5 L o w -N o is e A m p l if ie r D e s ig n .........................................................................................8 2
4.5.1
Amplifier Design Considerations.......................................................... 83
4.5.2 Circuit Description..................................................................................88
4.5.3 LNA Simulation Results........................................................................ 95
4.5.4
Discussion of Measured Results.......................................................... 98
4 .6 T h r e e - s t a g e R F A m p l if ie r D e s ig n ...........................................................................102
vii
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4.6.1
Design Description.............................................................................. 102
4.6.2
Discussion of Measured Results....................................................... 105
4.7 R e c e iv e r RF F r o n t -E n d In t e g r a t io n
and
E v a l u a t io n ...................................108
4.8 S u m m a r y ...............................................................................................................................113
Chapter 5 ................................................................................................................. 115
5.1 In t r o d u c t io n ..................................................................................................................... 115
5 .2 M e a s u r e m e n t P r o t o c o l D e v e l o p m e n t ................................................................ 116
5 .3 S o il M o is t u r e F ie ld E x p e r im e n t C a m p a ig n s ...................................................... 119
5.3.1
Field Experiments-1 ............................................................................ 119
5.3.2
Field Experiments-2............................................................................ 121
5 .4 FIM R R e s u l t s V a l id a t io n ............................................................................................. 123
5 .5 S u m m a r y ...............................................................................................................................123
Chapter 6 ................................................................................................................. 125
6.1 S u m m a r y
of
R e s e a r c h C o n t r ib u t io n s ..................................................................125
6 .2 F u t u r e R e s e a r c h O p p o r t u n it ie s ............................................................................ 126
Appendix A Photographs of Outreach Activities................................................... 128
Appendix B Vitae..................................................................................................... 130
R e f e r e n c e s ....................................................................................................................................133
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L is t o f T a b l e s
Table 1.1 List of handheld/portable radiometers either published or commercially
available with their respective operating wavelengths......................................6
Table 3.1 List of some commercially available FPGA evaluation boards along
with associated operating power requirements and cost............................... 42
Table 3.2 HMR circuit componenets with corresponding electrical properties....46
Table 3.3 List of voltage regulation unit components............................................ 53
Table 3.4 Extracted noise temperature of calibration reference loads.................64
Table 4.1 BPF specifications....................................................................................75
Table 4.2 LNA specifications................................................................................... 88
Table 4.3 LNA parts lists.......................................................................................... 95
Table 4.4 RF Amplifier specifications.................................................................... 103
Table 5.1 Handheld microwave radiometer data-logger sheet........................... 118
Table 5.2 Benchmarking of handheld radiometer microwave radiation outputs
against recently published results.................................................................. 123
ix
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L is t o f F ig u r e s
Figure 1.1 Photo of the IL1400 portable radiometer (Courtesy o f International
Light Inc.)......................................................................................................................7
Figure 1.2 Photograph of the Multispectral handheld radiometer MSR5 (Courtesy
ofCropscan Inc.)..................................................................................................8
Figure 1.3 Partitioned hardware illustration of HOPE demining system.............. 10
Figure 2.1 Graphical comparison of Planck’s relation of Blackbody theory and
the Raleigh-Jeans approximation for T=300K. Close agreement between
the two expressions is observed at frequencies below 500 GHz...................16
Figure 2.2 Block diagram of total power radiometer.............................................. 18
Figure 2.3 Block diagram of dicke radiometer....................................................... 20
Figure 3.1 Block diagram architecture of HMR...................................................... 31
Figure 3.2 Representation of a inset-line fed microstrip patch antenna.............33
Figure 3.3 Layout view of stacked patch antenna design and associated
dimensions [32].................................................................................................. 36
Figure 3.4 Representation of antenna horn sidewall design with notched edges.
The side panels are shown with the seams detached and flattened out. All
dimensions are in mm [32]................................................................................ 37
Figure 3.5 Cross-sectional representation showing patch, aperture, bucket and
microstrip assembly [32]....................................................................................38
Figure 3.6 Measured antenna pattern at varying scan angles............................. 39
x
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Figure 3.7 Block diagram of RF receiver front-end................................................41
Figure 3.8 Schematic of implemented analog Back-End processing unit
highlighting the four major building blocks of the circuitry..............................44
Figure 3.9 Comparison of loss due to dielectric in a microstrip line on different
substrate materials (courtesy of Rogers corporation).....................................49
Figure 3.10 Photos of integrated low-cost radiometer printed circuit board
showing electronic compartment on front side of the PCB and stacked patch
microstrip antenna on back side of the board................................................. 50
Figure 3.11 Representation of low-cost radiometer assembly. (a)-View of
instrument engineering prototype (b)-Close view of the front-panel displaying
specifications and information on operating the instrument........................... 51
Figure 3.12 Ouput responses characteristics of high-linearity, switching ±7V
voltage regulators, (a) Voltage output of PT78ST107V and (b) Output
voltage for PT78NR107V.................................................................................. 55
Figure 3.13 Block diagram of typical radiometer receiver calibration setup........ 59
Figure 3.14 ‘load’ and ‘receiver’ reflection parameters: (a) Real and Imaginary
parts of the reflection coefficients, (b) Magnitude of the reflection
coefficients..........................................................................................................63
Figure 3.15 HMR calibration curves; dashed line represents the case assuming
matched conditions, solid line denotes instrument calibration response
accounting for additional noise temperature contribution due to impedance
mismatch............................................................................................................ 65
Figure 4.1 Block diagram of direct-sampling digital radiometer system [38].......68
xi
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Figure 4.2 Design topologies for filters; (a) Microstrip Coupled-lines structure, (b)
Grounded XI4 Microstrip SIR structure............................................................73
Figure 4.3 Layout of filter in microstrip form........................................................... 75
Figure 4.4 Equivalent circuit schematic.................................................................. 76
Figure 4.5 Effects of Cgap on filter resonant frequency.......................................... 78
Figure 4.6 Photo of fabricated filter.........................................................................79
Figure 4.7 Wideband filter performance results..................................................... 80
Figure 4.8 Narrowband output match results......................................................... 81
Figure 4.9 Typical block diagram of a microwave transistor amplifier................. 83
Figure 4.10 Typical stabilization methods.............................................................. 85
Figure 4.11 Single-stub matching of two arbitrary impedances........................... 89
Figure 4.12 Gain and Noise circles of ATF-34143 at 1.413-GHz.........................90
Figure 4.13 Schematic of amplifier circuit in microstrip form................................ 94
Figure 4.14 Wideband simulation of LNA S-parameters performance. Device
exhibits large gain along with adequate input at output return loss
performance across bandwidth of design from 1400MHz to 1427MHz. AG
across design bandwidth is determined to be less than 0.2dB......................96
Figure 4.15 Simplified schematic of LNA noise figure performance. Simulated
noise figure performance across LNA bandwidth of interest is close to
desired value of 0.5dB.......................................................................................97
Figure 4.16 Simulated LNA stability factor performance. K-factor is greater than
unity across wideband simulation frequency................................................... 97
Figure 4.17 Photo of fabricated Low-noise amplifier............................................. 99
xii
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Figure 4.18 Measured effective Gain of fabricated Low-noise amplifier. LNA
displays similar characteristics at both input power with a measured gain
value of around 13.8dB at center design frequency of 1.413GHz.................99
Figure 4.19 Microstrip Layout of 3-stages RF amplifier. Shown of the figure are
the grounding walls used to prevent coupling effects resulting from close
proximity of some stub lines............................................................................104
Figure 4.20 S-parameters characteristics of the multi-stage amplifier as
predicted by ADS circuit simulator.................................................................106
Figure 4.21 Simplified schematic of VSWR. It is shown that across bandwidth of
interest (1400MHz - 1427MHz), the circuit displays adequate input and
output matching conditions............................................................................. 106
Figure 4.22 Simulated stability performance of the multi-stage amplifier.......... 107
Figure 4.23 Simulated amplifier performance at 1dB compression point. It is
shown that at compression, the circuit emulates almost 51 dB of gain.
Compression condition occurs at input power level greater or equal to 31.5dBm............................................................................................................107
Figure 4.24 Flow chart of methodology of Receiver Front-End integration....... 109
Figure 4.25 Simplified integrated Front-end circuit representation. The various
sections of the receivers are contained within sub-circuit blocks. The tuning
interconnecting lines are not shown............................................................... 110
Figure 4.26 Representation of integrated receiver Front-end design layout. The
components in the circuits are arranged in a configuration that optimizes real
estate use of the printed circuit board material. Also shown in the figure are
xiii
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the grounding walls used to isolate various sections of the design in
preventing potential coupling effects.............................................................. 111
Figure 4.27 S-parameters characteristics of integrated receiver Front-End as
predicted by ADS circuit simulator..................................................................112
Figure 4.28 Simulated noise figure performance of integrated receiver FrontEnd.................................................................................................................... 113
Figure 5.1 representation of the L-band handheld radiometer measured
brightness T and “in situ" soil moisture at 5 cm depth. The two datasets
display a linear relationship.............................................................................121
xiv
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L is t o f a b b r e v ia t io n s
A/D
Analog to Digital
BJT
Bipolar Junction Transistor
BPF
Bandpass Filter
CMOS
Complementary Metal Oxide Semiconductor
CAMRA
Center of Advanced Microwave Research and
Applications
COBE
Cosmic Background Explorer
DSDR
Direct Sampling Digital Radiometer
DSP
Digital Signal Processing
FET
Field-effect Transistor
FPAA
Field Programmable Analog Arrays
FPGA
Field Programmable Gate Arrays
GLOBE
Global Learning and Observations to Benefit
the Environment
GPR
Ground Penetrating Radar
GPS
Global Positioning System
GSFC
Goddard Space Flight Center
HPBW
Half power beamwidth
HEMT
High Electron Mobility Transistor
HMR
Handheld Microwave Radiometer
IMN, OMN
Input/Output Matching Network
xv
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IR
Infrared
LNA
Low Noise Amplifier
LSB, MSB
Least/Most Significant bit
MD
Metal Detector
MIC
Microwave Integrated Circuit
MMIC
Monolithic Microwave Integrated Circuit
MEMS
Micro-electromechanical Systems
NASA
National Aeronautic and Space Administration
NEAT
Noise Equivalent Temperature Difference
PALS
Passive/Active L-and-S band radiometer
PBMR
Pushbroom Microwave Radiometer
RF-FE
Radio-Frequency receiver Front-End
RFI
Radio-Frequency Interference
SISO
Single Input Single Output
SM
Soil Moisture
SMT
Surface Mount Technology
SSM/I
Special Sensor Microwave/Imager
SPDT
Single Pole Double Throw
TRM
TOPEX/Poseidon Microwave Radiometer
UV
Ultraviolet
xvi
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C hapter 1
I n t r o d u c t io n
1.1 M o t iv a t io n
“M icrow ave radiometers have changed the w ay we view the Earth. If only
we could get people to understand th e m r
Dr. Dorothy Zukor
Former Deputy Director,
Earth-Sun Exploration Division, NASA-GSFC
The ongoing pursuit of the National Aeronautics and Space Administration
(NASA) towards the development of more stable and more accurate radiometer
systems is a justification of these instruments usefulness in understanding our
home planet and its neighboring cosmos.
Microwave radiometers are highly
1
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sophisticated engineering devices capable of discriminating between changes in
a given object physical temperature and its surrounding background noise. With
the major recent advancements in RF technology and signal processing
techniques, the level of accuracy or sensitivity at which this discrimination is
made has significantly improved.
Today’s radiometer systems have proven
effective and essential at extracting geophysical parameters related to natural
processes (i.e. Ozone depletion, meteorological, hydrological and etc. ) affecting
all forms of life on Earth.
In the case of the hydrology process, microwave
radiometers are useful in disseminating information regarding the state of water
content within various soil layers. This information is commonly referred to as
soil moisture (SM) - A parameter directly affecting energy transfer fluxes between
ground and the atmosphere primarily by means of vegetation evapotranspiration
and land surfaces dry-down.
There is convincing evidence that regular
monitoring of SM can provide scientists with valuable insights in understanding
climate variability thus, enabling them to perform accurate weather forecasting
[1].
To this end, NASA and several other supporting institutions and space
agencies worldwide namely, European Space Agency (ESA) have invested
significant amount of resources towards the development of radiometer
instruments to study SM of land surfaces.
Prior research work done by many authors concluded that radiometric
observations at long microwave wavelengths are best suited for SM applications
[2]-[4].
This analogy stems from the greater penetration depth of the
electromagnetic wave associated with low frequencies, consequently enabling
2
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observation of large depth of soil.
Furthermore, in space-based radiometer
systems, as is the case for many of NASA’s radiometers, the low frequency
operation offers the added advantage of penetrating the earth’s ionosphere layer
with very little atmospheric attenuation.
In an article dating 2002, Sean O’Keefe (former NASA administrator)
expressed concerns over the future of NASA science and technology workforce
before the Congressional Science Committee [5]. This article states that NASA
“is finding it particularly difficult to hire people with engineering, science, and
information technology skills.” [5],
In the engineering discipline, particularly in
radiometry fields of concentration, a major contributor to this deficiency within the
pipeline is the inaccessibility of remote sensing instrumentation and techniques
to many high-education engineering institutions due to the high related costs. In
fact, remote sensing instrumentation remains at best a rare instance for many
educational institutions around the country due to high associated cost and
general inaccessibility of the science.
A study conducted by the Virginia Space Grant Consortium argued that
engaging students early in practical research projects provides an effective
means of stimulating continued interest in scientific research [6]. So it can be
deduced that accessibility to affordable yet, reliable scientific instrumentation
could potentially draw more talent within the remote sensing arena, ultimately
addressing NASA’s pipeline deficiency concerns.
3
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1.2 P r e v io u s W o r k
Radio astronomy found its debut in the mid 1900’s with the introduction of
the Dicke radiometer [7].
In 1972, the successful deployment of the first
LANDSAT in a series of seven, Landsat-1, into orbit by NASA brought for the first
time revolutionary imaging of the Earth from space. The first three LANDSATs
(Landsat-1, 2, 3) consisted essentially of spectral sensors operating within the
infrared (IR), near infrared (NIR) and visible frequency spectrum. Outputs from
these
satellite
advancement
based
in
instruments
mapping,
generated
agriculture,
numerous
planning
and
resources
geology
for
research.
Achievements by these earlier LANDSATs sensors confirmed the benefits of
space radiometry as a means of observing and studying the Earth. Since then,
many other satellite-platform instruments have been launched to either study the
Earth while orbiting it or, to observe into deep space in an effort to understand
the evolution of Earth from the “big bang” phenomenon and the continuous
expansion of the universe as an ongoing effect of this phenomenon.
Some
examples of earth orbiting sensors are: Topex/Poseidon Microwave Radiometer
(TMR) [8], the Special Sensor Microwave/Imager (SSM/I) [9] and Conical
Scanning Microwave Imager Sounder (CMIS) [10].
One example of space
pointing instruments is the Cosmic Background Explorer (COBE) [11].
Microwave radiometers are particularly useful in environmental studies. In
addition to microwave wavelengths abilities to penetrate through media that are
typically opaque at optical and infrared wavelengths, there are several other
motivations for using microwave sensors.
First, at microwave frequencies, the
4
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atmosphere is effectively transparent, providing all-weather coverage. Secondly,
vegetation is semi-transparent, allowing the observation of underlying surfaces.
Third, the microwave measurements do not require solar illumination, permitting
both day and night observations. In addition to satellite platform sensors, there
exist many other types of microwave radiometers including ground-based and
aircraft mounted instruments currently in use in a variety of research applications.
Handheld microwave radiometers can be perceived as portable hardware
instruments capable of reliable scientific functionalities. This compactness and
portability features of handheld microwave sensors in contrast to their groundbased counterparts could potentially lend themselves to various practical
considerations including: 1) Ease of deployment to virtually any remote test sites
otherwise inaccessible to large assemblies mounted on towers or carried by
trucks. 2) Accurate understanding of various ground variables such as biomass,
amount of ground cover in localized areas within fine resolution elements
thereby, leading to proper interpretation of satellite data.
Such information is
best attained by continuous monitoring of small test sites. Handheld radiometers
are well suited for this type of studies.
3) The large footprint resolutions of
satellite and aircraft mounted systems often necessitate measures in-place for
validating the retrieved data. These validation procedures commonly referred to
as “Ground-truth” can be well performed using portable microwave radiometers.
Over the recent years, some handheld sensors have been reported. While the
reported devices fulfill the portability requirement, none of them however has
demonstrated capability for remote sensing of soil moisture at low microwave
5
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frequencies.
T a b le-1.1
lists fe w o f th ese sensors along with their intended
scientific purpose.
T a b l e -1 .1 :
L is t
of
h andheld/ po rtable
r a d io m e t e r
e it h e r
p u b l is h e d
or
c o m m e r c ia l l y
AVAILABLE WITH RESPECTIVE OPERATING WAVELENGTHS.
International
Light Inc.
[12]
IL1400 Radiometer
& Photometer
UV
(400 to 700)-nm
Photometry,
UV curing
$1,495
Cropscan Inc.
[13]
Multispectral
Radiometer (MSR)
NIR
(750 to 900)-nm
Biochemical
content of
plants
N/A
European Space
Agency (ESA)
[14]
Handheld
Operational
Demining System
(HOPE)
Department of
Defense (DoD)
[15]
Hand-held
microwave
radiometer for
landmine
CAMRA/NASA
(This work)
Handheld
Microwave
Radiometer (HMR)
L, X Bands
Land-mine
detection
$14,000
X, Ku Bands
Land mine
detection
N/A
L-Band
surface soil
moisture
Measurements
6
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$200
The IL1400 photometer, depicted in Fig 1.1, is a handheld radiometer designed
for ultra violet (UV) curing and other exposure applications [12]. This device is
capable of measuring plants’ irradiations in a wide spectral range (250nm to
475nm) owing it to a combination of built-in light detectors.
The IL1400
automatically read the type and calibration information from each detector
therefore making substituting the detectors or using them in a given combination
convenient and effective.
Figure 1.1 - Photo of the IL1400 portable radiometer (courtesy o f InternationalLight Inc.)
7
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Figure 1.2 shows photograph of the Multispectral Radiometer (MSR5). It is
a five channel handheld sensor designed to measure incident and reflected
irradiation of vegetation canopies.
This instrument consists of a telescoping
support pole, analog to digital (A/D) converter, connecting cables, data logger
and operating software. The five channels are centered about the spectrum of
the five channel Landsat Thermatic Mapper (TM5) sensor with wavelengths from
450 nm to 1720 nm [13]. In the field, the radiometer is upheld above the canopy
covered surface at a fixed height through the use of the support pole.
resolution footprint element is taken to be equal to half the height.
The
Instrument
comes with an external calibration plate for use in calibrating the instrument prior
to field measurements. This portable device measures 80 x 80 x 100mm in size
and is operated using rechargeable 10V NiMH battery.
Figure 1.2 - Photograph of the Multispectral handheld radiometer
MSR5 (courtesy o f Cropsan Inc.)
8
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Figure 1.3 shows photographs of the Handheld Operational Demining
System (HOPE). The HOPE handheld system is part of a project funded by the
European Commission targeting the detection of hard to detect land mines [14].
The HOPE instrument includes a metal detector, stepped frequency ground
penetrating radar (GPR) and a multifrequency microwave radiometer (MWR), all
with imaging capabilities through the use of a high precision positioning system.
The instrument’s antenna which uses metal detector (MD) coils, is designed to
operate in the very near field for a ground resolution less than 10cm with a
system sensitivity of 1 kelvin. HOPE features operation at 1.7GHz (L-Band) and
7GHz (X-Band) and relies on the principle that the presence of a mine causes a
temperature anomaly at the surface of the host medium of uniform emissivity or
in other words, the soil disturbance causes a brightness temperature change.
9
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OPMS camera
MWR sky antenna
OPMS laptop
Antenna housing for
GPR and MWR
MD coils
MWR and
Top: HOPE head (a) and
nsor electronics
and power supply
(c) MWR backpack
bottom: GPR electronics box (b)
Figure 1.3- Partitioned hardware illustration of HOPE demining system
The handheld standoff mine detector system (HSTAMIDS) described in
[15] is part of an effort undertaken by the US Army to produce a practical
successor to currently prevalent MD. The HSTAMIDS is in many ways similar to
the HOPE instrument described above; it consists of multisensors integrated with
standard M D and G PR .
The G P R uses a frequency stepped technology
covering frequencies from 900-2750 MHz. The overall weight of the instrument
10
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including a belt attached electronics unit, an IR lightweight camera and the GPR
is approximately 10Lbs.
1.3 R e s e a r c h G o a l s A nd C o n t r ib u t io n s
The major objective of this research endeavor is the development of lowcost microwave instrumentation for remote sensing application of soil moisture.
The contributions of the research presented in this dissertation are two fold.
First, the development of a revolutionary low-cost handheld
microwave
radiometer suitable for soil moisture studies is reported. The instrument herein is
to-date, the most affordable and convenient (in terms of portability) microwave
hardware fully capable of surface soil moisture studies.
Second, the design
methodology and comprehensive evaluation of an integrated RF receiver frontend is outlined.
The first part of the research is expected to alleviate the challenges faced
by NASA regarding its workforce deficiency by potentially serving as an
education-outreach device.
The latter part of the research was performed in
support to efforts recently started by the Center of Microwave and RF
Engineering (COMSARE) at Morgan State University to develop the university’s
first Direct Sampling Digital Radiometer (DSDR). Below is the summary of goals
proposed for this research:
11
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•
Development of a down-converting Dicke handheld radiometer featuring
the following figures of merit:
•
1.
Center frequency: 1.413 GHz
2.
Sampling bandwidth:
3.
Sensitivity >0.4 °K
4.
Overall cost: < $200
25 MHz
Demonstration and validation of the instrument against gravimetric soil
moisture data.
•
Measurement
protocol
documentation
for using
the
instrument
in
classroom environment.
•
Design and evaluation of an integrated RF receiver front-end for a direct
sampling digital radiometer containing following components:
1.
Low-noise amplifier
2.
Bandpass filters
3.
RF gain amplifier
1.4 O r g a n iz a t io n
of
D is s e r t a t io n
The contents of the dissertation are organized in this report in sequential
order.
Chapter 2 of the document discusses background work relevant to
microwave radiometry fundamentals. It also looks at various radiometer systems
designs from a cost perspective.
Soil moisture is then introduced and its
12
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relevance to the frame of this work is highlighted.
Chapter 3 presents a
comprehensive description of the low-cost handheld microwave radiometer
instrument developed. All major building blocks of the systems are discussed.
An examination of antenna pattern correction analysis and that of the instrument
calibration procedure are also included in this section.
Chapter 4 reviews the
design methodology for the development of the integrated RF front-end building
blocks consisting of bandpass filter, low-noise amplifier and RF gain amplifier. A
full characterization followed by measurement results for each component is
discussed.
In Chapter 5, soil moisture measurement data from series of field
experiments and outreach activities are discussed. A measurement protocol for
using the instrument is also presented.
Finally, Chapter 6 summarizes the
dissertation and makes suggestions for further research efforts.
13
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C hapter 2
R a d io m e t e r S y s t e m s
2.1 I n t r o d u c t io n
The primary focus of this dissertation as previously described in Chapter1, is the development of a low-cost microwave remote sensing instrument
capable of soil moisture observations.
This chapter presents a background
discussion of concepts and considerations pertaining to radiometer systems.
The chapter begins with underlying fundamentals and theories governing
microwave radiometry.
Then, it introduces various system configurations and
attaches qualitative and quantitative figure of merits for each type.
Next, sub­
components or units with greater impact to overall system cost and performance
are detailed.
In the later section of the chapter, soil moisture as relates to this
14
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research is introduced.
Various methods commonly used for estimating this
parameter are also discussed.
2.2 F u n d a m e n t a l s
of
M ic r o w a v e R a d io m e t r y
A major discovery by Max Planck during the first half of 20th century
revealed that all matter naturally emits electromagnetic energy.
This emitted
energy is proportional to frequency and temperature state of the matter.
His
theory was formulated as the well-known Planck’s law of blackbody which is
governed by the relation
*(/)=
2h f 3 (
1
exp(/z / / kT) - 1
eq (2.1)
where B ( f ) is the spectral brightness in tym~2sr~lHz~xJ, h is the Plank constant
(=6.626 xl0~34[Js]), T represents brightness temperature expressed in degrees
Kelvin, and k is the Boltzmann constant(=1.38* 10“23[ J AT1]).
Brightness temperature T
is the physical quantity radiometers are
essentially designed to determine. The higher the temperature, the more energy
is emitted according to (2.1).
We consider the case when h f / k T « 1, then
exp(/j f / k T ) - l &h f / kT leading to:
15
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2f2
2
B ( f ) = ^ L - k T = — kT
c
eq (2.2)
A
where A = d f is the free-space wavelength.
Expression (2.2) is known as the Raleigh-Jeans law.
The condition
h f / k T « 1 resulting in the above simplification is easy to satisfy at low
frequency values and ambient temperatures as shown in fig. 2.1.
Fixing
r = 300°A: in the graphical representation, we determine the relative error
between (2.1) and (2.2) to be less than 0.9% a t/ = 100GHz.
1.E-07
1.E-08
1.E-09
1.E-10
Z'
1.E-11
N
I
1.E-12
3L
1.E-13
%
1.E-14
cjT
1.E-15
^
1.E-16
§
1.E-17
0Q
1.E-18
Planck's relation
Raleigh-Jeans
approximation
1.E-19
1.E-20
1.E-21
1.E+08
1.E+09
1.E+10
1.E+11
1.E+12
1.E+13
1.E+14
1.E+15
Freq [Hz]
Figure 2.1
Graphical comparison of Planck’s relation of blackbody theory and the Raleigh-Jeans
approximation for T=300°K. Close agreement between the two expressions is
observed at frequencies below 500 GHz.
16
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Because it gives a linear relationship between spectral brightness and
temperature at a given frequency, Raleigh-Jeans approximation has been widely
adopted in radiometry for Earth remote sensing applications since frequencies
suited for these studies are generally less than 100 GHz.
Theoretically,
microwave
radiometers can be conceived
as highly
sensitive receivers that measure thermal emission or incoherent noise signal
radiating from surrounding objects.
material’s
electromagnetic
This emission is also dependent upon
properties,
i.e.
permeability,
conductivity
and
permittivity. Together, these properties effectively make up the “emissivity, e, of
the material, e is a quantity with values ranging between zero and unity. With
unity being that of a blackbody or perfect emitter.
An ideal metal has an
emissivity of zero. For a given object, emitted brightness is related to e through
where 7 ^ represents the physical or thermometric temperature of the object.
The subscript e is added to T to help differentiate between physical and
brightness temperature.
17
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2 .3 R a d io m e t e r S y s t e m s C o n f ig u r a t io n s
The general and most elementary radiometer type is the total power
radiometer shown in fig. 2.2.
Mixer
Pre-Amplifier
Integrator
Amplifier
IF-Filter
Detector
Figure 2.2 - Block diagram of total power radiometer
It consists of an antenna, a mixer in combination with a local oscillator, several
amplification stages, filtering blocks, a detector and post-detection processing
unit. The local oscillator serves the purpose of converting the incoming collected
antenna signal to baseband or intermediate frequency (IF) while retaining its RF
power level unchanged. This system optionally also includes a pre-amplification
stage serving the purpose of reducing overall system noise factor as will be
shown later.
In theory, the pre-detection power measurable at detector’s input
terminal for this system is dictated by the expression [16]:
PPr e - ^ = k B G T sys
eq (2.4)
where B represents system radiometric bandwidth (Hz) ideally set by the
filtering blocks, G .denotes overall gain of all amplification stages, Tsys = T A + Trec
18
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represents system noise temperature which incorporates both, antenna sensed
brightness temperature and receiver self-noise temperature.
The detection stage is commonly realized using a diode detector with
square law behavior.
This mode of operation enforces a direct relationship
between input and output powers across the detector. The mean voltage output
measurable at diode output terminal is theoretically given by
eq (2.5)
where a represents the diode sensitivity constant.
In reality however, the detector’s output signal is merely the average of the
instantaneous voltage observed by the antenna. Obtaining the true mean of the
signal requires further filtering and smoothing of the signal waveform.
This is
commonly achieved using an integrating circuit.
The radiometric sensitivity, AT in kelvin, of a radiometer represents the
smallest change in emitted microwave radiation that can be sensed by its
electronics assembly. For the total power radiometer it is expressed as [16]:
1/2
eq (2.6a)
where B t is the product bandwidth integration time constant, the term (AGI G)
denotes gain fluctuations present within the system.
Because it is desired for
AT to be of low values, the quantity AG should be minimized to a point it can be
19
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ignored as for instance ( A G / G ) « 1 0 4. This however depends on the quality of
electronic parts used in realizing the system.
One method often adopted in
practice to reduce effects of gain variations is the use of long integration
periods, t . In the absence of gain fluctuations, expression (2.6a) reduces to
AT=- A t l l ec
v Bt
eq(2.6b)
Design requirements for imaging radiometers however, often do not
accommodate large integration periods since long r entails that the radiometer
antenna’s main-beam must remain focus toward the scene over an extended
duration.
Alone, gain fluctuations represent the greatest limitation of the total
gain radiometer.
The Dicke radiometer, illustrated in Fig. 2.3, addresses the gain variations
shortfalls of the total power radiometer by adding an RF switch, known as a
Dicke switch inserted at antenna’s output terminal, and a synchronous detector
or demodulator located at detector’s output port.
Square
Wave
RF
Switch
if
Mixer
Integrator
Amplifier
ANT
K
Pre-Amplifier
Local
Oscillator
Synchronous
detector
T ref
D
:
V qut
Detector
IF-Filter
. (-300° K)
Figure 2.3 - Block diagram of Dicke radiometer
20
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The RF switch, often a multi-state switch, allows the receiver to
periodically rotate between antenna and a known reference load. This RF switch
operates synchronously with the demodulator via a square-wave control signal.
Considering the case of a single pole double throw (SPDT) switch, and taking
dwelling time at each state to be equaled, the measurable voltage at detector’s
output of dicke radiometer is expressed as [16]:
V„ = ~ a k B G [(T ,
+ r,J -(r„ + r j ]
eq (2.7a)
where Tret represents the reference load temperature. Simplification of (2.7a)
yields
Vd = ^ a k B G { T t - T „ , )
The term
eq (2.7b)
refers to the half dwelling period by the receiver to each one of the
two terminals of the RF switch assuming a perfect 50% duty cycle square wave
control signal. While greatly reducing the complexity of accurately determining
antenna temperature 7^when load reference temperature is known, the Dicke
radiometer suffers from a reduced radiometric sensitivity stemming from lower
antenna observation dwelling period.
21
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1/2
„„
\ l( T A+TrJ + 2 (t, . , + tJ
A T = ----------------------------Bt
eq (2.8)
+
In treating the expression above, if we assume Tref =TA, the sensitivity of the
Dicke radiometer only amount to half that of the ideal total power radiometer as
given in (2.6b). However, when other losses in the system are considered, the
actual sensitivity further reduces.
When satisfying the condition Tref =TA, the
radiometer is known to be balanced.
There exist other types of radiometers. However, these often incorporate
principles of both the total power and dicke radiometers.
Some examples of
other types of instruments include the feedback gain control radiometer [17] and
the noise injection radiometer [18] to enumerate a few.
In general these
instruments employ more reference loads (i.e. noise diodes) in their operation
scheme as to yield higher accuracy in determining instrument’s response
characteristics.
Doing so however, results in direct cost implications on
instrument development budget as it requires the purchase of additional
components.
2 .4 C o s t C o n s id e ra tio n s
In principle, development costs for radiometer systems depend upon the
science mission to fulfill combined with the very type of instrumentation.
22
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Associated payload launching costs in addition to rigorous flight testing render
development of satellite-platform based instruments highly resources exhaustive.
Furthermore, space instrumentation requires use of radiation resistant, so-called
“rad-hard” electronics to endure harsh environmental conditions in space. These
components can amount to several thousand dollars as they are custom
designed specifically for such applications. On the other hand, budgeting factors
for Earth-bound instruments mainly incorporate acquisition costs for hardware
and assembly.
Also, electronics housing of these systems does not require
exotic features such as hermetic packaging in contrast to space instrumentation
[19]. Moreover, with recent advances in the semiconductor technology sector,
high-performance hardware compatible with typical radiometers requirements
can often be acquired commercially. In general, major costs driving components
associated with development of total power L-band ground-based systems
include antenna, amplifiers, mixer, filters and detector.
For Dicke and noise-
injection configurations, the list is expanded to include RF-switches, couplers and
also noise diodes used as additional reference loads.
It thus appears that
hardware acquisition alone for radiometers can be quite costly.
Among other considerations with notable implications on development
budget for radiometer instrumentation include: 1) Need for thermal control to
maintain adequate stability of the system and 2) Addressing the issue of radio
frequency interference (RFI). While efficient internal calibration is paramount in
realizing a stable system, the importance of a thermal control unit to regulate
front-end electronics assembly temperature cannot be overstated.
23
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For most
systems, temperature regulation is realized by housing all RF parts in an
enclosure equipped with fans automatically powered to force internal and
external air exchanges thereby, regulating temperature within the enclosure at a
preset value [20].
Radio frequency interference (RFI) is a phenomenon which occurs in
radiometer systems under the form of uncontrolled inter-modulation products
from man-made sources surrounding the instrument’s operating frequency band.
RFI can severely impair the operation of radiometers.
L-Band sensors are
especially susceptible to RFI due to the proliferation of radars systems and
communications links at this portion of the microwave spectrum. For example,
the passive/active L-and-S band (PALS) radiometer experienced strong levels of
RFI contamination, primarily within its L-band channels, during the 1999
Southern Great Plains experiment in Oklahoma [21]. Also, recorded data by the
L-band Pushbroom Microwave Radiometer (PBMR) also displayed significant
levels of RFI infiltration during experiments conducted at the USDA Walnut Gulch
watershed [22].
To date, there exist no comprehensive means to fully mitigate RFI without
compromising continuity in the recorded microwave data. Commonly, adopted
strategies involve the use of multiple channels as opposed to a single channel to
enable detection of RFI by means of comparing captured information from the
various channels [23].
Other systems utilize finely tuned narrowband filters
placed in front of the first stage of amplifications [20]-[22],
Digital receiver
24
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technology provides another approach of achieving improved RFI resolution both
in time and frequency [24].
2 .5 S o il M o is tu re (an a p p lic a tio n )
2 .5 .1
S o il M o is t u r e
d e f in e d
Soil moisture (SM) can be quantitatively defined as the amount of water
held between soil particles in the unsaturated layer of Earth’s surface.
critical
component
of
many
geo-systems
affecting
climate
It is a
interactions.
Knowledge of near-surface soil moisture, within 3cm to 5cm of surface, is
important in prediction of hydrology processes and global natural resources
budgeting [2-4].
SM is a quantity detectable by microwave radiometers.
Explanation for this fact lies within the electrical conductivity properties of water.
Work by Hallikainen et al. [25] showed that there is a large contrast in dielectric
constant between water and solids with values ranging from k = 1 (air), 3 < k < 7
(dry soil) and k * 80 (free-water). The ability to retrieve SM content information
by means of remote sensing is greatly owed to this characteristic.
25
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2.5.2 C o n v e n tio n a l te c h n iq u e s o f e s tim a tin g s o il m o is tu re
There exist several traditional techniques used for point measurement of
SM.
The oldest, however most accurate, is gravimetric measurement which
essentially consists of deriving volumetric SM content through computation of
mass differences for a given soil sample.
The method generally involves
collecting a sample from the ground at a particular depth and analyzing it in a
laboratory environment where it is weighted, oven dried for a period of time then,
weighted again [26].
The difference between the two masses represents the
water content. The gravimetric measurement technique, while reliable, is very
intensive, resource- and time-consuming.
Neutron thermalization, also known as neutron probe, takes advantage of
the scientifically known fact that rapid changes in soil hydrogen molecules
content are almost completely due to changes in soil water content.
As a
consequence, emitted neutrons from a radioactive source are thermalized, or
slowed by the hydrogen atoms contained within the soil by repeated collision with
the nuclei of soil materials. Water content can be derived from count of slow
neutrons resulting from this attenuation [27]. Although useful for measuring large
soil volume and allowing for scanning at higher soil depths, this approach suffers
from several limitations. First, it is insensitive to near soil surface because of the
sphere of influence emanating from the neutron source, the size of which varies
with soil water content.
Second, the use of radioactive source presents a
radiation hazard thus, making it impractical.
26
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A more recent approach used for point measurement of SM is timedomain reflectometry, TDR, principally based on soil capacitance and soil
dielectric constant as a measure of SM. TDR measures the propagating velocity
of a fast-rise electromagnetic pulse traveling along the sensor [28]. In this case,
the propagation constant of the soil medium, K a, can be derived from analysis of
the pulse dissipation (reflection) as a function of the probe-length (L) and travel
time (t) according to the relation
K a=
where
c
(ctV
v2 L ;
eq (2.9)
« 3 x l0 8/w5_1 is speed of light.
The soil’s relative permittivity is related to its electric susceptibility,^, and
electrical conductivity, <y, through
£r =b + Z e ) - j
G)£n
'0
= e'r - j s ' r
eq(2.10)
where co is the angular velocity at frequency of interest, f 0.is free-space
permittivity.
To derive the electric properties of the scene under observation
such as sr , one must study the behavior of the electromagnetic wave incident on
that surface. The general time-domain form of a homogeneous plane wave for
the electric field in the z direction is given by:
27
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E ( z , t ) = E 0 exp
where n = n ' - j n
-J
I
271
\
T
/
eq (2.11)
-n z - c o t
is the real and imaginary parts of the index of refraction and
are related to complex s r through
n =
1/2
I
. 2
' r + ^ r
+ * r
eq (2.12a)
/2
1/2
n =
- ^ s r2 +£^
/2
eq (2.12b)
TDR, although generally accurate for most soils, does not adapt well to soils with
high water retention capability such as clay.
2.5.3 R e m o te s e n s in g r e t r i e v a l o f s o il m o is tu re
The ability to remotely retrieve SM information is largely owed to the wide
contrast in permittivity between water (£r »80) and soil (£r «3).
Compared to
conventional approaches, this procedure offers a dual advantage. First, it allows
for collection of soil information that is spatially distributed, and more importantly,
the observations are performed rapidly.
community that observation
It is a consensus within the scientific
at L-Band is best suited for soil moisture
measurements owing it to the deeper penetration of the surface by long
wavelengths. However, difficulties arise in the interpretation of the captured data
and retrieving SM information. This is partially due to the influence of the soil
28
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surface hydraulic properties on the microwave emission. This effect has been
determined to be particularly large for structured soils with dense pores
arrangement [29]. Work by Schmugge [30] successfully related the emissivity of
soil surface to the fresnel relation for an electromagnetic wave incident on it as
follow:
eq (2.13)
where R(a) represents the reflectivity of the soil surface. This quantity is
polarization dependent.
In this same work, Schmugge demonstrated that
horizontal polarization yields greater accuracy in passive microwave radiometry
observations.
The general relation of R(a) for horizontally polarized wave is
given by:
f
R ( H ,a ) =
I—
V
c o sa -^s -sin (a)
eq (2.14)
cosa + ^ £-sin2(a)
where e is the complex dielectric constant of the medium, and a gives the
instrument’s antenna angle of incidence. It is possible to derive brightness TB for
known esurf values and effective soil surface temperatures according to (2.3).
29
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C hapter 3
H a n d h e l d M ic r o w a v e R a d io m e t e r
3.1 In tro d u c tio n
In Chapter 2, a comprehensive background review of radiometer systems
was presented.
An examination of major integrant building blocks typical of
these systems along with quantitative figure of merits necessary to fulfill their
science requirements was provided.
detailed
characterization
of
a
This chapter presents the design and
low-cost
microwave
handheld
radiometer
developed under this research effort. Included in the description of this system
are the methodology for parts selection, a detailed look at the building blocks of
the developed instrument and presentation of its assembly.
antenna pattern correction analysis is also provided.
Discussion of
In the latter part of the
30
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chapter, the discussion is extended to include a study of the calibration
procedure conducted on the instrument.
3 .2 S y s t e m D e s c r ip t io n
Figure 3.1 shows the architectural block diagram of the implemented
handheld radiometer. As shown, the instrument uses a conventional super­
heterodyne receiver configuration. The depicted system architecture consists of
three major sections as reported in [31] including: The antenna section, receiver
RF
Front-End
unit and
Back-End
signal
processing
unit.
Two
major
specifications for the design included first, low-cost requirements which restricted
the choice of parts used to realize the instrument. The second specification was
low-power dissipation stemming from the handheld aspect of the instrument
which; ought to be powered using a battery. Consequently, it is paramount for all
electronics of the radiometer to dissipate as little power as possible to yield an
extended battery lifetime.
^
DSB
Miter
, SquarMrtW
n' AMl> PP Filter, D<tc(tor
LO
RFFront-End Unit
Figure 3.1 -B lock diagram of architecture of HMR
31
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3.2.1 A n te n n a
The radiometer antenna consists of probe-fed stacked patch design [32].
There were two motivations for selecting this design approach. First, probe-feed
mechanism renders the antenna performance less susceptible to variations in
material properties.
This was especially important in this case given that the
dielectric substrate used is FR-4 w ith ^ =3.4±0.2 and 7an£ = 0.23 + 04. Second,
the stacked-patch approach allows for higher achievable antenna gain. A typical
representation of a microstrip patch antenna is shown in figure 3.2. In theory,
design of microstrip patch antennas is governed by the following relations [33]:
120^r
z„=-
eff
,
W
— + 1 .3 9 3 + 0.667 In — + 1.444
h
h
fo r^ > i.
eq (3.1)
h
Where W denotes the width of the microstrip feed-line; and h is the substrate
thickness. The effective dielectric constant is related to the relative permittivity
through the expression:
n -
£R+ \ + £R ~l
^
-
2
~—
2
+0.04 V
v
W/
V
-\—
1/2
1+ 12—
I 1 + 1 2 —
W
—
; forW/h< 1
h.
1/2
; fo r W / h> 1
32
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
eq (3.2)
For a regular microstrip patch antenna, the resonant frequency of the radiating
element can be determined from:
f r = — r = ------------- 7= .
eq (3.3)
l^n ^ E o (L + 2 A L ) ^
with,
(eefr + 0 . 3 ) ( W / h + 0.264)
AL « 0. 412/ z— ^
e
q
(3.4)
(eeff- 0 . 2 5 8 ) ( W / h + 0.8)
Where L corresponds to the size of the patch element and
al
denotes a distance
between radiating patch edges and wall-edges at which the electromagnetic
fields are computed.
Figure 3.2 Representation of a inset-line fed microstrip patch antenna
To improve directivity characteristics of antenna, a metallic horn bucket
was designed to house antenna element [32]. Figures 3.3, 3.4 and 3.5 show the
33
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
layout of the antenna, representation of metallic horn and the cross-section view
of antenna and horn bucket assembly respectively. An interesting feature of the
metallic horn as shown on the figures is the notched edges in the bucket walls.
These notched edges measure a quarter-wavelength in size and help to reduce
antenna back-lobe effects.
The two patches used in the design are essentially air-spaced and are
held apart using 0.25” long nylon standoffs as shown on Fig 3.5. These spacers
in turn, are held in place using nylon nuts and bolts.
In the design, the lower
patch is located 0.25” above ground plane 1 while the upper patch is elevated by
a distance 0.645” above the PCB surface. The upper patch is rectangular shape
and measures 82.1mm x 105mm in size while the lower patch is square shape
and measures 97mm x 97mm. The probe element feeding the antenna is the
aluminum screw used to hold the assembly together. This probe is connected
underneath to a 50Q microstrip line terminating at one input of the dicke switch
located in the receiver RF Front-End.
Simulation of the antenna design and metallic horn assembly was
performed through a contractor at NASA/GSFC. Printed circuit board fabrication
was done using an outside vendor. Upon fabrication of the design, evaluation of
antenna performance was conducted at the Goddard Automated Antenna
Measurement System (GAAMS).
In principle, characterization of antenna
performance is performed by interpretation of E-plane and H-plane field patterns.
Respectively, these two patterns describe the characteristics of electric field
vector and magnetic field vector behavior exhibited by the antenna. Considering
34
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
a spherical coordinate system, a two dimensional pattern in each of the plane
can be obtained by fixing one of the angle {9 o r ^ ) while varying the other.
Keeping <j> constant, and varying 9 (0°< 9 < 180°) gives the elevation pattern.
Similarly, keeping 9 constant, and varying (j> (0°< (/>< 360°) gives the azimuthal
pattern.
Figure 3.6 shows the measured directivity characteristics of HMR
antenna. These measurement patterns were obtained by mounting the antenna
on a positioner at a distance sufficiently exceeding the far-field distance away
from a transmitter. The measurement procedure involved using the positioner to
scan the antenna across the transmitter from -180° to 180° in given step
increments. Simultaneously, the transmitter was rotated for its polarization to be
either aligned or orthogonal to the antenna’s polarization as to result in co- or
cross-polarized measured patterns.
For a given antenna, these patterns are derived from computation of
electric (E) and magnetic (H) fields in far-region of antenna according to the
following expressions [33]:
(K h
sm 0 — a
\
•
eq (3.5)
F(<t>) =
;
(H-plane)
35
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
eq (3.6)
Angles Gand# essentially represent the vertical and horizontal orientation
of the radiation as the antenna moves in the corresponding XY plane of a
spherical coordinates system. Parameter kQ denotes the wave dispersion and W
represents the dimension of the patch element in the radiating direction while h is
substrate height. The measured results of the implemented antenna are found to
be in close agreement with simulation and reveal 3dB beamwidth of 27° for the
structure with sidelobes suppression better than -20dB.
Lower patch
Upper patch
R6.69mm
Clearance
hole
22 m il wide
{istrip
R3.79mm pad
6 x 20mil vias
arranged
3.39mm from
feed point
centre
26.25mm
37mm
12 x 20mil
vias arranged
7.09mm from
feed point
centre
82.1mm
97mm
Figure 3.3 Layout view of stacked patch antenna design and associated dimensions [32]
36
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
159.0
106
Side panels
367.3
B o tto m
318.0
305.2
-o106
Figure 3.4 Representation of antenna horn sidewall design with notched edges. The side panels
are shown with the seams detached and flattened out. All dimensions are in mm [32]
37
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
#6-32 aluminium
M -F 0.25” diam
standoff
#6-32 M -F 0.25
diam. nvlon standoffs
r Datch
lower patch
bucket
clearance
I hole
0.99 i+mm
3M ™ tape
type 941N
FR4
Ground plane 1
305mm
Ground plane 2
microstnp lme
0.559mm wide
#6-32 Nylon bolt
S M A connector
Figure 3.5 Cross-sectional representation showing patch, aperture, bucket and microstrip
assembly [32]
38
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
1 30 -150 -120
-90
0
-60
60
90
120
150
Magnitude [dB]
3dB beamwidth: 19 deg
Return Loss: > 18dB
Directivity:
14.1dBi
Elevation angle (degree)
—1— 1,35GHz (co-pol) —
1,4GHz (co-pol) — - 1,5GHz (co-pol)
Figure 3.6 Measured antenna pattern at varying scan angles.
39
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
100
3 .2 .2
R e c e iv e r R F F r o n t -E n d U n it
The handheld unit receiver follows a superheterodyne architecture as
shown in figure 3.1, consisting of a mixer and local oscillator components for
down-converting the incoming RF signal into lower frequencies components. For
improved sensitivity, the radiometer uses a pre-amplifier stage (NF=3dB) located
between dicke switch and mixer. Being the first gain stage in the cascade chain,
this pre-amplifier allows for significant reduction in overall system noise figure
based on expressions (2.1-2.8) previously discussed in Chapter-2.
The
schematic of the receiver unit is depicted in figure 3.7 below.
During operation, the radiated RF energy captured by antenna enters the
system through the switch when in its open-state and is immediately amplified
through the pre-amplifier prior from reaching the mixer. A voltage controlled local
oscillator operating at a frequency very close to that of the RF signal is used as
second input to the mixer.
The resulting intermediate frequency (IF) signal
output of the mixer is then processed by a low-pass filter and subjected to two
additional stages of amplifications. Next, the signal is filtered again prior of being
sent to a detector.
It is important to note that, throughout this operation, the
dicke switch continuously switches between antenna input and an internal
reference load terminal for purpose of calibration. The particular switch in use in
the handheld radiometer is a single-pole double throw (SPDT) CMOS based
integrated circuit surface mount (SMT) component. This component is actuated
in the design through the use of a transistor-transistor logic (TTL) clock signal
generated from a 555Timer operating at 50% duty cycle.
40
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
VCC = +7
C14
ANT-IN
_ L _ C26
+5V
LAT-5
SWITCH
(PE4246)
C19
C8
U5
C24
TTL=3V
C28
670Hz
MIXER
(RMS-11F)
C15
U6
L11
C13
J - v 'V V '
vco
C2
U3
U4
C4
Hh
FILTER
(PLP-21.4)
R6
C10‘
AAV
R5
C18
Detector
To back-end
Figure 3.7-Block diagram of RF receiver Front-End
41
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
3 .2 .3
B a c k -E n d P r o c e s s in g U n it
The instrument Back-End was implemented following a fully analog
approach for simplicity.
The primary motivation for not opting to use
commercially available digital processing units such as “field programmable gate
arrays” (FPGA) or “digital signal processors” (DSP) was driven by advanced
knowledge needed to use pre-configurable logic. Table 3.1 illustrates a selective
list of commercially available FPGAs along with their respective cost and
operating power requirements from some of industry leading vendors including
Xilink, Altera and Lattice Semiconductors. Prolonging the handheld radiometer
battery operating lifetime requires use of low power consumption electronics.
Thus, the use of an evaluation board was not appropriate. Finally, it was decided
an analog approach was simpler to use in this application.
T a b l e -3 .1 :
L is t o f
some
c o m m e r c ia l l y
a v a il a b l e
FPGA
e v a l u a t io n
boards
along
w it h
ASSOCIATED OPERATING POWER REQUIREMENTS AND COST.
Model Name
Manufacturer
Cyclone IM
Altera
16.0 V
$995.00
LatticeECP2
Lattice
12.0V
$1,295.00
Spartan-3
Xilink
5.0V (0.2A to 1.5A)
$495.00
Virtex II Pro FPGA
Xilink
5.0V (@ 3A)
$595.00
Supply Power
(Vcc)
Unit Cost
(evaluation board)
The implemented Back-End circuitry is one that demonstrates the basic
functionality of a Dicke radiometer. Figure 3.8 shows the implemented back-end
42
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
circuitry for the instrument.
To guarantee synchronous operation between the
Back-End unit and the front-end dicke switch, a transistor-transistor logic clocking
circuit was developed. This logic circuit generates a square wave signal at 50%
duty cycle. The back-end processing unit is implemented using high precision
operational amplifiers and essentially consists of four stages. At the first stage of
this circuit is an adder and a subtracter realized using an inverting and non­
inverting op-amps both with a gain of unity. These two share a common input
originating from the detector. The second stage contains a SPDT analog switch
having for inputs, the adder and subtracter.
Synchronous operation between
dicke switch and analog switch is enforced by the 555-Timer control signal as
shown in figure 3.1. The third stage consists of an integrator sub-circuit. Special
interest was placed in realizing this phase of the design as its performance
presents direct implications on the overall sensitivity of the system. Particularly,
since the integration time constant as previously discussed in Chapter-2 is set
here for the design.
The final stage of the Back-End is composed of an
additional gain stage to increase the level of the processed output signal and a
buffer circuit for further cleaning of the signal.
43
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
J"L
TTL control
1KQ
CFA227 2
GTO /
■ W r
100kfl
RF Front-End
receiver
(Detector OUT)
WV
0FA227 4
0V1V
1kQ
I
1k!)
20kd
AAAr
200kC
15uF
5kQ
Fig 3.8 Schematic of implemented analog Back-End processing unit highlighting the four major
building blocks of the circuitry
44
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
3 .3 P a r t s S e l e c t io n
The handheld microwave radiometer is intended to provide accessibility to
remote sensing instrumentation to higher-education institutions that could not
otherwise afford the typical associated high cost of more sophisticated
instruments. For this purpose, the handheld device ought to be low-cost. This
particular aspect of the design translated into budget constraints during creation
of parts inventory for HMR. As a result, tradeoff between cost and performance
represented the dominant factor in selection of electronic parts for the system.
Among other considerations for the design were size and weight, spatial
resolution, ease of calibration. The design philosophy of the radiometer is to use
a conventional, if not dated, architecture with low-cost parts. A comprehensive
list of major electronic components used in the development of HMR along with
their respective cost is shown in Table 3.2.
HMR system uses an integrated circuit architecture designed on FR-4, a
low-cost printed-circuit board material. In analyzing the IF information resulting
from the down-converting operation, the system relies on the use of cost effective
low-pass IF filters costing only ten dollars rather than using RF band-definition
filters which can cost several hundred dollars.
The amplifiers are plastic-
packaged parts costing only -US$2 each. Several stages of gain are used to
amplify the signal to the desired detector input power level. The receiver along
45
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
with remainder of the system is powered using a single 12-volts rechargeable
battery.
The dicke switch is a single-pole double throw (SPDT) CMOS based
integrated circuit surface mount (SMT) acquired from Peregrine Inc. with
adequate insertion loss of 1dB at frequencies up to 2GHz costing only about
US$3. The second switch is the analog switch located between within the analog
Back-End module.
In the design, all electronics components are carefully
housed inside on-board shielding enclosures. These enclosures provided a cost
effective means for protecting the system against radio-frequency interference
(RFI).
In practice, it is often desired to maintain the electronics at regulated
temperature which is achieved through cooling techniques.
However, these
techniques were not adopted in HMR on ground of expense.
T a b l e -3 .2 :
HMR c ir c u it
COMPONENT
RF dicke
switch
c o m p o n e n t s w it h c o r r e s p o n d in g e l e c t r ic a l p r o p e r t ie s .
PART NUMBER
PE4246
Packaged
amplifiers
ERA-8SM
Filters
PLP-21.4
Mixer
RMS-11F
MAJOR ELECTRICAL
CHARACTERISTICS
COST
(US$)
DISTRIBUTOR
50 Q Non-Reflective
I.L. = 0.8dB
Isolation = 55dB
3.16
Mini-circuits
2.45
Mini-circuits
12.95
Mini-circuits
4.95
Mini-circuits
NF = 3.1dB
Gain (@1GHz) = 25dB
Pout (@1dB)=12.5dBm
IP3=25dBm
I.L. < 1dB
VSWR (passband) =1.7:1
Conversion Loss <7dB
LO-RF Isolation = 36dB
LO-IF Isolation = 29dB
46
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
Voltage
CLV1440E
$6.95
Z-COMM. Inc
DHM020BB
Zero-Bias Schottky
detector
Max. Flatness: 0.3dB
Dynamic range: -50dBm to
+20dBm
125.00
Herotek Inc.
LAT
VSWR=1.22:1
1.95
Mini-circuits
0 .8 1 0 0
Digikey
controlled
oscillator
(VCO)
Diode detector
Attenuators
(SMT)
Voltage
regulators
1)
3 Vdc
2)
5 Vdc
±7V dc
regulators
High precision
operational
amplifiers
resistors,
capacitors and
inductors
(SMT)
resistors,
capacitors and
inductors
(CHIP)
Antenna
standoffs and
screws
Battery
Freq: 1400-1486 MHz
Power output: 0 ± 3dBm
Supply power: 5Vdc
(25mA)
P
m a x
= 0 .5 W
296-2759-1-ND
LM78L05AC-ND
Type: Linear
lu A x = 1 0 0 m A
0 .2 8 8 0
PT78ST107V
PT78NR107V
Type: Switching
Im a x ^ 550mA
Line regulation = ± 0.2%V0
12.975
12.975
Texas
Instrument
1.62
Digikey
(1/4W) 1%; 5% and 5%
tolerance respectively
0.042
Digikey
KXBK-ND
(1/4W)1%; 10% and 5%
tolerance respectively
0.070
Digikey
140406HMN
140406HMN
Nylon and
Aluminum
0.00
(Sample
Above Board
Electronics
(ABE)
OPA277PA-ND
Gain BW=1MHz
Offset <10pV
Low drift: 0.1 pV/°C
)
FB 12-1.4
12 V (1.4Ah) rechargeable
Capacity: 20hrs (70mA to
10.5V)
Weight=1.32 Lbs
15.25
Power Sonic
The printed circuit board material of choice for HMR is FR-4.
This
substrate selection, while primarily driven by the cost requirements of the design,
is also based upon the electrical properties of the material at the operating
47
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
frequency of interest.
Among other aspects to consider in the selection of a
substrate include the dielectric loss tangent,T a n S , which gives an indication of
the fraction of RF power dissipated within the dielectric. The second aspect to
consider is material permittivity, s R , which is a comparison measure of how
similar the selected dielectric is to free space ( s R =1).
For a given microstrip
line, the effective permittivity of the substrate dielectric is related to the phase
velocity of propagating signal according to
vP = - ^ =
eq (3.7)
where, c is speed of light and v^.is phase velocity which, can also be expressed
as follow
vP = X f
eq (3.8)
A = — c=
eq (3.9)
Substituting (3.8) into (3.7) yields
Eq. (3.9) clearly shows the dependence of electrical wavelength upon material
dielectric. This relation in essence represents the fundamental governing relation
in inherent ability to realize miniaturized circuits simply through the choice of
high-permittivity substrates.
Fig 3.9 below compares the signal attenuation in a matched 50-Q
transmission line due to dielectric loss for various materials. Although it can be
48
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
observed that at high microwave frequencies, FR-4 displays the greater loss, it
can be observed that at low frequencies, under 2-GHz, the loss exhibited by all
the materials are approximately equal.
This observation represented a key
attribute for our selection of FR-4 as substrate of choice given that the operating
frequency of HMR design is centered at 1,413GHz.
C h a rt 3: Microstrip Insertion Loss
(0.030" D ielectric Thickness)
0.000
—
m m
i
JL
U
-
1.200
F re q u e n c y . GHz
103003
—
FIFE
Woven Glasi*04003--— *04350
ITG low
Epoxy/fPO
IT/Epoxy
FM
Fig 3.9 Comparison of loss due to dielectric in a microstrip line on different substrate materials
(courtesy o f Rogers Corporation)
The integrated electronics board of HMR is illustrated in figure 3.10. The
top-side of this board contains the circuitry of the system consisting of the frontend receiver, detector and analog back-end processing unit. The patch antenna
aperture is contained on the back-side of the printed circuit board.
illustrates the assembled engineering prototype of the HMR.
49
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
Fig 3.11
Fig.3.10 -Photos of integrated low-cost radiometer printed circuit board showing electronic
compartment on front side of the PCB and stacked patch microstrip antenna on back
side of the board.
50
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
F ig .3 .1 1
operating the instrument
51
Further
r e p r o d u c t io n
of the copyright owner. Further repr
Reproduced with permission
prohibited without permission.
3 .4 V o l t a g e R e g u l a t io n U n it
Regulating the supply power is a critical aspect of all radiometer designs
to ensure constant power level. This practice helps alleviate potential swifts in
biasing conditions of active components within the system.
For portable
microwave sensors such as HMR, the voltage regulation aspect represents an
even more pertinent bottleneck issue due to the very nature of power supplies
used in this case.
Handheld systems are powered using dc batteries which,
unlike ac power generators typically utilized in ground-based instruments
counterparts, have a limited voltage and current handling capability with finite
depletion lifetime. Therefore, the importance of the voltage regulation unit for this
type of instrumentation cannot be overstated.
The HMR is powered using a single 12V rechargeable sealed lead-acid
battery (Power Sonic, part# FB12-1.4).
This battery, in addition to reducing
system maintenance costs by being rechargeable, also offers a high current
handling capability of 1.4-Amp.hr.
For purposes of extended battery lifetime
between recharging cycles, a “Turn ON and OFF” power switch is made
available on the front panel of the instrument.
The voltage regulation unit for
HMR was realized using commercially available high-linearity voltage regulator
parts. In combination to the battery supply, these components provide clean and
stable voltages at various stages of the system.
A list of components used to
realize the voltage regulation circuitry of HMR system is depicted table-3.3. The
unit consists of a variety of switching DC-to-DC converters essential in cleaning
52
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
the DC power by effectively removing any picked up noise from the bus lines.
Outputs of these converters are filtered to remove switching noise. The resulting
voltages therefore, have a ripple less than 10 millivolts (<±0.2%v0). Another
major consideration for choosing switching units is efficiency which, in this case
was determined to be greater than 85 percent. This unit sets and maintains the
voltage across the various amplification stages of the receiver front-end at
constant potential of 7 volts. In the analog back-end, the voltage regulation unit
provides the necessary positive and negative 7V to the dual polarity, highprecision operational amplifiers. Furthermore, this unit functions as power supply
to additional regulators also housed inside the radiometer.
T a b l e -3 .3 : L is t o f
v o l t a g e r e g u l a t io n u n it c o m p o n e n t s .
PARTS
VENDOR
PART NUMBER
UNIT-COST
Digital output display
Lascar
Electronics
EMV-1025S-01
$29
Power switch
RadioShack
N/A
$3
Battery
Power Sonic
PS-1212
$12
Battery holder
Home Depot
N/A
$2
Voltage regulator (Neg.)
Digikey
PT78NR107V
$12.95
Voltage regulator (Pos.)
Digikey
PT78ST107V
$12.95
53
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
An investigation was conducted to study the linearity of the switching DC
to DC converters parts used in realizing the voltage regulation unit notably,
components PT78NR107V and PT78ST107V.
battery was attached to the
V !N
oscilloscope was connected to
V 0
In both cases, HMR 12-volt
terminal of the regulator components and an
ut
lead. The output voltage was recorded over
a continuous period of 1.5 minutes in 5 seconds increment as shown in fig 3.12.
From the information in the figures, it is possible to determine the degree of
flatness of these electronic components.
Here, we define the effective line
regulation of the regulator components as the percentage difference between
their maximum and
minimum
readings.
Hence, the positive regulator,
PT78ST107V, displays a line regulation of 1% of expected output while,
PT78NR107V exhibits a 0.02% of
V
out
line regulation. Together, these values
were found to be in close agreement to manufacturer specifications.
54
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
>
7.08
4)
O)
(0
O
>
3
Q.
3
o
7.06
7.04
7.02
0
10
20
40
30
50
60
70
80
Tim e (Sec)
(a)
_7
^
-------------------------------------------------------------------------------------------------
-7.002
---------------------
>
<u
S’ -7.004
O
>
3 -7.006 -
a
3
o
-7.008 -
-7.01
I "
0
"
I 1"
10
1i
20
t ^
30
1 ‘- i 1 1
40
1 i <■
..■ 1 111 "
50
60
1 1 i 111 "
70
80
Time (Sec)
(b)
Figure 3.12 -O utput responses characteristics of high-linearity, switching ±7V voltage regulators,
(a) Voltage output of PT78ST107V and (b) Output voltage for PT78NR107V
55
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
3 .5 A n t e n n a P a t t e r n C o r r e c t io n A n a l y s is
In radio astronomy studies, the accuracy of the measurements heavily
depends on how well the antenna pattern and all associated uncertainties are
known. While it is desired to realize an antenna exhibiting a narrow beam with
no sidelobes characteristics, in practice however, one must accounts for
contributions of surrounding objects within the line-of-sight or beamwidth of the
antenna.
A pattern correction analysis was conducted on measured HMR
antenna gain pattern in an effort to isolate noise contribution of surrounding
sources from the data. A method centralized around the half-power beamwidth
(HPBW) was adopted for this procedure.
The technique assumes that for a
symmetrically shape antenna pattern in respect of the boresight axis, half or
more of the captured radiated power is contained by the antenna’s main-beam
within the interval
0r , ± 8 \
C e n te r
\m a g ( 8 ce„ ,cr ) - 3 d B
I
J
eq (3.10)
where 0 represents the antenna scan angle.
In general, the antenna radiometric temperature is proportional to the input power
of the receiver. As a result, it may be described as an integration combining the
antenna gain and the brightness temperature summed over all directions [35] as
follow:
56
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
eq (3.11)
Ta = Jr5(Q')G(Q',Q)JQ'
where G(Q!,Q) is the gain of the antenna in the direction Q'. This quantity is
often normalized over the solid angle dQ' such that
eq (3.12)
In practice, G(Q',Q) is readily obtained by conducting testing of the antenna
structure in an anechoic chamber or antenna test range.
For known antenna
gain pattern G , TB values can be obtained through inversion of (3.11).
The antenna correction algorithm we have implemented closely follows
that of [35] but incorporates two variations. First, it is assumed that most of the
power captured by the antenna is concentrated within the ± HPBW region
according to (3.10) and that, the remaining power is uniformly distributed across
the Earth and cold sky.
Second, the average brightness temperature
contributions outside the main-lobe but within Earth’s limb are equated to
correspond to ambient temperature fixed at 290°K. The following approximation
is thus applicable
TA * t1- (p + <l)\TB + PTamb + Acosmic
cosmic
eq (3.1 3)
where Tamb =290°K and Tcosmic =2.7°K ; ^.corresponds to the mean brightness
temperature concentrated across the HPBW region,
p and q are coefficients
readily obtained from analysis of gain pattern measurements as follow
eq (3.14a)
57
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
n
q = 'ln: Jg(0, </>= 0)sin(9)d9
eq (3.14b)
where 9X and 62.are delimiting scan angles giving boundaries of the fraction of
p and q relevant for these computations. Choice for setting </>= 0 is determined
by the single-polarization characteristic of the antenna.
Precisely, only the co­
polarization gain patterns are utilized in the algorithm.
The mean brightness
temperature can therefore be deduced as
3 .6 C a l ib r a t io n
o f In s tr u m e n t
While a 2-point calibration yields many useful characteristics of a
microwave radiometer (i.e. Gain, offset, Y-factor, etc.), it often does not account
for losses due to impedance mismatch. It is common practice to compensate for
gain fluctuation within the receiver system by switching back and forth between
two (or more) temperature-controlled reference terminals as shown in figure 3.13.
Although useful in compensating for the fluctuation, these internal references do
not offer insight into antenna losses. As a result, understanding the mismatch
effects to the system is necessary for complete characterization of the microwave
sensor noise temperature. Neglecting mismatch effects, principally at antenna-
58
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receiver interface, whenever estimating the effective input noise temperature of a
system results in errors [16, 35].
Thot
rhot
TAnt
LNA
rcold
Tcold
Figure 3.13- Block diagram of typical radiometer receiver calibration set up.
As reported in [34], the calibration of the HMR was tested under two
considerations: 1) Assuming perfectly matched conditions, and 2) Taking into
accounts noise contribution due to impedance mismatch. This experiment was
carried out in a laboratory and consisted of a two-point calibration procedure for
total-power radiometers similar to that described in [16]. During the procedure,
the antenna was replaced with a matched load (WEINSCHEL #1419). The “hot”
point was taken to be 295K. The load was then dipped into liquid nitrogen (78K)
for the ‘cold’ calibration point. In the two cases, the load was connected to the
system front-end using a low-loss, 6-feet long flexible cable with a loss factor of
0.35dB/ft. The corresponding output noise temperature is estimated to be 178K.
For single port microwave systems, under matched conditions (i.e.
impedances at receiver input and ambient-load termination are conjugate of each
59
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other), assuming there are no sources of mismatch present within the system,
the delivered input noise temperature to the receiver can be found to be [16]:
eq (3.10)
where
L:
represents the loss factor under matched conditions
To: room temperature (295 K)
Tioad'■effective antenna noise temperature
When taking into account effects due to impedance mismatch at the loadreceiver interface, (3.10) transforms into [36]:
Tm = [ M ] [ [ y ] T Phy + (1 - y)T0 + (1 - M ) T Kec,
where
eq (3.11)
eq (3.12)
In the above expression, TLoad and r Recare the complex reflection coefficients at
the ambient-load and the receiver input terminals respectively. These frequency
dependent quantities were measured using a HP8510C vector network analyzer.
The term Ls is the loss parameter of the network analyzer. The mismatch factor
(M) of the system, appearing in (3.11) is explicitly expressed as:
60
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This mismatch parameter can be defined as the proportion of the available power
from the noise generator (i.e. termination-load) delivered to the receiver and
varies between values of 0 -to -1 for a passive system; with “ 1 ” representing the
perfectly matched case (i.e. \YLoad \ = |rRJ = 0).
Plots of
YLoad and r Rec are
shown in figure 3.14. As it can be observed, not only are the real components of
these reflection coefficients varying as function of frequency, but also does their
phase thus, further justifying the need to treat these parameters as complex
quantities. Through expansion, the denominator of (3.13) can be shown to be
[36]:
| i - r R„ r £Mj f = i - 2 | r R„ r tMjc o s « + | r R„ r w f
eq (3.14)
with,
|r
L°ad
| SWRLoad- 1
SWRLoad +1,
eq (3.15)
where S W R denotes measured values of voltage standing wave ratio done using
a network analyzer.
I r fev |= SWRrcv 1 ,
eq (3.16)
*cv sff7?Rcv+ r
r » =Sn +
,
*
22
Rev
61
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eq (3.17)
For simplicity purposes, our analysis will assume the HMR to be a perfectly
unilateral system; in which case, there is no reverse transmission scattering
parameter (i.e. S 12 = 0). Expressions (3.17) and (3.18) above then reduce to:
eq (3.19)
eq (3.20)
In (3.14), 0 refers to the combined phases of the reflection coefficients.
eq (3.21)
For a given receiver system, the LNA is predominantly responsible for
mismatch due its close proximity to the antenna.
In our design, a monolithic
packaged amplifier component (ERA-8 SM) acquired from Mini-Circuits Inc. was
used at the pre-amplification stage. This component features a VSWR of 1.7 at
1.413GHz corresponding to a reflection coefficient of 0.26. This component in
addition to the interconnecting transmission line is attributed to the measured
|r Rec| = 0.33 that can be observed in figure 3.6.2(b).
The resulting mismatch
factor is about 0.86 at our frequency of interest.
62
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0.4
0 .4
Rec
0.3
0 .3
0.2
0.2
j*********** **
<D
a:
0
*
- 0.1
“ *“ *
—
- 0.2 -;
Load
-0.3 -
0.1
-
0.2
V
-0.4
1
-
1.1
1.2
frequency (GHz)
0.5
1.413 GHz
Rec
0.4
0.3
0.2
Load
1
1.1
1.2
1.3
1.4
1.5
1.6
frequency (GHz)
(b)
Figure 3.14- ‘load’ and ‘receiver’ reflection parameters: (a) Real and Imaginary parts of the
reflection coefficients, (b) Magnitude of the reflection coefficients
63
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Using these VNA measurements along with expressions (3.10-3.21), it is
possible to analytically determine the effective receiver noise temperature for the
various calibration reference targets while accounting for mismatch effects.
Table-3.4 lists the results of this investigation for HMR.
T a b l e -3 .4 : E x t r a c t e d
n o is e t e m p e r a t u r e o f c a l ib r a t io n r e f e r e n c e l o a d s
Termination-Load TPhy
TtN assuming matched
conditions
T/w considering effects
due to mismatches
295 K
295 K
304.15 K
78 K
178.46 K
223.72 K
1
c\i
II
-j
£
Having analytically determined the receiver noise temperature to be TRec =
460K, and taking To = 295K, the HMR calibration analysis was performed for the
two cases described above.
The resulting curves from that investigation are
shown in figure 3.15.
To relate this mismatch to actual radiometer soil scenes’ readings, the
antenna along with horn assembly reflection coefficient was measured using the
network analyzer and the corresponding M was determined by substituting YAnt
for Tioarf into (3.13). The input noise temperature for the given “cold” and “hot”
loads described above was found from (3.11).
The derived information was
useful in determining the system response parameters (i.e. gain and offset).
Taking the system output voltage to be linearly proportional to the microwave
64
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radiation, these parameters subsequently were used to estimate the brightness T
of any other natural scene under observation by the radiometer.
350
Teal [K]
300
250
200
150
200
250
300
350
400
450
Vout [mV]
— ♦— Curve (Considering Mismatch)
Curve (Assuming Match)
Figure 3.15- HMR calibration curves; dashed line represents the case assuming matched
conditions, solid line denotes instrument calibration response accounting for
additional noise temperature contribution due to impedance mismatch.
3 .7 S u m m a r y
In this chapter, an elaborative description of the low-cost handheld
microwave radiometer developed within the framework of this dissertation was
65
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presented.
While the instrument configuration follows that of a basic dicke
radiometer, special attention was given in the various electronic parts selections
for purpose of minimizing development cost.
To that very end, cost efficient
approaches were investigated for RFI mitigation and temperature regulating units
were omitted in this instrument.
Together these aspects, while representing
potential limitations to the proper operation of the device, were accounted for
through a detailed calibration procedure also presented in this chapter.
For
added robustness to accuracy of the calibration, effects due to impedance
mismatch were taken into consideration during the procedure.
Moreover, a
characterization of the instrument was further investigated by performing an
antenna pattern correction analysis to effectively eliminate the contributions due
to background sources and sidelobes in the antenna field of view.
While
Chapter-5 of this document investigates the field experiments conducted using
the handheld radiometer, the upcoming chapter will first present the design and
characterization of a radiometer microwave integrated circuit receiver RF frontend module also developed within the framework of this dissertation research.
66
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C hapter 4
I n t e g r a t e d R F F r o n t -E n d D e v e l o p m e n t F o r
a D ir e c t S a m p l in g D ig it a l R a d io m e t e r
4.1 I n t r o d u c t io n
This chapter outlines the development of an integrated receiver front-end
designed for use in a direct sampling digital radiometer.
A discussion of all
individual building blocks of the receiver including their design methodology is
provided.
microwave
During the discussion, quantitative figure of merits, pertinent to
radiometers,
for
each
of
the
components
are
examined.
Measurements results for each component design are presented and discussed.
Finally, a thorough evaluation of the integrated receiver is also performed.
67
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4.2
D e s c rip tio n o f
DSDR System
The Center of Advanced Microwave Research and Applications (CAMRA)
at Morgan state University is developing a Direct Sampling Digital Radiometer
(DSDR) for purpose of soil moisture studies. The project is aimed at reinforcing
CAMRA capabilities in developing scientific instrumentations to support NASA’s
mission. This DSDR will be in many ways similar to that proposed by Fischman
[37] in 1998 with major addition of a high-speed Analog to Digital Converter
(ADC) containing at its core, a track-and-hold (T/H) circuit and comparator as
investigated by Thompson in his thesis dating 2003 [38]. Fig. 4.1 illustrates the
architecture of the DSDR system as proposed by Thompson. In the architecture,
the T/H circuit of the ADC serves the purpose of undersampling the RF noise
signal sensed by the antenna.
This undersampling operation results in an IF
alias that is directly converted into digital form thereby, eliminating the need of
down-converting the RF signal.
SWITCH
LNA
BPF
RF AMP I
t /h
circuit
Comparator
Calibrate
101101
BW
Measure
ANT
RF Analog
/ \
Digital
Fig 4.1 Block diagram of direct-sampling digital radiometer system [38]
68
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DSP
While DSDR is a large project effort involving several research centers within
CAMRA, the work reported in this dissertation focuses on the development and
procurement of the RF Front-End.
4 .3 W o r k O b je c t iv e s
The main objective of this portion of the research is to provide support to
DSDR by developing and providing the Receiver RF Front-End for the
instrument.
As seen in figure 4.1, this receiver comprises of a Low-Noise
Amplifier (LNA), Band-Pass Filter (BPF) and a RF gain amplifier. It is desired for
these components to be integrated unto a single microwave integrated circuit
(MIC) printed circuit board with two interface connectors access (i.e. SMA) for
ease of future integration with both, an antenna and a digital Back-End.
The specifications for the respective building blocks are driven by science
requirements for soil moisture studies.
An important radiometer performance
factor for soil moisture applications relates to system sensitivity, AT, as
previously discussed in chapter-2. This parameter denotes the minimum emitted
thermal energy from the soil scene that can be disseminated by the radiometer.
This energy is directly proportional to the amount of water content of the scene.
In designing radiometer systems, it is desired for AT to have a very low Kelvin
value which corresponds to superior system sensitivity.
between AT and overall receiver noise temperature
The relationship
Tsys was previously
69
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highlighted through expression (2.6) for an ideal total power radiometer. Tsys, on
the other hand, is directly related to system overall noise figure (NF) performance
according to,
T sys =
To
( N F -1)
eq(4.1)
where T0 = 290° K is ambient physical temperature.
In a cascade receiver such as DSDR, the overall system NF is primarily
governed by the first active stage or more precisely, the LNA. As a result, careful
considerations ought to be placed in designing this component.
4 .4 B a n d -P a s s F il t e r D esig n
In microwave radiometers as in other communication systems, the filter
plays the essential role of band selection while simultaneously preventing out-ofband interferences from reaching the sensitive receiver.
The proliferation of
modern mobile communication units and military radar systems has now made Lband radiometer systems particularly prone to radio-frequency interference (RFI).
Although the frequency allocation from 1400 MFIz to 1427 MFIz has been
designated by the Federal Communications Commission (FCC) as a protected
band for radio-astronomy observations, radiometers operating within the band
remain exposed to contamination from other man-made signal sources operating
near the protected-band edges. It appears therefore that in designing a filter for
70
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use in a radiometer system, the three most important criteria one ought to satisfy
include: 1 ) maximally flat insertion loss across the passband; 2 ) high out-of-band
signal rejection, and 3) sharp roll off at band edges.
4 .4 .1
F il t e r D e s ig n C o n s id e r a t io n s
Any given design of microwave filters begins with gathering of design
requirements or considerations including center frequency band and bandwidth
of operation. For bandpass filters, the band center can be determined by
/o = VyTTT
where,
/j
eq (4-2)
and f 2represent the lower and upper edges of the passband
respectively. The fractional bandwidth is given as follow
BW = Jf i2 - fJii
w = -----/»
4 /J i
eq tA
(4 .3O\)
While it is possible to realize filters using strictly lumped elements, at microwave
frequencies this practice presents some drawbacks. Foremost, lumped elements
are generally only available for limited range of values thus, making it difficult to
achieve high design accuracy. Also, unlike their discrete counterparts, lumped
components are prone to temperature drift adversely affecting their tolerance
factor.
Such drift often result in disparity of actual component values from
specified units. A well-known method commonly used in translating a lumped
71
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design into transmission line sections is the Richard’s. This method consists of
substituting open- and short-circuited transmission lines for the lumped elements
present in the circuits based on the following relations
Q = tan(/7 /) = tan 6
eq (4.4)
jX L = jQ L = jL tan 6
eq (4.5)
jB c - jC lC = jC tan 9
eq (4.6)
where / denotes the transmission line physical length;
P = 2k !X
is the
propagation constant; Xi. and Be represent an inductor reactance and a capacitor
susceptance respectively. It thus appears that a short circuit can be substituted
for an inductor while an open circuit for a capacitor.
In brief, this technique
consists of assembling shorted stubs, open stubs and cascaded transmission
lines all with similar phase length.
While the method discussed above is practical for common filters, it
suffers from adaptability to more robust designs especially those with small size
and narrow fractional bandwidth requirements.
Two alternative approaches
better suited for such condition include the coupled lines method and steppedimpedance resonators (SIR) technique.
Figure 4.2(a) shows a typical
configuration of microstrip coupled-lines. In this formation, the input impedance
of the structure is given as
Z m
= ~ J y l Z o e ’ Z oo c o t 0
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eq (4.7)
where
Z0e:
even modes impedance
Zoo'-
odd modes impedance
One interesting observation to mention is, for# = 90°, expression (4.7) reduces to
Zin = 0 ; which implies that there will be a short circuit at the input of the structure
thus, resulting in a transmission zero. Owing to this characteristic, coupled-lines
structures are often utilized to extend the stopband frequency of filter designs by
suppressing spurious modes [45].
(a)
(b)
Fig 4.2 -Design topologies for filters; (a) Microstrip Coupled-lines structure, (b) Grounded
Microstrip SIR structure
XIA
The microstrip line SIR structure, shown in fig 4.2 (b), relies on the
capacitive and inductive coupling at resonator ends to achieve filtering.
For a
grounded XIA SIR as shown of the figure, the input admittance at the open end
can be expressed as:
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tan(<9, )tan(<92) - i ?
,N
eq (4.8)
2 ta n (0 ,)+ i? ta n (0 2)
where:
0,:
Electrical length of transmission line Z,
B2:
Electrical length of transmission line Z 2
R:
Stepped impedance ratio Z 2/ Z x
Spurious modes are minimized at resonance; for Ym = 0. This condition
occurs only when
r
= Y tan(0,)tan(02)
eq (4.9)
Recently work on filter design exploiting characteristics of SIR structures have
been reported [46]
4.4.2
D
e s ig n
M
eth o d o lo g y
The specifications for the filter developed under this effort are listed in
table 4.1.
Fig. 4.3 and Fig. 4.4 show the layout representation and equivalent
circuit of the microstrip filter respectively. The design, consisting of two coupled
resonators, uses a combination of meander lines of varying widths and the
coupling mechanism of the two resonators to achieve the bandpass response.
The first resonator is based on the meander loop concept described in [39] for
purpose of miniaturization. This segment is coupled on the left-hand side to a
quarter-wavelength stub.
74
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T a b l e -4 .1 :
BPF
S p e c if ic a t io n s
Parameters
Targeted values
Pass band
1 .4 -1 .4 2 7 GHz
Insertion Loss (In-band)
> -3dB
Return Loss (In-band)
< -15 dB
Out-of-Band rejection
< -45 dB
Figure 4.3 Layout of filter in microstrip form
75
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Cl
C lr >Lg
Figure 4.4 Equivalent circuit schematic
Parameter g,sets the first transmission zero of the filter with segment
/, = 0.08^o( kg0 denotes the guided wavelength at operating frequency).
The
second resonator is similar to that of slow-wave resonator from [40]. The edges
of the resonator are modified to accommodate the inset section extending from
the first resonator thereby providing maximum coupling between the two
elements.
In the design, we set l 2 =0A52AgO, a n d g 2 = 0. 3m m .
The input and
output feeding lines are weakly coupled to the structure by placing them at a
wide gap distance away so as not to disturb the propagating mode of the filter.
The resonators are coupled through elementCgap shown in fig.4.4. By adjusting
the values of the various circuit elements, a bandpass response can be
achieved. For our operating frequency of interest centered at 1.41-GHz with BW
=
27MHz,
corresponding
values
76
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are:
c,
= \O pF ,c2 = 12PF,
c3 =4.8pF, c4 =1.2p F , L p = 9nH , Lg =2nH,
and
Cgap = 9.35pF-
This
basic circuit is useful in gaining insight about the scheme of operation of the filter.
For instance, the coupling effect between the two resonators can be investigated
here by simply varying c
. It can be observed in fig.4.5 that as this capacitance
value is increased, corresponding to a smaller gap distance between the
resonators, the filter resonance swifts to lower frequencies regions
77
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CO
"O
-— Cgap=9 pF
CO
- - Cgap=9.35 pF
-*-C g a p -9 .7 5 pF
Cgap=10.5 pF
1.3
1.35
1.4
1.45
1.5
1.55
Frequency [GHz]
Figure 4.5 Effects of Cgap on filter resonant frequency
The commercially available Method-of-Moments (MOM) based planar EM
simulator Ansoft Designer v3 was extensively used in simulating the filter. The
substrate material selected
s R = 10.2 and
in our investigation is RT/Duroid 6010 with
h = 1.27mm. For good impedance match within passband, width of
ports is set to w, =1.19mm .
78
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4 .4 .3
D is c u s s io n
of
M easured R esults
The fabricated filter structure is represented on fig.4.6.
The filter
measures 17x21 mm in size. Measurements testing of the structure were carried
out using a HP8510C vector network analyzer (VNA). Figures.4.7 and 4.8 below
show measured performance data collected on the fabricated filter.
Good
agreement between simulation and measurements can be observed. The filter
structure displays a better than -35 dB of stop-band at frequencies up to 3GHz
(< 2/„). Effects of harmonics become visible at 3.2-GHz with higher order modes
being excited in the structure.
Fig.4.6 Photo of fabricated filter
79
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Magnitude (dB)
-10
-15
-20
-25
-30
-35
-40
Simulated
— - Measured
Simulated
— Measured
-45
-50
(S11)
(S11)
(S21)
(S21)
-55
1
1.5
2
2.5
3
Frequency (GHz)
Figure.4 7 - Wideband filter performance results
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3.5
Magnitude (dB)
-10
-15
-20
Simulated (S22)
-25
— - Measured (S22)
-30
1.2
1.25
1.3
1.35
1.4
1.45
1.5
1.55
1 .6
Frequency (GHz)
Fig.4.8 - Narrowband output match results
81
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1.65
4 .5 L o w -N o is e A m p l if ie r D e sig n
Low
noise
amplifier
(LNA)
represents
a
critical
building
block
in
communication systems. It is an electronic circuit within the receiver that serves
the purpose of amplifying the low-voltage signal incoming from the antenna while
minimizing the amount of electronic noise added to the low-voltage signal. For
highly sensitive receivers such as microwave radiometers therefore, LNA
governs the overall system sensitivity NEAT (Noise Equivalent Temperature
Difference) by setting the noise figure (NF).
To satisfy stringent sensitivity
requirements for radiometer systems, it is desired for the LNA to have high gain
with maximally flat response across pass-band, good return loss corresponding
to the matching condition at input terminal of the device and most importantly,
feature a low NF.
To that end, GaAs Metal Semiconductor Field Effect
(MESFETs) transistors and other lll-V compound semiconductors are often
consider ideal candidates for LNAs due to their ability to yield high gain with
minimal NF [41].
With recent advancements of the semiconductor industry, low-cost superiorelectrical-performance packaged devices have now become commercially
available. Few such devices include the ATF low-noise series pHEMT devices
by Agilent Technologies [42], The superior ultra-low noise performance of these
devices in addition to their high gain and high linearity make them particularly
attractive for radiometers LNA designs.
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4.5.1
A
m p l if ie r
D
e s ig n
C
o n s id e r a t io n s
In designing a LNA, the most important criteria include stability, gain, dc
power requirements and most of all, noise. Considerations for such design have
been well reported in literature. Theoretically, synthesis of LNA design requires
understanding of its corresponding two-port network. Fig. 4.9 above illustrates
the typical block diagram of a microwave amplifier and consists of the active
device. It usually consists of a FET or BJT surrounded by an input and output
matching networks.
These matching networks are used to achieve maximum
power transfer between input and output ports of the device. In the figure, Es and
Zs denote input source generator and its impedance. The transistor is terminated
at the output by a load with an impedance Zi_. For this network, it is possible to
derive fundamental equations necessary for the LNA design [43].
Input
Matching
Network
(IMN)
Active Core
(FET or BJT)
n,in
[S] =
Output
Matching
Network
(OMN)
out
S 11
& I2
S 21
S 22
Figure 4.9 -Typical block diagram of a microwave transistor amplifier.
83
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In order to achieve maximum power transfer, the following expressions must hold
true:
r IN= r ;
eq(4.11a)
rOOT=r;
eq (4.11b)
and,
Since the stability of a LNA describes its ability to resist oscillation, it is therefore
desired to an unconditionally stable design.
For a LNA design to qualify as
unconditionally stable, the following relations must apply:
|rw|<l
eq (4.12a)
|ro[/r|<l
eq (4.12b)
A potentially unstable network can be made stable by resistive loading,
introducing negative feedback, or implementing series feedback. Figure 4.10
depicts circuit diagrams of typical stabilization methods.
An index commonly
used to measure the stability of amplifier circuits is the K factor expressed as:
84
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K
1-|5u |2|522|2+1A|2
21Sl2S2l |
eq (4.13)
with,
A = SuS22 —S\2$2\
eq (4.14)
Figure 4.10 -Typical stabilization methods
The power gain criteria of the LNA design represents the ratio of increase
of the low-level signal level during its transfer from input to output of the
electronic circuit.
In the typical LNA design, there are three types of gains
including the transducer power gain (Gj), the operating power gain (Gp) and the
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available gain
( G a ).
While the latter represents the ratio of available powers at
both the network and the source, Gp on the other hand represents the ratio of the
actual power delivered to the load and that input to the network.
Gj
characterizes the ratio of the actual power delivered to the load and that available
at the source. It is the most common of the three types of gain and is useful in
the design of amplifier circuits.
Several literature work including [43] have
focused on transducer power gain in synthesizing microwave amplifiers.
The
quantity Gy is governed by:
It is possible to decompose (4.15) into three components representing the gain
contributions from the input network, the output network, and the active core.
G t = Gs + G0 + Gl \dB ]
eq (4.16)
where
eq (4.17a)
eq (4.17b)
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eq (4.17c)
Other important considerations in the design of amplifier circuits include the
unilateral and bilateral conditions. These two cases relate the effects of the FET
or BJT device on the matching networks. In the bilateral case, the effects of the
active device are of significant value that it cannot be neglected. In the unilateral
case on the other hand, the active core has negligible effects.
Under such
condition, it is possible to make the following assumptions; |S-i2|=0 and r m « Sn.
In addressing the low-noise criteria for LNAs, it is common practice to use
FET devices over their bipolar counterparts.
There are several underlying
semiconductor physics considerations for this selection but the two most
pertinent to the microwave radiometer designer include:
1) FETs exhibit low
source resistance at low microwave frequencies. 2) FETs have a small gate-tochannel separation which leads to high transconductance ( g m).
A GaAs
MESFETs with gate lengths as small as 0.1-jt/m was reported in [44].
Large
values of g m result in minimum noise figure for the transistor according to the
relation:
eq (4.18)
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where a denotes operating frequency and
Cgs represents the gate-source
junction capacitance.
4.5.2 C ir c u it D e s c r ip tio n
Specifications for the LNA design are listed in Table 4.2. The design is
based on an ATF-34143 pHEMT packaged device by Agilent Technologies. The
design is implemented in a common source configuration for high output gain.
Table-4.2: LNA S p e c i f i c a t i o n s
Parameters
Targeted values
Frequency range
1.4 - 1.427 GHz
Gain
> 15 dB
A Gain (gain flatness across band)
< 0.2 dB
VSWR in
<1.5:1
VSWR out
< 1.3:1
NF
< 0.5 dB
Agilent Technologies’ Advanced
Design
System
(ADS)
simulation
environment was extensively used to obtain graphical solutions including stability
circles, small signal performance characteristics for the LNA.
Both input and
output matching networks are realized using microstrip lines for ease of
88
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fabrication and MIC integration of LNA block with remaining receiver components
on a single substrate.
The IMN and OMN are designed using a single-stub
matching technique approach. The objective in this technique is to determine the
proper transmission line length and type that can emulate the appropriate
reactance and susceptance values for the design. A pictorial representation of a
typical single-stub matching network is shown on figure 4.11. Using transmission
line theory, it is possible to derive fundamental expressions to fully characterize
the diagram shown below [43].
Figure 4.11 Single-stub matching of two arbitrary impedances.
In practice, the development of IMN and OMN starts with the selection of a
unique bias point satisfying the power requirements for the design.
Such
condition can be easily determined using CAD tools by performing DC analysis of
current-voltage (IV) transfer curves of the transistor device.
In our design, the
selected bias condition is: VDs = 3V and VGs = -0.4V; corresponding to lDs=63mA.
The resulting DC power consumption of the device is determined to be ~180mW.
89
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
The reflection coefficient values required for matching networks are obtained
graphically using ADS circuit simulator. The simulation setup for this procedure
includes ideal RF-choking inductor and DC-blocking capacitors around the
transistor device with input of output ports connected to 50Q terminations.
Figure 4.12 shows resulting reflection coefficients from this analysis graphically
determined through generation of Gain and noise circles at center frequency of
operation.
U
CD
O
CJ
• pH
- pH
UO
m cd
2 O
m1
i nd e p ( m l ) = 5 1
m2
38
c i r _ p t s
( 0 . 0 0 0
to
2 0 1 . 0 0 0 )
Figure 4.12 - Gain and Noise circles of ATF-34143 at 1.413-GHz.
90
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The desired input reflection coefficient for the matching network can be
graphically determined as that associated with minimum achievable NF at design
center frequency.
This complex quantity is denoted on the graph as the
interception point of the specified noise circle with highest corresponding gain
circle. As thus,
and,
r f f F = r Ms>
eq (4.19)
r M = Z u s ~ Z°
eq (4.20)
JMS
“r1 Zm0a
Substituting (4.19) into (4.20) gives:
7
_
MS ~
i +' *r N F
*
.
p
* — t NF
_ 1 + (0.431 + y 0.402)
~ 1 -(0 .4 3 1 + y'0.402)
eq (4.21)
= 1.34462 + y l .6 5 6 4 8 0 ,
Accordingly, a single-stub input matching network consisting of a series line and
shunt shorted stub can be realized using the following relations:
eq (4.22)
■*IN
YIN= Y '+ Y stub
eq (4.23)
Y ' = ~ = G + jB
eq (4.24)
Z ' = Z 0^ +jZot
^0 j Z i t
eq (4.25)
Ystub = -JB
eq (4.26)
91
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In (4.25), it is assumed that the series line has a characteristic impedance of Z q.
It can be also shown that
eq (4.28)
The length, I,
of the shunt line or stub can be derived using the following
relation:
eq (4.29)
Similarly, output impedance values necessary to design the optimal output
network can also be determined. Figure 4.13 shows the final schematic of the
implemented LNA in microstrip form. The circuit is implemented on RT/Duroid
6010 with sR =10.2 and substrate thickness of 1.27mm. All through-hole vias in
the designs including shorted-stubs vias for IMN, OMN and transistor’s source
terminal vias measure between 0.4mm and 0.6mm in diameter. For purpose of
stability, a 24.9 Q has been added in series with the drain terminal of the device.
The input and output matching networks have been bent in “hook shapes” for
92
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
efficient real estate use of PCB material. All RF components within the design
notably, DC blocking capacitors are low equivalent series resistance (ESR) parts
with adequate self-resonance characteristics; carefully selected as not to have
adverse effects on circuit performance.
Bypass capacitors of value 33nF are
placed at close proximity of power lines to warranty clean DC signal. Table 4.3
lists all component parts used in the design.
93
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
ro
CM
o
cn
o
o
o
LU
alit =
oo„
Figure 4.13 - Schematic of amplifier circuit in microstrip form
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
T able -4.3: LNA Parts L ist
V alue
Unit
D istributor
Part Num ber
24.9
Q
Res, 1/10W, 1%, 0603
1
Digikey
RT0603FRE0724R9L-ND
100
nH
lnd.1008, 10%
2
Digikey
M1188CT-ND
12
PF
Cap,0603, 5%
2
Digikey
06035J120JAWTR-ND
33000
PF
Cap, X5R, 0603, 10%
2
Digikey
GMK107BJ333KA-T-ND
-
-
1
Digikey
516-1506-1-ND
-
-
Packaged device,
ATF-34143
Connectors, SMA (F)
2
Digikey
J502-ND
4 .5 .3
Description
Q ty
L N A S im u l a t io n R e s u l t s
Simulation data on the LNA as predicted by ADS is depicted on figures
4.14, 4.15 and 4.16.
It clearly shows a simulated gain of 15.2dB at center
operating frequency. The LNA displays an input return loss of approximately 13.6dB with corresponding noise figure of about 0.53dB; which is close to
desired value.
The actual NF from measurements is expected to be slightly
higher due mainly due to noise contribution of soldered SMAs at device ports.
The output return loss is predicted to be -12.4dB which is reasonable for
adequate output power transfer from the LNA.
95
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dB(S(2,2))
dB(S(1,1))
dB(S(2,1))
freq=41.413GHz
dB(S(2,1))=15.224
1.413GHz
2,2))=-12.
=1.413GHz
freq, GHz
Figure 4.14 - Wideband simulation of LNA S-parameters performance. Device exhibits large
gain along with adequate input at output return loss performance across bandwidth
of design from 1400MHz to 1427MHz.
A G across design bandwidth is
determined to be less than 0.2dB.
96
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2.0
1. 8—
1.6—
1.4—
c
0 . 8—
m5
0 .6—
freq=1.413GHz
nf(2)=0.527___
0 .4 -
02
.
—
—
1.0
0.5
2.0
1.5
2.5
3.0
freq, GHz
Figure 4.15 -Simplified schematic of LNA noise figure performance. Simulated noise figure
performance across LNA bandwidth of interest is close to desired value of 0.5dB.
2.2
2.0------1. 8—
m4
m4
freq= 1.413GHz
StabFact1=1.436
0.8
0.8
1.0
1.2
1.4
1.6
1.8
2.0
2.2
2.4
2.6
2.8
3.0
freq, GHz
Figure 4.16 - Simulated LNA stability factor performance. K-factor is greater than unity across
wideband simulation frequency.
97
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4.5.4
D
is c u s s io n o f
M
easured
R
esults
A photo of manufactured LNA is shown in figure 4.17. Due to HP8510C
vector network analyzer (VNA) equipment malfunction, small-signal parameter
measurements could not be conducted on the manufactured circuit. Instead, a
non-conventional testing procedure was developed to access the performance of
LNA. The experiment utilized a function generator (Agilent E4433B), spectrum
analyzer (Agilent E4403B), DC power supply. During the procedure, the function
generator was connected to the input of LNA with output attached to spectrum
analyzer. The function generator was used to provide an input signal at a fixed
power level but varying frequencies. The center frequency of the design (1.413
GHz) was chosen as test frequency.
Then, the frequency was carefully
incremented and decremented in equal steps as to yield a full sweep across
bandwidth of design. At each frequency step, the spectrum analyzer was used to
measure LNA output.
The effective LNA gain, readily determined from the
difference between input power and measured output power, is shown in fig 4.18.
It should be noted that this procedure only permits evaluation of LNA linear gain
power performance. Measurements revealed that LNA exhibits a linear power
gain higher than 13.8dB across design bandwidth, which is found to be in close
agreement to simulated performance.
98
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Figure 4.17 - Photo of fabricated Low-noise amplifier.
20
18
Measured Gain [dB]
16
14
12
10
8
6
— At Pin=-50dBm
4
At Pin=-25dBm
2
0
1.36
1.38
1.4
1.42
1.44
1.46
1.48
Freq [GHz]
Figure 4.18 - Measured effective Gain of fabricated Low-noise amplifier. LNA displays similar
characteristics at both input power with a measured gain value of around 13.8dB
at center design frequency of 1,413GHz
Another
notable
observation
indicating
proper
operation
of
the
manufactured circuit involved the observed DC power status of circuit during the
99
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experiment.
Indeed, DC power performance check revealed that at operating
bias condition: V Ds = 3.01V and V Gs = -0.42V, LNA draws a desirable Ids = 67mA
as displayed on the power supply which is in close agreement to simulation value
of 63mA.
Using a similar experimental setup, it was possible to conduct an
uncalibrated measurement of the LNA noise figure.
In the procedure, a 50Q
termination load was attached to LNA input port with output connected to the
spectrum analyzer.
The DC power supply was used to provide appropriate
biasing conditions for proper operation of the LNA. The noise figure was readily
computed from the measured output power displayed on the spectrum analyzer
with the input of the LNA terminated.
eq (4.30)
meas
Where,
k
represents Bolzmann constant; B
is the measurements
bandwidth as displayed on the spectrum analyzer screen; GLNA is LNA effective
gain. Under ideal conditions, in absence of mismatches, a 50Q termination load
effectively acts as a perfect blackbody with emissivity of unity as follow:
eq (4.31)
phy
TLf oad
nnrl - T pi h y
100
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eq (4.32)
Taking the physical load temperature to be Tphy= 2 9 5 K , the noise
temperature contributed by the LNA can be determined from (4.30) as follow:
T
1 LNA -
eq (4.33)
295
\ kBGLNA
From (4.33) above, LNA noise figure is obtained by using the relation
T
l NA
~
T 0 {T L N A
0
~
-295
eq (4.34)
\kB G LNA
Taking T0 = 295K , LNA noise figure expressed in [dB] is given as follow:
P meas
'
NFlna =1 0 * Log f
\ T»kBGLNAj
eq (4.35)
In practice however, the emissivity of a matched load varies within the
following interval:
0.9 < e < 1
eq (4.36)
Thus, re-performing the above analysis for e = 0.9 yields a total range for
absolute “best” and “worst” LNA noise figure performance:
10 * Log
NFLNA =
f
10 * Log
V
\ T()kBGLNAj
T0kBGINA
; fo r e « 1 (best case )
eq (4.37)
+ 0.1 ; fo r e
= 0.9 (worstcase)
101
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LNA noise figure was measured to vary between 0.602dB for best case to
0.95dB for worst case analysis.
4.6
T h re e -s ta g e
RF A m p lifie r
Design
The RF gain amplifier represents another essential entity of radiometer
systems receiver front-ends.
Commonly located after the LNA, the primary
function of the RF amplifier is to substantially increase the amplitude of signal
waveform originating from the BPF and LNA stages. In general, the objective of
the gain amplifier is to raise the low incoming power signal to a level readable by
the detector component. To this respect, most RF amplifier circuits consist of
multiple stage designs. It should be noted that in performing this amplification,
the amplifier component should not distort the waveform of the signal.
It thus
seems that the functionality of the RF amplifier is in many ways similar to that of
the LNA component.
However, contrary to the LNA however, in designing the
RF amplifier, electronic noise level does not represent an important design
consideration.
4 .6 .1
D e s ig n D e s c r ip t io n
The RF amplifier investigated for this effort is a 3-stage amplifier featuring
a linear power gain higher than 50dB across the operating frequency region. For
102
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convenience purposes, this design was implemented using same transistor part
as the LNA, ATF-34143. In addition, similarly to the LNA, the matching networks
design philosophy of the RF amplifier is entirely microstrip line based.
The
specifications of the amplifier are given in table-4.4.
T a b l e -4 .4 : R f -A m p l if ie r S p e c if ic a t io n s
Parameters
Targeted values
Frequency range
1 .4 -1 .4 2 7 GHz
Gain
> 50 dB
Input return loss
< -18 dB
Output return loss
< -18 dB
Input at P1dB
- 40 dBm
The layout of the realized multi-stage amplifier is illustrated in figure 4.19. The
design utilizes single-stubs for IMN and OMN.
Inter-stage isolation is
implemented using capacitors which are also part of the inter-stages matching
networks.
The input and output DC blocking capacitors have values of 56pF
while 12pF capacitors are used at inter-stages.
A 1 5 0 resistor is added in
series with the drain terminal of the first stage amplifier for stability purposes.
Also depicted in figure 4.19, are the grounding walls.
These walls serve to
prevent coupling effects resulting from close proximity of some stub lines. The
103
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vias used in the grounding walls are carefully placed at a distance A015 proximity
of one another (where X0 is wavelength at design center frequency).
ATF-34143
Grounding
walls
Figure 4.19- Microstrip Layout of 3-stages RF amplifier. Shown of the figure are the grounding
walls used to prevent coupling effects resulting from close proximity of some stub
lines.
104
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4 .6 .2
D is c u s s io n
of
M easured R esults
Although the RF amplifier circuit was manufactured, measurements were
not performed due to HP8510C VNA equipment malfunction. As a result, our
discussion is limited to simulated performance of the circuits as predicted by ADS
circuit simulation. Small signal characteristics of the amplifier are illustrated in
figure 4.20. It is observed that the design meets and even surpasses gain power
specification with a considerable margin.
The gain at center frequency of
1413MHz is simulated to be approximately 51 dB. The input and output matching
conditions of the circuit are depicted in figure 4.21 through representation of
simulated VSWR performance.
For unconditional stability characteristics, the
device must exhibits a K-factor greater or equal to unity across design frequency.
Stability performance of the amplifier is illustrated in figure 4.22. The simulated
performance of amplifier at 1dB compression point is illustrated in figure 4.23. It
is observed that amplifier exhibits compression at respectable input power level
of -31.5dBm.
105
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40
ml
freq=1r400GHz
dB(S(2.1)1=52.432
CM
m2
freq=1.427GHz
dB(S(2,1))=51.594
COCO CO
m CD CD
T 3 -O
T3
0-
-20
1.0
1.2
1.4
1.6
1.8
2.0
freq, GHz
Figure 4.20 S-parameters characteristics of the multi-stage amplifier as predicted by ADS circuit
simulator
m3
freq=1,400GHz
VSWR in=1.300
m4
freq=1,427GHz
VSWR in=1.417
3c
O|o:
CO
rr\
CO
>
4 -
m5
freq=1,400GHz
VSWR_out=1.724
m6
freq=1,427GHz
VSWR out=1.420
freq, GHz
Figure 4.21 Simplified schematic of VSWR. It is shown that across bandwidth of interest
(1400MHz - 1427MHz), the circuit displays adequate input and output matching
conditions.
106
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m7
freq= 1.400GHz
Stab -a c t!-1.980
.......
m8
freq=1.427GHz
StabFact1=2 222
o
ro
u_
2-
1.0
1.4
1.2
2.0
1.8
1.6
freq, GHz
Figure 4.22 Simulated stability performance of the multi-stage amplifier
P i n
C o m p r e s s i o n
- 3 1 . 5 0 0
-
1
0 4 2
m 4-
00
m.4
5040-
m 1
30m 2
2 0
-
100-60
- 5 5
-50
—45
- 4
0
-35
-30
-25
-20
P i n
Figure 4.23 Simulated amplifier performance at 1dB compression point. It is shown that at
compression, the circuit emulates almost 51 dB of gain. Compression condition
occurs at input power level greater or equal to -31,5dBm.
107
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4 .7 R e c e iv e r R F F ro n t-E n d In te g r a tio n and E v a lu a tio n
Upon confirming adequate performance of the individual circuit blocks
through measurements, integration of the front-end was conducted.
A design
protocol for realizing the microwave integrated circuit (MIC) of the receiver frontend was developed and is provided in figure 4.24. The ADS simulator is used as
the primary design environment for this integration. In the procedure, the filter is
represented by its S-parameter touchstone files exported from Ansoft Designer.
To ensure minimal deviation in performance of the circuit blocks, the optimization
of the integrated front-end is limited to tuning only the interconnecting lines at
component blocks interfaces.
A simplified representation of the integrated
receiver front-end is illustrated in figure 4.25. Both, LNA and RF amplifier appear
in the figure as sub-circuits contained inside design blocks. The final layout of
integrated receiver Front-end is provided in figure 4.26. In the design, the circuit
blocks are arranged in a configuration that maximizes the efficient real estate use
of the printed circuit board material.
Simulated S-parameters performance of the integrated receiver Front-End
is illustrated in figure 4.27. The simulated gain at center frequency of design is in
agreement with expectations. This observation indicates convincing evidence to
the validity of this integration methodology used since this value can readily be
determined by subtracting all losses contributions due to filters from total gain of
amplification stages. The noise figure characteristic of the receiver is illustrated
in figure 4.28.
108
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Re-Simulation of
BPF in planar EM
Ansoft Designer
Re-Simulation of
LNA and RF AMP
in ADS
Exporting
simulation data in
Touchstone format
Integration of all
system blocks into
single schematic
Optimizing response of receiver
module by tuning only inter­
blocks stubs and line widths
Generation
of layout
Export of gerber
files for
manufacturing
Figure 4.24 Flow chart of methodology of Receiver Front-End integration
1 09
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
B PF_Blo
?
Term
Terml
NUITF=1
Z=50Ohm
H
AM P_Block
Term
Ter m2
Num=2
Z = 5 0 0 tm
S2P
SNP1
S2P
SNP2
File^CAMyjdomairflDesiQner_projects'My_filter_s2p_data.s2p"
File=X:\Myj3omain\Designer_projects\MyJilter_s2pjlata.$2p"
O P T IO N S
S -P A R A M E T E R S
Options
Options 1
Temp= 27
S Param
SP1
Start= 1.0 GHz
TnorrF=27
V_RelTol=1e-6
V_AbsTol=1e-6V
Stops3.0GHz
Step=5.0MHz
MSub
MSUB
MSubl
H=50rrtl
ErsiO.2
M u rs i
Cords 1.0E+50
Hu=1.0&*-G33mm
l_RetTol=1e-6
l_AbsTol=1e-12 A
GiveAll Warning s=^es
WaxWarnings=10
T =35 um
TanDs0.0023
RoughsQmm
Figure 4.25 Simplified integrated Front-end circuit representation. The various sections of the
receivers are contained within sub-circuit blocks. The tuning interconnecting lines
are not shown.
110
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J J
l
0
M ic ro s trip
M ) W 0
ffl
|
e g - a |D
K if
0 " |
m
A fg fl
\
-r
|v ,s cond
i
P
Q
jfc
;i o
0
i
'K
r
il
ERI C
CH I K k
ND 0
su
DS DR
r e c e i v e r
COMSAHE,
M
Dee eraij e i
.
3 0 0
F r o n t - E n d
'
Figure 4.26 Representation of integrated receiver Front-end design layout. The components in
the circuits are arranged in a configuration that optimizes real estate use of the
printed circuit board material. Also shown in the figure are the grounding walls used
to isolate various sections of the design in preventing potential coupling effects.
111
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
rfrT
o—
0—
C\2
ml
freq=1.4‘
dB(S(2,1 ))=60.236
[m2
—i —i
3GHz
C\2 C\2 i —i
m 2.
minm
^ xl
o
m3
freq=1.413GHz
~
dB(S(1,1))=-12.253
—
1 . 0
1 . 2
1 . 4
f r e q ,
1 .6
1. 8
2 . 0
GHz
Figure 4.27 S-parameters characteristics of integrated receiver Front-End as predicted by ADS
circuit simulator.
112
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
10—
8
CQ
—
7 —
£
m4
....
freq=1,413GHz
nf(2)=0.607
3 —
2 —.
1 . 0
1 , 2
1
1 . 4
f r
e q
,
.
6
1
.
8
2 . □
G H z
Figure 4.28 Simulated noise figure performance of integrated receiver Front-End.
4 .8 S u m m a r y
In this chapter, a detailed description of a receiver RF front-end was
presented.
The receiver is deigned in support to ongoing research efforts to
develop a digital radiometer by COMSARE at Morgan State University. During
the discussion, specifications for the various circuit blocks constituting the
receiver were provided. Design methodology for each of the circuit components
was discussed and simulations and measured results (where it applied) were
examined. The components were integrated into single MIC circuit following a
procedure also discussed herein.
The realized receiver features a gain
113
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performance of approximately 60dB with an outstanding simulated noise figure of
0.61dB.
114
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C hapter 5
H a n d h e l d R a d io m e t e r D e p l o y m e n t
(F ie l d E x p e r im e n t s )
5.1 I n t r o d u c t io n
In this chapter, the application of the Handheld Microwave Radiometer
(HMR) previously described in Chapter 2 to surface soil moisture is outlined. The
evaluation of the instrument is performed through a series of field experiments.
The measurement protocol followed during the field experiments is provided in
this section of the document.
To fulfill the objective of the instrument as an
education outreach tool, the field experiments were two-phased. The first phase
consisted of collecting remote sensing data and determining the brightness
temperature of measured soil scenes.
Phase two consisted of the validation
stage, achieved here by collecting soil samples for gravimetric analysis.
115
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5.2 M e a s u r e m e n t P r o t o c o l D e v e l o p m e n t
The Handheld Microwave Radiometer (HMR) is an innovative instrument
capable of scientific soil moisture sensing.
readings displayed on a digital panel.
The instrument outputs voltage
These readings are proportional to the
amount of water content of the observed soil area hence, emissivity of the
surface.
For optimum performance when using the handheld microwave
radiometer to retrieved soil water content information, an easy to follow protocol
was developed.
students,
the
Since the ultimate users of the HMR tool will be grade-level
protocol
was
rendered
simple
enough
to follow
and
is
complemented with a data-logger sheet for manual recording of the data. In all,
proper usage of HMR follows a nine steps guideline entailing:
1.
Begin by completing the top of the “Handheld radiometer data-logger” sheet.
2.
Locate an area of soil to be examined and enter a description of its settings in the
“Handheld Radiometer data-logger” sheet.
3.
Mark the experiment area by placing for example, small flags at all four corners
to denote the different horizons.
4.
Prior to measuring selected soil area, perform calibration of the instrument by
aiming it vertically towards clear sky.
5.
Wait 5 seconds for the panel meter’s reading to settle at stable value. Record
this value in the data-logger.
6.
Now, proceed to the measurements by aiming the instrument towards the
selected soil area at 30° (tracked using protractor attached to right hand side of
panel).
116
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
7.
Maintain the instrument as steady as possible at constant height above ground.
Wait 5 seconds and record this value.
8.
Repeat steps 7 and 8 for each incidence angle from 30° continuing through the
angles of 40°, 50° and 60°.
9.
Prior to performing each measurement, repeat steps 4 and 5. to ensure proper
calibration of instrument.
The data-logger in itself, shown on table-5.1, allows the instrument user to
enter personal information. The user is also required to enter precise study site
location information including longitude and latitude GPS coordinates, exact time
and date of the experiment procedure. In addition, weather condition as well as
ambient temperature information on the ground must be provided.
117
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T a b l e -5 .1 : H a n d h e l d
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5.3 S o il M o is t u r e F ie l d E x p e r im e n t C a m p a ig n s
5 .3 .1
F ie l d E x p e r im e n t s -1
To fulfill the purpose of developing an educational tool the handheld
radiometer brightness T readings needed to be scientifically reliable.
As a proof
of concept to demonstrate the reliability of the instrument, preliminary field
experiments were performed during the summer of 2005. The experiment was
carried out at NASA Goddard Space Flight Center in Greenbelt, Maryland, USA.
The study site, located on a volleyball field near the recreation center, consisted
of sandy bare soil with two different moisture content values and free of
vegetation. The investigation consisted of two phases and was carried during
matinal hours in order to avoid radiation contribution due to solar illumination.
During phase-1, variations in radiometer output as function of incidence
angle were recorded. The look angles used for the observations were (30, 45,
60) degrees.
For practical purposes, natural targets were used as external
calibration targets. One such target was the sky which displays low-radiation at
L-band. A large contrast in voltage output was observed between sky radiation
and soil measurements. While this observation was inline with our expectation,
the recorded sky radiation (160K) was much higher than the theoretical value of
10 K at L-Band. It was later determined that cause of this observation was due
to large null offset (Vnuii = 12.3mV).
119
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Assessment of instrument stability was
performed throughout the
experiment, during external calibration steps, by pointing the instrument directly
toward clear sky. This procedure was repeated each time prior from recording
soil radiation data. Over the experiment period of 1 hour, a standard deviation in
measured radiation of only 2 K was recorded when observing the sky.
During phase-2, in situ soil samples were collected at 5cm depth profile
and processed at the Goddard Soil Lab to determine gravimetric soil water
content.
This phase of the procedure was done in accordance to Global
Learning and Observations to Benefit the Environment (GLOBE) gravimetric soil
moisture protocol [48]. For a given soil sample, volumetric water content can be
determined by computing the ratio of differences between its original mass, and
the corresponding dry mass of the sample.
Weight
Gravimetric soil moisture =
wet
- Weight
d ry
Weight
(5.1)
d ry
A total of 8 independent soil samples, respective to all cardinal orientation
(north, south, east and west) of the field, were collected of which, half
corresponded to dry soil with gravimetric moisture content values ranging from
0.018 g/g to 0.028 g/g and the remainder, to wet soil with moisture content
values 0.063 g/g to 0.095 g/g.
Regression analysis performed on the data
obtained from both investigations (phase-1 and phase-2) revealed that there was
a relationship between the sensed emission and the amount of water content.
120
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Figure-5.1 shows the sensitivity of the handheld radiometer measured radiation
to variations in water content on the bare sandy soil.
290
280 -
270 Q.
260 -
250 -
o>
240 -
230 4
0.00
0.02
0.04
0.06
0.08
0.10
0.12
Gravimetric water content [g/g]
Figure 5.1- representation of the L-band handheld radiometer measured brightness T and “in situ"
soil moisture at 5 cm depth. The two datasets display a linear relationship.
5 .3 .2
F ie l d E x p e r im e n t s -2
An educational outreach activity using HMR was conducted at the
Baltimore Freedom Academy, an urban high school located in Baltimore, MD,
USA. The study site (39.29016° N, 76.59616° W), located adjacent to the school,
was a baseball field with non-uniform grass cover.
The investigation team
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included 11 students with classification ranging from K-10 to K-12 levels. The
instructor team was composed of three graduate-level college students and one
teacher from the Baltimore Academy.
During the course of the activity, students were introduced to basic
concepts of remote sensing.
Specifically, the information divulged included
briefing students on impacts of surface soil moisture on weather patterns and
precipitations prediction.
Students were then taught the benefits of monitoring
soil moisture and how relatively simple much monitoring could be performed
using radiometers; in particular handheld radiometer. Following this brief tutorial
session, students were given an opportunity for “hands-on” experiment with
HMR. The measurements field experiment procedure was two-phased. The first
phase consisted of collecting remote sensing data and determining the
brightness temperature of measured soil scenes.
Phase two consisted of the
validation stage, achieved here by collecting soil samples for gravimetric
analysis. During this phase, soil samples were collected from each horizon and
processed at the Morgan State University Soil Lab to determine volumetric water
content.
The outreach activity lasted 70 minutes.
In the end, high remarks and
satisfactory feedbacks were received from students and the teacher who
expressed strong interest in conducting similar educational activity in the future.
122
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5.4 HMR R e s u lts
V a lid a tio n
The accuracy of HMR was assessed by benchmarking its output
brightness T values to published results.
As thus, work reported in [47] was
chosen to perform this assessment. Basis for this selection was the similarity in
experimental site conditions between [47] and that of the tests carried out under
this research during field experiments.
During the first field experiment described in section 5.3.1, the recorded
brightness temperatures over the study site for the various look angles were
found to be within range of some recently published values [47].
Table 5.2
compares corresponding brightness T of the handheld radiometer to previously
demonstrated instrumentation.
T a b le
5.2
- B e n c h m a r k in g o f H a n d h e ld
R a d io m e te r M ic r o w a v e R a d ia tio n O u t p u t s A g a in s t
R e c e n t l y P u b lis h e d R e s u lt s
Dry soil
267-276
0.018-0.028
250-270
0.02-0.04
Wet soil
238-256
0.063-0.095
160-180
0.18-0.2
5.5 S u m m a r y
In this chapter, the capability of HMR in detecting surface soil moisture
was demonstrated. This assessment was carried out through a series of field
experiments. A measurement protocol was developed for proper use of HMR.
123
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All experiments performed were two-phased; a remote retrieval of soil data using
HMR, and soil sample collection phase for gravimetric analysis. Data from both
phases were compared to each other to determine repeatability of the
instrument. Also, the accuracy of HMR was successively benchmarked by
comparing its output TB to that of other previously demonstrated instruments.
Some of the field experiments were conducted as part of educational outreach
efforts; performed as preliminary steps to highlight feasibility for potential
deployment of HMR into classroom environment.
124
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Chapter 6
Conclusions
6.1 S u m m a r y
of
R e s e a r c h C o n t r ib u t io n s
This research focused on the development of low-cost microwave
instrumentation for soil moisture application through remote sensing science. As
thus, this work introduced and demonstrated a revolutionary cost effective
handheld microwave radiometer featuring operation at L-Band. The instrument
described within this dissertation is to date, the most affordable and portable
microwave device fully capable of soil moisture studies at L-Band. This handheld
radiometer was designed in the effort to address the deficiency of NASA’s
mathematics, engineering and science personnel pipeline.
To that end, the
handheld radiometer will provide accessibility to remote sensing science to
higher education institutions and other organizations that would not have been
125
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otherwise able to afford the high instrumentation cost associated with the
science. . The developed instrument has a target cost of under US$200 and total
weight around 2 kg. The instrument will benefit the remote sensing community
by being a tool for educating students about passive microwave remote sensing.
In addition to the handheld radiometer, this work also contributed to
current research efforts ongoing within the center of microwave satellite and RF
engineering (COMSARE) at Morgan State University.
Precisely, through the
design and fabrication of a low-noise receiver RF Front-End (RF-FE) to be
integrated with a direct-sampling digital radiometer currently under development
at Morgan State University. The design of this RF-FE incorporates at its core: a
low-noise amplifier, an innovative highly compact bandpass filter and a 3-stage
RF amplifier. The realized receiver features a respectable gain performance of
approximately 60dB with an outstanding simulated noise figure of 0.61 dB.
6.2 F u t u r e R e s e a r c h O p p o r t u n it ie s
While goals of this research endeavor were successfully met, there still
remains some room for improvements. Notably on HMR, one area of possible
improvement would be addition of an automatic data-logger.
Currently, while
HMR can be operated by single user, manual recording of data leads to
requirement of two people in order to complete successful field experiment
126
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procedure.
As a result, adding an automatic data-logger will enhance overall
HMR experience.
The deployment of HMR into school environment is expected to continue
as part of the center of advanced microwave research and applications
(CAMRA)’s pilot-schools outreach efforts. There are several notable motivations
for this: 1) It could serve to show continuous initiative by CAMRA in cultivating
interest of young high-school students in pursuing higher-education into various
NASA related fields. 2) It could also potentially spur similar interest in current
CAMRA undergraduate students by enabling them to become more involved in
the deployment phase. For example, CAMRA undergraduate pipeline could be
used to carry on the mission of taking the HMR instrument for introduction into
the ATMO pilot-schools.
As for the receiver Front-End, dependant upon results from remaining
fabricated circuits measurements, a second iteration of the design could improve
a few areas. Specifically, matching networks could be tuned based on measured
data to improve input and output match and an inter-stage matching network
utilizing a more robust design philosophy could be developed to improve isolation
thus, enhancing overall circuit performance.
In addition, upon successful integration of receiver Front-End with digital
Back-End also under development within CAMRA, lessons learned from overall
design experience will be of valuable use in the ultimate pursuit of a radiometer
on-a-chip.
127
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A p p e n d ix A
Photographs of outreach activities
128
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Figure A1.1 - Photograph of group of students in process of collecting soil samples for gravimetric analysis.
The collection of soil samples was done in accordance to the previously demonstrated GLOBE protocol for
determining volumetric soil water content.
Figure A1.2 - Photograph of entire field experiment team during the educational outreach activity. Team
consisted of 11 students with classification ranging from K-10 to K-12, a high-school teacher and three
graduate-level college students. Activity highlighted the first deployment of HMR into classroom
environment.
129
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A p p e n d ix B
V itae
130
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V it a e
Eric Chikando was born in Victoria, Cameroon, on May 09, 1979 to
Emmanuel and Emilienne Chikando. He received his BSEE from Morgan State
University with high honors in May 2002. As a recipient of a CAMRA graduate
fellowship, he enrolled in the M. Eng/D. Eng graduate program at Morgan State
University during Fall 2002.
During his tenure as a graduate student, Mr.
Chikando held various internships, researches and faculty positions. In 2003, he
was a visiting research student at the Microwave Instrument and Technology
Branch (M.I.T.B) of NASA Goddard Space Flight Center located in Greenbelt,
Maryland.
At NASA, he assisted in the design and fabrication of RF
MicroElectroMechanical Systems (MEMS) switches at Ka-Band for low-noise
radiometer instruments.
In the fall of 2005, he joined Coppin State University
located in Baltimore, Maryland as an adjunct faculty member in the department of
mathematics and computer sciences.
In 2006, as a summer intern at Intel
Corporation based in Chandler, Arizona, he worked on the development and
implementation of small-form factor multi-band antennas for WiMAX and WiFi
applications. The greater part of his doctoral research work was performed at
M.I.T.B. of NASA Goddard Space Flight Center located in Greenbelt, Maryland
Eric Chikando expects to earn a D.Eng degree from Morgan State
University in May 2007. His work for the Doctorate of Engineering degree was
sponsored by the Center of Advanced Microwave Research and Applications
(CAMRA) at Morgan State University.
The experience gained as a graduate
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student was in the area of RF and microwave engineering. His doctoral research
focused on the development of low-cost microwave instrumentation for remote
sensing applications. Following is a selected list of Mr. Chikando publications.
1
M. Li, F. Pellerano, E. Chikando, and C. Nwabuelo, “RF MEMS for Microwave Instruments,"
NASA Goddard Space Flight Center Director’s Discretionary Fund annual report, pp.34-36,
2003.
2
J. Piepmeier, F. Cornelis, E. Chikando, and L. Hand, “Handheld Microwave Radiometer for
Education and Outreach,” NASA Goddard Space Flight Center Director’s Discretionary Fund
annual report, 2004.
3
E. Chikando, J. Piepmeier, A. Dujari, and C. White, “Handheld Microwave Radiometer for
Education and Outreach,” IEEE, International Geosciences and Remote Sensing Symposium
2005, Vol. 1, pp. 4 2 8 -4 3 0 .
4
E. Chikando, J. Piepmeier, E. Levine, and C. White, “Handheld L-Band Microwave
Radiometer,” IEEE, Microwave Radiometer specialists conference 2006, pp. 55 - 59.
5
E. Chikando, M. Reece, and C. White, “Ultra Wide-Stopband Planar Microstrip Bandpass
Filter Above Defected Ground Structure,” IEEE, Sarnoff 2007 (Accepted).
6
E. Chikando, J. Piepmeier, J. Whitney and C. White, “Handheld Microwave Radiometer
(HMR) into Classroom Environment -A n Introduction,” IEEE, International Geosciences and
Remote Sensing Symposium 2007 (Pending).
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