close

Вход

Забыли?

вход по аккаунту

?

Development of Advanced Microwave/Millimeter-Wave Multi-Antenna Systems on Liquid Crystal Polymer

код для вставкиСкачать
Development of Advanced Microwave/Millimeter-Wave Multi-Antenna Systems on
Liquid Crystal Polymer
By
Jia-Chi Samuel Chieh
B.S. (University of California, San Diego) 2004
M.S. (University of Southern California) 2007
DISSERTATION
Submitted in partial satisfaction of the requirements for the degree of
DOCTOR OF PHILOSOPHY
in
Electrical Engineering
in the
OFFICE OF GRADUATE STUDIES
of the
UNIVERSITY OF CALIFORNIA
DAVIS
Approved:
_____________________________________
Anh-Vu Pham, Chair
_____________________________________
Neville C. Luhmann, Jr.
_____________________________________
M. Saif Islam
Committee in Charge
2012
-i-
UMI Number: 3540488
All rights reserved
INFORMATION TO ALL USERS
The quality of this reproduction is dependent upon the quality of the copy submitted.
In the unlikely event that the author did not send a complete manuscript
and there are missing pages, these will be noted. Also, if material had to be removed,
a note will indicate the deletion.
UMI 3540488
Published by ProQuest LLC (2012). Copyright in the Dissertation held by the Author.
Microform Edition © ProQuest LLC.
All rights reserved. This work is protected against
unauthorized copying under Title 17, United States Code
ProQuest LLC.
789 East Eisenhower Parkway
P.O. Box 1346
Ann Arbor, MI 48106 - 1346
DEO GRATIAS
-ii-
Table of Contents
Dedication ........................................................................................................................... ii
Table of Contents ............................................................................................................... iii
List of Figures .................................................................................................................... vi
List of Tables .................................................................................................................... xii
Acknowledgements .......................................................................................................... xiii
Publications ....................................................................................................................... xv
Abstract ............................................................................................................................ xvi
Chapter 1
Introduction ..................................................................................................... 1
1.1
Survey of state-of-the-art Phased Arrays ............................................................ 3
1.2
Challenges and Motivations ................................................................................ 7
1.3
Liquid Crystal Polymer Benefits ........................................................................ 7
1.4
Dissertation Outline .......................................................................................... 10
Chapter 2
Lightweight Broadband Phased Array Module on LCP ............................... 13
2.1
Introduction ....................................................................................................... 13
2.2
Design of broadband passive components ........................................................ 17
2.3
Design of passive 8-Element Antenna Array.................................................... 25
2.4
LCP Manufacturing Process ............................................................................. 29
2.5
Active Array System Design and Interface ...................................................... 32
2.6
Measurement Results ........................................................................................ 35
2.7
Conclusions ....................................................................................................... 40
2.8
Acknowledgements ........................................................................................... 41
-iii-
Chapter 3
A Bidirectional X-Band Antenna Array on LCP .......................................... 44
3.1
Introduction ....................................................................................................... 44
3.2
Overview of Metamaterials for Antennas ......................................................... 47
3.3
Design of Metamaterial Inspired Wideband Slot Antenna ............................... 52
3.4
Design of an Bidirectional X-Band Antenna Array .......................................... 61
3.5
Measurement Results ........................................................................................ 65
3.6
Conclusions ....................................................................................................... 72
3.7
Acknowledgments............................................................................................. 73
Chapter 4
W-Band antennas for Phased Array applications on LCP ............................ 76
4.1
Introduction ....................................................................................................... 76
4.2
Materials considerations ................................................................................... 79
4.3
T-Junction Design ............................................................................................. 80
4.4
Substrate Integrated Waveguide Slot Antenna on LCP .................................... 83
4.5
Series fed Patch Array on LCP ......................................................................... 89
4.6
Measurement Scheme using Zero-Bias Diode Detector ................................... 96
4.7
Measurement Results ...................................................................................... 102
4.8
Conclusion ...................................................................................................... 106
4.9
Acknowledgments........................................................................................... 107
Chapter 5
Dual Polarized Scalable W-Band Phased Arrays on LCP .......................... 109
5.1
Introduction ..................................................................................................... 109
5.2
Aperture Coupled Dual Polarized Antenna Topology .................................... 111
5.3
Low Isolation Design ...................................................................................... 113
5.4
High Isolation Design ..................................................................................... 120
5.5
2 x 2 Antenna Array Tile Concept .................................................................. 126
-iv-
5.6
Antenna Prototype and Measurement Results ................................................ 132
5.7
Conclusion ...................................................................................................... 145
Chapter 6
Conclusion and Future Work ...................................................................... 150
Appendix A Verilog Code for FPGA ............................................................................ 153
-v-
List of Figures
Fig. 1.1 Single Antenna vs. Multiple Antenna Beamforming System ............................... 2
Fig. 1.2.1 Beam steering through coherent phase addition................................................. 3
Fig. 1.2.2. RF Phase Shifting Architecture. ........................................................................ 4
Fig. 1.2.3. LO Phase Shifting Architecture... ..................................................................... 5
Fig. 1.2.4. Digital Beamformer Architecture ...................................................................... 6
Fig. 1.4.1. Generic molecular structures of two types of LCP thermoplastic resin ............ 8
Fig. 2.1.1. Wave fronts for an AESA using (a) time delay (b) phase shifter.................... 14
Fig. 2.1.2. Array factor for (a) high fractional bandwidth (b) low fractional bandwidth . 16
Fig. 2.2.1. Comparison of VSWR for differing number of cascading sections across
frequency of interest. ......................................................................................... 18
Fig. 2.2.2. Photographs of the designed 2-18 GHz Wilkinson power combiner on LCP. 18
Fig. 2.2.3. Measured frequency response of thin film 100 Ohm resistor. ........................ 19
Fig. 2.2.4. Photograph of physical test setup used to make measurements ...................... 21
Fig. 2.2.5. Simulation and measurement results for S11. (b) S21, S23. (c) S22. (d)
Measured magnitude imbalance between output ports ...................................... 21
Fig. 2.2.6. Diagram of a folded Wilkinson power combiner on a multi-layer LCP
substrate ............................................................................................................. 22
Fig. 2.2.7. Photograph of folded multi-layer Wilkinson power combiner (a) top side (b)
bottom side (c) Photograph of equivalent planar implementation ..................... 23
Fig. 2.2.8. Simulation and Measurement results for S11 (b) S22 (c) S21 (d) S23 ........... 24
Fig. 2.3.1. Designed tapered slotted antenna element ...................................................... 25
Fig. 2.3.2. Measured azimuthal radiation pattern of individual antenna at (a) 32 GHz (c)
36 GHz ............................................................................................................... 26
Fig. 2.3.3. Designed tapered slotted antenna element ...................................................... 27
Fig. 2.3.4. Measured return loss of passive array ............................................................. 28
Fig. 2.3.5. Measured radiation pattern and gain ............................................................... 28
Fig. 2.4.1. Layer buildup of the LCP before lamination................................................... 30
-vi-
Fig. 2.4.2. Side view cross section of the multi-chip phased array module ..................... 30
Fig. 2.4.3. Bondwire compensation model ....................................................................... 31
Fig. 2.4.4. Predicted simulation results of bondwire compensation ................................. 32
Fig. 2.5.1. Serial to Parallel Interface ............................................................................... 33
Fig. 2.5.2. System level schematic of linear phased array................................................ 34
Fig. 2.5.3. Phased array module top side (top left), phased array module bottom side (top
right), LNA assembly (bottom left), phase shifter assembly (bottom right) ..... 35
Fig. 2.6.1. Test Setup ........................................................................................................ 36
Fig. 2.6.2. Measured radiation pattern at 34 GHz ............................................................ 38
Fig. 2.6.3. Measured 3-D antenna pattern for scan angles at (a) boresight (b) 15o (c) 30o
............................................................................................................................ 39
Fig. 2.6.4. Calibrated Gain Measurements ....................................................................... 40
Fig. 3.1.1. Typical Planar Antennas ................................................................................. 45
Fig. 3.2.1. Equivalent CRLH transmission line model ..................................................... 48
Fig. 3.2.2. Dispersion diagram of CRLH transmission line ............................................. 49
Fig. 3.3.1. Geometry of the proposed antenna (a) bottom conductor (b) top conductor .. 53
Fig. 3.3.2. Equivalent lumped element model of coupling element ................................. 54
Fig. 3.3.3. Simulated Co- and Cross-polarization radiation patterns at 11.5 GHz on (a) 5
mil (b) 10 mil (c) 15 mil (d) 20 mil LCP substrates .......................................... 56
Fig. 3.3.4. (a) Parametric study on ground plane size (b) simulated S11 for respective
ground plane sizes .............................................................................................. 58
Fig. 3.3.5. Co- and Cross-polarization patterns for various ground plane sizes ............... 58
Fig. 3.3.6. The fabricated prototype antenna (a) Front side (b) back side ........................ 59
Fig. 3.3.7. Measured and simulated return loss curves for proposed antenna .................. 60
Fig. 3.3.8. Measured and simulated normalized radiation pattern in the E and H planes 60
Fig. 3.3.9. Simulated surface current distribution at 11.5 GHz ........................................ 61
Fig. 3.4.1. Measured and simulated mutual coupling in the (a) H-Plane (b) E-Plane
orientation. Inset shows fabricated test structures for each orientation ............. 62
Fig. 3.4.2. 3D model of 16-element concurrent dual beam antenna array ....................... 63
Fig. 3.4.3. Simulated radiation patterns in the (a) E-Plane and (b) H-Plane .................... 64
Fig. 3.5.1. Fabricated 4x4 concurrent dual beam antenna array ....................................... 65
-vii-
Fig. 3.5.2. Measured and simulated return loss for 4x4 antenna array ............................. 66
Fig. 3.5.3. Measured radiation pattern in the elevation plane at (a) 10 GHz (b) 12 GHz 67
Fig. 3.5.4. 2D Plot of the radiation pattern in the elevation plane .................................... 68
Fig. 3.5.5. 3D Plot of the radiation pattern in the elevation plane .................................... 68
Fig. 3.5.6. Measured radiation pattern in the azimuth plane at (a) 10 GHz (b) 12 GHz .. 69
Fig. 3.5.7. 2D Plot of the radiation pattern in the azimuth plane...................................... 70
Fig. 3.5.8. 3D Plot of the radiation pattern in the azimuth plane...................................... 70
Fig. 3.5.9. Calibrated gain measurement .......................................................................... 71
Fig. 4.1.1. Oxygen absorption at various frequencies ...................................................... 77
Fig. 4.1.2. Potential Millimeter Wave High Bandwidth Backhaul ................................... 78
Fig. 4.3.1. T-Junction power combiner on LCP ............................................................... 81
Fig. 4.3.2. Simulated S-Parameters of the T-Junction combiner ...................................... 82
Fig. 4.4.1. Schematic of the SIW single slot antenna. Dimensions are: L = 3.4 mm, LTrans
= 0.98 mm, WTrans = 1.34 mm, LSlot = 1.2 mm, WSlot = 0.127 mm, WOffset = 0.722
mm, LOffset = 1.66 mm, DVia = 0.2032 mm, GVia = 0.6096 mm.......................... 85
Fig. 4.4.2. Return loss of SIW single slot antenna ........................................................... 85
Fig. 4.4.3. Schematic of the SIW eight-by-eight slot antenna array with corporate feed
network .............................................................................................................. 86
Fig. 4.4.4. Return loss of the SIW array varying the slot widths...................................... 87
Fig. 4.4.5. 3D E-Field distribution for a single longitudinal slice .................................... 88
Fig. 4.5.1. W-band 4 element series fed patch antenna array on LCP.............................. 90
Fig. 4.5.2. Simulated return loss of 4 element series fed patch array ............................... 90
Fig. 4.5.3. Simulated radiation pattern of 4 element series fed patch array ..................... 91
Fig. 4.5.4. W-band 16-element series fed patch array ...................................................... 92
Fig. 4.5.5. Simulated return loss of W-band 16-element series fed patch array............... 92
Fig. 4.5.6. Simulated radiation pattern of W-band 16-element series fed patch array ..... 93
Fig. 4.5.7. W-band 64-element series fed patch array ...................................................... 94
Fig. 4.5.8. Simulated return loss of W-band 64-element series fed patch array............... 94
Fig. 4.5.9 Simulated radiation pattern of W-band 64-element series fed patch array ...... 95
Fig. 4.6.1. Fully assembled W-band antenna array detectors (a) SIW array (b) series fed
patch array.......................................................................................................... 97
-viii-
Fig. 4.6.2. (a). Schematic of PBG LPF. Dimensions are: L1 = 0.71 mm, L2 = 0.81 mm,
W1 = 0.76 mm, L3 = 0.71 mm, W2 = 1.37 mm, L4 = 0.81 mm, W3 = 0.15 mm
(b). Sonnet simulated S-parameters ................................................................... 98
Fig. 4.6.3. Experimental measurement setup .................................................................... 99
Fig. 4.6.4. Physical setup used for radiation pattern measurements on the transmit side
.......................................................................................................................... 100
Fig. 4.6.5. Physical setup used for radiation pattern measurements on the receive side 101
Fig. 4.6.6. Antenna array under test inside the anechoic chamber ................................. 101
Fig. 4.7.1. Measured and simulated radiation patterns of the SIW array in the (a) Azimuth
(b) Elevation planes ......................................................................................... 103
Fig. 4.7.2. Measured sensitivity of SIW antenna array with diode detector................... 104
Fig. 4.7.3. Measured and simulated radiation patterns of the series fed patch array in the
(a) Azimuth (b) Elevation planes ..................................................................... 105
Fig. 4.7.4. Measured sensitivity of patch antenna array with diode detector ................. 106
Fig. 5.2.1. Conceptual diagram showing a W-band 2 x 2 antenna array tile with a chipon-board assembly for the MMIC.................................................................... 112
Fig. 5.3.1. Dual polarized aperture fed stacked patch on LCP/Kapton .......................... 113
Fig. 5.3.2. Antenna Geometries for various layers ......................................................... 114
Fig. 5.3.3. LCP/Kapton Stackup for Dual Polarized Aperture Fed Stacked Patch
Antennas .......................................................................................................... 114
Fig. 5.3.4. HFSS simulation model with independent polarization feedlines ................ 117
Fig. 5.3.5. Simulated return loss for both vertical (red) and horizontal (purple)
polarizations ..................................................................................................... 117
Fig. 5.3.6. Simulated isolation between orthogonal polarization ports .......................... 119
Fig. 5.3.7. Simulated vertical co- and cross-polarized radiation patterns in the
(a) azimuth (b) elevation planes ...................................................................................... 119
Fig. 5.3.8. Simulated horizontal co- and cross-polarized radiation patterns in the
(a) azimuth (b) elevation planes ...................................................................................... 120
Fig. 5.4.1. High isolation dual polarized aperture fed stacked patch on LCP/Kapton ... 121
Fig. 5.4.2. Antenna Geometries for various layers ......................................................... 122
-ix-
Fig. 5.4.3. Schematic of balun fed patch (a) isolation (b) primary mode (c) higher order
mode................................................................................................................. 123
Fig. 5.4.4. Simulated return loss for both vertical (purple) and horizontal (red)
polarizations ..................................................................................................... 124
Fig. 5.4.5 Simulated isolation between orthogonal polarization ports ........................... 125
Fig. 5.3.6. Simulated vertical co- and cross-polarized radiation patterns in the
(a) azimuth (b) elevation planes ...................................................................................... 126
Fig. 5.3.7. Simulated horizontal co- and cross-polarized radiation patterns in the
(a) azimuth (b) elevation planes ...................................................................................... 126
Fig. 5.5.1. 2 x 2 tile array (a) base line model (b) dual axis of symmetry model ........... 128
Fig. 5.5.2. Illustration of higher mode cancellation through symmetry (a) dominant (b)
high order mode ............................................................................................... 128
Fig. 5.5.3. HFSS simulation model for non-symmetric 2 x 2 tile (a) top (b) bottom side
.......................................................................................................................... 130
Fig. 5.5.4. HFSS simulation model for dual symmetric 2 x 2 tile (a) top (b) bottom side
.......................................................................................................................... 130
Fig. 5.5.5. Simulated radiation pattern at broadside (a) base line model (b) dual
symmetric model .............................................................................................. 132
Fig. 5.5.6. Simulated radiation pattern at 10º (a) base line model (b) dual symmetric
model................................................................................................................ 132
Fig. 5.5.7. Simulated radiation pattern at 30º (a) base line model (b) dual symmetric
model................................................................................................................ 133
Fig. 5.5.8. Simulated radiation pattern at 45º (a) base line model (b) dual symmetric
model................................................................................................................ 133
Fig. 5.6.1. PCB manufacturing process for LCP/Kapton antenna prototypes ................ 135
Fig. 5.6.2. Top view of a single antenna element prototype ........................................... 135
Fig. 5.6.3. Top view of a 8-element linear antenna array that has been misaligned ...... 136
Fig. 5.6.4. Test boards for the single antenna element for both the (a) horizontal (b)
vertical polarizations ........................................................................................ 137
Fig. 5.6.5. Block diagram of measurement setup ........................................................... 138
Fig. 5.6.5. Physical measurement setup .......................................................................... 138
-x-
Fig. 5.6.7. Close-up of measurement setup .................................................................... 139
Fig. 5.6.8. Measured and simulated H-plane radiation patterns in the vertical polarization
at (a) 95 (b) 80 (c) 105 GHz............................................................................. 141
Fig. 5.6.9. Measured and simulated H-plane radiation patterns in the horizontal
polarization at (a) 95 (b) 80 (c) 105 GHz ........................................................ 142
Fig. 5.6.10. Test boards for an 8-element array in both the (a) horizontal (b) vertical
polarizations ..................................................................................................... 143
Fig. 5.6.11. Measured and simulated H-plane radiation patterns in the vertical
polarization at (a) 95 (b) 80 (c) 105 GHz ........................................................ 145
Fig. 5.6.12. Measured and simulated H-plane radiation patterns in the horizontal
polarization at (a) 95 (b) 80 (c) 105 GHz ........................................................ 146
-xi-
List of Tables
Table 1.1 Comparison of Modern Wireless Standards ..................................................... 2
Table 2.1 Wilkinson Power Combiner Parameters ......................................................... 19
Table 3.1 Antenna Comparison ....................................................................................... 45
Table 3.2 Proposed Antenna Dimensions ....................................................................... 53
-xii-
Acknowledgements
First and foremost, I would like to sincerely thank my advisor Prof. Anh-Vu
Pham for his unwavering support through my doctoral studies. His level of energy and
passion is hard to match and what I have learned from him goes beyond academics. His
support has resulted not only in original research, but also opportunities to present my
research at international conferences all over the world. I look forward to many years of
friendship and professional collaborations with him.
I would also like to acknowledge and thank my dissertation committee members,
Prof. Neville Luhmann Jr., and Prof. Saif Islam for taking the time to be part of my
qualifying exam and for their questions and comments which have been valuable to my
research. Their expertise in the area of microwaves has been a valuable resource for my
studies at UC Davis.
This work wouldn’t be possible without the funding and support from a number
of sponsors. From the Boeing Company, I am grateful to Tim Lee for support and
valuable discussions on wideband passive components. From the AFRL (Air Force
Research Labs) I am grateful to Tom Dalrymple for his support for my work on phased
arrays on LCP. I am indebted to Brian Garber, Dave Kuhl, and Dr. Kunia Aihara for
assistance in testing my phased array at the AFRL. I will never forget the long hours
spent at WPAFB debugging the FPGA code, I am truly thankful to you. I am especially
grateful to Dr. Alex Pidwerbetsky and George Kannell from LGS Innovations for the
opportunity to do a summer internship in New Jersey. Their mentorship, support, and
vision resulted in a research collaboration that is a large part of this dissertation.
-xiii-
Special thanks to my great colleagues from MML: Dr. Kunia Aihara, Dr. Mark
McGrath, Dr. Andy Chen, Dr. Morgan Chen, Dr. Chi Law, Dr. Yiren Wang, Alexander
Stammeroff, Hai Ta, Cuong Nguyen, Will Chin, Jeffery Curtis, Binh Pham, and Van
Duong. I am grateful for all the valuable technical discussions we’ve had, and even more
grateful for your friendship that have made graduate school bearable. Special thanks also
to the Plasma Diagnostics group: Huan Liao, Xiangyu Kong, Tianran Liang, and Jiali
Lai. Thank you for being generous with your time and knowledge. To Cornelius Chin, I
am grateful for all the time we spent talking microwave.
My graduate studies wouldn’t be possible without the support network of friends.
To Mike Hong, thank you for being a willing participant on all our adventures and
especially for being willing to drive cross-country with me to New Jersey. To Cameron
Ashizawa, thank you and your family for always being there for me and showing me
Christ like love. To Stanley Hsu, you are awesome, I could not have asked for a better
roommate and friend. To my friends from the East Bay: Timothy Hwang, Payton Chu,
Jesse Chui, Philip and Phyllis Wang, David Yu and Eric Lee thanks for being you.
Finally, this work wouldn’t be possible without the love, support, encouragement,
and sacrifices made by my parents and family. My dad is the wisest man I know and I am
thankful for all the knowledge and advice he has shared with me over the years. My mom
has been a pillar of strength for the whole family and has continuously spoiled me even
through graduate school. To my sister Grace, I could not be more proud of you, you are
truly a beautiful person. My new brother Stephen, thank you for your support and kind
words. To both my grandmothers whose strength, love, and faithfulness have been
inspirational and influential.
-xiv-
Publications
Chieh, J.S.; Pham, A.V., Kannell, G.; Pidwerbetsky, A.; " A W-Band 8 x 8 Series Fed Patch Array Detector
on Liquid Crystal Polymer," Submitted to Antennas and Propagation Society International Symposium
(APSURSI) 2012 on 1/15/12
Chieh, J.S.; Pham, A.V., “A Bidirectional Microstrip X-Band Antenna Array on Liquid Crystal Polymer for
Beamforming Applications," Submitted to Transactions on Antennas and Propagation on 2/6/12
Chieh, J.S.; Pham, A.V.; T. W. Dalrymple, D. G. Kuhl, B. B. Garber, K. Aihara, “A Millimeter Wave
Phased Array Multi-Chip Module on Liquid Crystal Polymer," Submitted to Transactions on Advanced
Packaging on 7/6/11
Chieh, J.S.; Pham, A.V., A.V., Kannell, G.; Pidwerbetsky, A.; " A Low Cost 8 x 8 W-Band Substrate
Integrated Waveguide Antenna Array Detector on LCP," To be submitted to Letters on Antennas and
Propagation on 3/1/12
Chieh, J.S.; Pham, A.V.; , "A Bidirectional X-Band Antenna Array on Liquid Crystal Polymer," Antennas
and Propagation Society International Symposium (APSURSI), 2011 IEEE , vol., no., pp.1-4, 3-8 July 2011
Wang, Y.; Chieh, J.S.; Pham, A.V.; , "A Wideband and High Gain V-band EBG Patch Antenna on Liquid
Crystal Polymer," Antennas and Propagation Society International Symposium (APSURSI), 2011 IEEE ,
vol., no., pp.1-4, 3-8 July 2011
J.S. Chieh and A.V. Pham, Development of a Wide Bandwidth Wilkinson Power Divider on Multilayer
Organic Substrates, Microwave Opt Technology Letters, July 2010, Issue 52#7
J. S. Chieh, A. Pham, T. W. Dalrymple, D. G. Kuhl, B. B. Garber, K. Aihara, "A Light Weight 8-Element
Broadband Phased Array Receiver on Liquid Crystal Polymer," in IEEE International Microwave
Symposium(IMS), Anaheim, CA, May 2010
J.S. Chieh, A.V. Pham, T. Dalrymple, K. Aihara, D. Kuhl, B. Garber, “A broadband 8-Element Phased
Array Antenna Receiver on Liquid Crystal Polymer in the Ka-Band,” in Proceedings of GOMAC, Reno,
NV, March 2010
Chieh, J.-C.S.; Anh-Vu Pham; , "Development of a broadband Wilkinson power combiner on Liquid
Crystal Polymer," Microwave Conference, 2009. APMC 2009. Asia Pacific , vol., no., pp.2068-2071, 7-10
Dec.2009
Book Chapters
H. Ta, M.J. Chen, K. Aihara, A. Chen, J.S. Chieh and A.V. Pham, “LCP for Passive Components,” in “LCP
for Microwave Packages and Modules”, Cambridge Press, June 25th 2012.
M.J. Chen, K. Aihara, A. Chen, J.S. Chieh and A.V. Pham, “LCP for System Design,” in “LCP for
Microwave Packages and Modules”, Cambridge Press, June 25th 2012.
-xv-
Jia-Chi Samuel Chieh
April 2012
University of California Davis
Department of Electrical and Computer Engineering
Abstract
Multi-antenna systems are increasingly becoming more and more ubiquitous as
technology improves and has proven to be advantageous in many communication
applications. This dissertation addresses the challenges faced by multi-antenna systems in
a variety of applications with a wide range of operating frequencies from the X-Band
radio frequency (8 GHz – 12 GHz) all the way up to the W-Band (75 - 110 GHz)
millimeter wave frequency regime.
A novel wideband, light-weight, electronically controlled, linear 8-element
phased array antenna receiver module has been demonstrated on LCP (liquid crystal
polymer). Operating in the Ka-band (26.5– 40 GHz), this phased array module
demonstrates the concept of a scalable phased array utilizing high density integration
offered through the use of multi-layer LCP.
-xvi-
A novel wideband planar wide slot antenna with a metamaterial inspired reactive
coupling element has also been developed. Utilizing the bidirectional radiation pattern of
the single antenna, a concurrent dual beam antenna array is proposed and successfully
designed. This novel bidirectional array operates in the X-Band (8 - 12 GHz).
In addition, state-of-the-art W-Band (75 – 110 GHz) antenna arrays for phased
array applications are presented. A SIW (Substrate Integrated Waveguide) slot antenna
array and a series-fed patch antenna array are designed and successfully demonstrated.
Measurement techniques utilizing Zero-Bias detector beam-lead diodes are also
discussed.
Finally a truly scalable W-Band dual polarized, low cross polarization, phased
array system is described and proposed. Antenna topology choices are discussed with
respect to a scalable system level perspective. A multilayer LCP 2 x 2 tile array concept
is also presented. Finally, some prototype antennas and antenna arrays are presented with
simulation and measurement results.
-xvii-
1
Chapter 1
Introduction
Emerging state-of-the-art wireless applications require enhanced performance
which can be achieved through the use of multi-antenna systems. As demand for
additional functionality increases in wireless devices (i.e. streaming live video and
audio), the limits of traditional radio architectures are being reached and new topologies
must be adopted. The two major challenges encountered in modern wireless radio
systems are bandwidth and spatial diversity. As can be seen in Table 1.1, modern
wireless standards utilize larger channel bandwidths and/or multiple antennas topologies.
In the case of streaming high definition video, 2 standards have been proposed,
WiDi and WiGig, both of which take advantage of the unlicensed 7 GHz of bandwidth in
the V-band (60 GHz). Although a large bandwidth is achieved, transmission of
millimeter-wave signals are severely attenuated at the V-band due to oxygen absorption
[1], and therefore spatial directivity through the use of a multiple antenna beamformer is
2
necessary [2]. This is illustrated in Figure 1.1. Large bandwidths are not necessarily
always available, and for applications such as 802.11n and LTE, data throughput is
Table 1.1.
Comparison of Modern Wireless Standards
Standard
Max. Bandwidth Max. Data Rate
UWB
802.11n (4 antenna)
LTE (1 antenna)
WirelessHD (WiDi)
Wireless Gigabit Alliance (WiGig)
520 MHz
40 MHz
20 MHz
7 GHz
7 GHz
~ 80 Mbits/s
~ 1.1 Gbits/s
~ 86.4 Mbits/s
~ 25 Gbits/s
~ 7 Gbits/s
Fig. 1.1 Single Antenna vs. Multiple Antenna Beamforming System
enhanced through the use of MIMO (multiple-input multiple-output) using spatial
multiplexing. The use of multiple antennas in this instance also enhances reception
diversity, mitigating channel fading.
3
It is apparent that more and more future wireless applications will adopt multiple
antennas schemes. The main focus of this work is on advanced scalable multiple antenna
systems with an emphasis
asis on scalable designs.
1.2 Survey of State-of
of-the-Art Phased Arrays
The phased array antenna system offers the unique fundamental ability to
electronically scan an antenna beam, providing rapid beam steering and element failure
reliability. Applications
ns of the phased array are numerous and include radar [3],
[
SATCOM [4],
], microwave imaging [[5],
], and more recently in commercial W-PAN
W
60GHz
links [6].
]. Moreover, it has been widely reported in literature that the phased array can
improve sensitivity by a fact
factor
or of 10log(n) and reduce interfering signals by increasing
the spatial directivity [7],
], [[8].
A beam is electronically steered through control of the individual radiating
element so that the total signal coherently adds in phased to the direction of the desired
beam direction. Figure 1.2
1.2.1 demonstrates this concept.
Fig. 1.
1.2.1 Beam steering through coherent phase addition
4
Thus the path length difference can be defined as:
∆X=δsin θ
(1)
where ∆X is the path length difference, δ is the inter-element spacing, and θ is the beam
angle relative to the normal. If the frequency of the radiated signal is known, the one can
convert this path length to a phase difference. This can be defined as:
∆Φ=2π(∆X⁄λ)
(2)
where ∆Φ is defined as the phase difference. Clearly, the beam direction can be adjusted
by simply maintaining a phase difference between radiating elements.
Three main architectures have been implemented in the phased array system and
differ in the method used for phase shifting. In the more traditional architecture, phase
shifting is achieved in the RF (radio frequency) domain. This architecture is shown in
Figure 1.2.2
5
Fig. 1.2.2. RF Phase Shifting Architecture
Within the traditional architecture, many methods have been used to accomplish
the RF phase shift. A piezoelectric transducer (PET) phase shift method has been
demonstrated in [9], [10] and although achieves broadband operation, it has high voltage
requirements and the apparatus’ physical size limits the level of integration necessary. An
RF-MEMS phased shifter has been developed at the Ka-Band and offers a good solution
[11] as well as the popular ferrite phase shifters as described in [12].
An alternative architecture recently proposed and demonstrated by [13]-[15], is
through phase shifting in the IF domain. This is shown in Figure 1.2.3. The main benefit
of using this method is that a dedicated phase shifter is not necessary, reducing
complexity. In this case, the phase shift mechanism is built into the mixer and operates on
the same principles as a retroactive array. The drawback of using this method is that the
bandwidth is dictated by the mixer.
6
Fig. 1.2.3. LO Phase Shifting Architecture
A final architecture that is gaining momentum is the digital beamformer. In this
topology, the phase shift is done in the digital domain. Although this technique is elegant
in its simplicity, it requires digital signal processing which is computationally intensive
and therefore generally requires more power. This architecture is shown in Figure 1.2.4.
7
Fig. 1.2.4. Digital Beamformer Architecture
The active electronically scanned array (AESA) has seen renewed interest in
recent years in part because of the improvement of very large scale integration (VLSI) to
integrate and realize multiple T/R elements onto a single chip, providing a compact and
cost effective solution [16], [17]. However, as much as VLSI has allowed for the
minimization of the receiver system components, the principal component, the antenna
array itself, has a fundamental size limitation that trades performance, complexity and
cost with aperture size, a primary restriction to the level of integration that can be
achieved. Although recently in literature, a phased array transceiver with on-chip
antennas has been reported [18], the antenna performance (i.e. gain, directivity, and
bandwidth) is limited because of its size and the integrated circuits’ lossy substrate. A
novel lens structure had to be implemented to recover performance.
8
1.3 Challenges and motivations
Modern day phased arrays can have tens to hundreds to thousands of elements.
Transmit/Receive (T/R) modules are the method of choice when building a scalable
phased array. To maintain scalability in phased array antenna systems, two criteria need
to be met. Firstly, thin light weight miniaturized modules are necessary. Secondly, these
modules need to have high density integration using 3D technology. Brick module
architectures traditionally suffers from complicated interconnects, which result from the
complex digital control circuitry, and are typically extremely bulky and heavy when
conventional materials such as Duriod or LTCC is used [19].
What is truly needed is a material that light weight, low in density, has the ability
to support multi-layers with a vast assortment of interconnect options, and low in loss up
to the millimeter wave regime. Much of this work explores the possibility of integrating
components on the sub-array level directly onto a low-cost and flexible substrate, mainly
LCP, to provide lightweight structures for potentially large arrays.
1.4 Liquid Crystal Polymer Benefits
LCP is a thermoplastic, an aromatic polymer with electrical and mechanical
properties which are superior to similar materials such as Polyimide or Kapton. LCP
films are developed typically with pelletized LCP resin. The molecular structures of two
generic LCP thermoplastic resins [20] are shown in Fig. 1.4.1.
9
(a)
(b)
Fig. 1.4.1. Generic molecular structures of two types of LCP thermoplastic resin
Fig. 1.4.1a describes
ibes the LCP resin that has the attributes of a high melting point
temperature, in the ranges of 300
300-350oC. Fig. 1.4.1b shows the LCP resin of a lower
melting point temperature material, in the ranges from 200
200-250oC.
When compared to a similar material ssuch
uch as Polyimide the main advantages of
LCP is that it has 1) Lower permittivity and loss tangent that re
remain
main constant with
humidity. 2) Adjustable coefficient of thermal expansion (CTE). 3) Chemical resistance.
4) Dimensional
ional stability, and finally 5) Thermo-plasticity,
plasticity, which allows for direct heat
lamination without the need for additional adhesives.
Additionally, when compared to ceramic materials such as low temperature coco
fired ceramics (LTCC) or high
high-temperature co-fired
fired ceramics (HTCC) it is more cost
c
10
effective. The reason for this is because LCP can be manufactured using typical PCB
processes, which yield panel sizes that are 5-10 times the size of a typical ceramic board.
This allows for a cost effective solution for volume manufacturing.
Finally, for antenna applications, LCP is a great candidate material because it has
superb electrical performance which has been demonstrated up to the W-Band [21], [22].
Its permittivity is much lower than that of LTCC which in turn means that antenna
efficiencies can be increased. Also, with the ability to design antennas on very thin
layers, substrate modes prevalent in the millimeter-wave regime can be suppressed. LCP
can enable advanced phased array antennas with the ability to support multiple layer
boards on a homogenous substrate media. Moreover, LCP is an organic substrate that is
mechanically strong yet light and flexible; great qualities for space based applications. It
is demonstrated in this work that LCP provides a convenient platform for mechanically
flexible electronics. Passive antenna structures can be designed directly into the LCP
build. Active semiconductor chips can be packaged into this platform which enables
potentially large scalable systems. This dissertation will demonstrate advanced research
based on LCP builds that extend electrical performance and mechanical functionality of
today’s antenna array modules.
11
1.5 Outline of Dissertation
This dissertation presents the design and development of advanced antenna arrays
and phased array modules on LCP. The electrical designs of antenna elements, passive
components such as power combiners, chip-on-board assemblies for MMICs are studied.
Chapter 2 will introduce a millimeter-wave multi-chip phased array receiver module in
the Ka-Band (32-37 GHz) built on LCP. Chapter 3 will discuss a bidirectional microstrip
X-band antenna array on LCP for concurrent beamforming applications. Chapter 4
introduces novel antenna array detectors in the W-Band (75-110 GHz) built on an LCP
substrate. Chapter 5 investigates the possibility of creating a truly scalable dual-polarized
W-Band phased array system on LCP which include the packaging of active MMICs.
Chapter 6 includes summary and conclusion of this dissertation, and future work that will
further improve advanced phased arrays on LCP.
12
References
[1] Yujiri, L.; Shoucri, M.; Moffa, P.; , "Passive millimeter wave imaging," Microwave
Magazine, IEEE , vol.4, no.3, pp. 39- 50, Sept. 2003
[2] A. M. Niknejad and H. Hashemi (editors), mm-Wave Silicon Technology: 60 GHz and
Beyond, Springer, 2008
[3] E. Brookner, Practical Phased-Array Antenna Systems. Norwood, MA: Artech
House, 1991
[4] Kunath, R. R.; Lee, R. Q.; Martzaklis, K. S.; Shalkhauser, K. A.; Downey, A. N.;
Simons, R.. , “A Vertically integrated Ka-band phased array antenna,” International
Communication Satellite Systems Conference 14th 1992, AIAA, Mar. 22-26, 1992
[5] B. D. Steinberg, Microwave imaging with large antenna arrays, NY: WileyInterscience, 1983
[6] L. Caetano, 60 GHz Architecture for Wireless Video Display, SiBEAM [ONLINE],
http://www.sibeam.com/whtpapers/60_GHz_for_WirelessHD_3_06.pdf
[7] Hajimiri, A.; Hashemi, H.; Natarajan, A.; Guan, X.; Komijani, A., "Integrated Phased
Array Systems in Silicon," Proceedings of the IEEE , vol.93, no.9, pp.1637-1655, Sept.
2005
[8] Parker, D.; Zimmermann, D.C., "Phased arrays - part 1: theory and
architectures," Microwave Theory and Techniques, IEEE Transactions on , vol.50, no.3,
pp.678-687, Mar 2002
[9] Tae-Yeoul Yun; Kai Chang, "A low-cost 8 to 26.5 GHz phased array antenna using a
piezoelectric transducer controlled phase shifter," Antennas and Propagation, IEEE
Transactions on , vol.49, no.9, pp.1290-1298, Sep 2001
[10] Lu Yang; Domier, C.W.; Luhmann, N.C., "Ka-Band True Time Delay E-plane Beam
Scanning and Broadening Phased Array System using Antipodal Elliptically-Tapered
Slot Antennas," Antennas and Propagation Society International Symposium 2006,
IEEE , vol., no., pp.2213-2216, 9-14 July 2006
[11] Hacker, J.B.; Mihailovich, R.E.; Moonil Kim; DeNatale, J.F., "A Ka-band 3-bit RF
MEMS true-time-delay network," Microwave Theory and Techniques, IEEE
Transactions on , vol.51, no.1, pp. 305-308, Jan 2003
[12] Parker, D.; Zimmermann, D.C., "Phased arrays-part II: implementations,
applications, and future trends," Microwave Theory and Techniques, IEEE Transactions
on , vol.50, no.3, pp.688-698, Mar 2002
13
[13] Ramant, S.; Barker, N.S.; Rebeiz, G.M., "A W-band dielectric-lens-based integrated
monopulse radar receiver ," Microwave Symposium Digest, 1998 IEEE MTT-S
International , vol.2, no., pp.517-520 vol.2, 7-12 Jun 1998
[14] Ramant, S.; Barker, N.S.; Rebeiz, G.M., "A W-band dielectric-lens-based integrated
monopulse radar receiver ," Microwave Symposium Digest, 1998 IEEE MTT-S
International , vol.2, no., pp.517-520 vol.2, 7-12 Jun 1998
[15] Xiang Guan; Hashemi, H.; Hajimiri, A., "A fully integrated 24-GHz eight-element
phased-array receiver in silicon," Solid-State Circuits, IEEE Journal of , vol.39, no.12,
pp. 2311-2320, Dec. 2004
[16] Hashemi, H.; Xiang Guan; Komijani, A.; Hajimiri, A., "A 24-GHz SiGe phasedarray receiver-LO phase-shifting approach," Microwave Theory and Techniques, IEEE
Transactions on , vol.53, no.2, pp.614-626, Feb. 2005
[17] Kwang-JIn Koh; Rebeiz, G.M., "An X- and Ku-Band 8-Element Phased-Array
Receiver in 0.18-um SiGe BiCMOS Technology" Solid-State Circuits, IEEE Journal of,
vol.43, no.6, pp. 1360-1371, June. 2008
[18] Babakhani, A.; Xiang Guan; Komijani, A.; Natarajan, A.; Hajimiri, A.; , "A 77-GHz
Phased-Array Transceiver With On-Chip Antennas in Silicon: Receiver and
Antennas," Solid-State Circuits, IEEE Journal of , vol.41, no.12, pp.2795-2806, Dec.
2006
[19] N. Fourikis, Advanced Array Systems, Applications and RF Technologis, CA;
Academic Press, 2000
[20] Balde and W. John, Foldable Flex and Thinned Silicon Multichip Packaging
Technology. Norwell, MA: Kluwer, 2003
[21] Thompson, D.; Kirby, P.; Papapolymeron, J.; Tentzeris, M.M.; , " W-band
characterization of finite ground coplanar transmission lines on liquid crystal polymer
(LCP) substrates," Electronic Components and Technology Conference, 2003.
Proceedings. 53rd , vol., no., pp. 1652- 1655, May 27-30, 2003
[22] Smith, S.L.; Dyadyuk, V.; , "Measurement of the dielectric properties of Rogers
R/flex 3850 liquid crystalline polymer substrate in V and W band," Antennas and
Propagation Society International Symposium, 2005 IEEE , vol.4B, no., pp. 435- 438 vol.
4B, 3-8 July 2005
14
Chapter 2
Lightweight Broadband Phased Array Module on LCP
This chapter explores the possibility of integrating components on the sub-array
level directly onto a low-cost and flexible substrate, mainly LCP, to provide lightweight
structures for potentially large arrays. More specifically a 8-channel broadband phased
array receiver hybrid module is built on a LCP substrate and operates in the Ka-Band
(32-37 GHz). The module includes a broadband antenna aperture, corporate power
combining network, chip-on-board MMICs, and digital control circuitry. The resulting
phased array module achieves ±30̊ beam steering capability across a 5 GHz bandwidth
with less than 6.75̊ of beam squint. The total size of the array is 100 mm x 100 mm with a
board thickness of 16 mils and a total weight of 12.3 grams. This work demonstrates LCP
as an enabling technology for light weight, high density, large scale phased array antenna
systems.
The phased array is named aptly that because of the predominant use of a phase
shifter to steer the direction of the beam. Ideally a frequency independent time delay
15
would be used to sum the signals from each antenna creating a linear wavefront as shown
in Fig. 2.1.1a.. For sinusoidal signals, it is well known that a phase shift is sufficient,
however because phase shifters operate with a 360o modulus, it creates a discontinuous
di
wavefront as shown in Fig. 2.1.1
2.1.1b.
b. Although a good approximation at narrow
bandwidths, the phase shift is independent of frequency, resulting in spatial distortion of
wideband signals. The distinction between the need for a TTD rather than a phase
ph
shifter
has been documented in [[1]] and explained in terms of intersymbol interference (ISI).
(a)
(b)
Fig. 2.1.1
2.1.1. Wave fronts for an AESA using (a) time delay (b) phase shifter
In an ideal situation using time delay elements, the relationship between the phase and
the frequency is
(3)
(4)
As can be seen, in the ideal relationship, although the time delay is independent of
frequency, the phase is not. This however, is not the ca
case
se for the phase shifter. To see
16
the effects of the phase shift approximation, one has to only look at the array bandwidth
as described in [2].
It is well known that the array factor for isotropic sources is defined as
, (5)
where the maxima occurs at !"# $ !"#$ . If we continue to set f=fo and using the
taylor expansion approximation
!"# !"#$ ≈ $ &'!$
(6)
we can determine the following relationship.
∆&'!$ 0
(7)
Now, setting θ=θo we can determine the following relationship.
∆!"#$ ∆&'!
(8)
Using (5) and (6), a fundamental relationship between frequency, bandwidth, and beam
shift can be determined.
17
∆
∆ * + ,#
(9)
This relationship dictates the bandwidth of the phased array and shows that for a given
bandwidth, depending on the frequency, the scan angle of the array will shift, this is
also known as beam squint.
Since beam squint is dependent on the fractional bandwidth, it can be seen that
for a given bandwidth, beam squint will always be more severe at lower frequencies.
Figure 2.1.2 illustrates this point with 2 simulated 7-element arrays steered to 15o.
Figure 2.1.2a shows an array with a center frequency of 34.5 GHz operating at 32 GHz.
Figure 2.1.2b shows an array with a center frequency of 4.5 GHz operating at 2 GHz.
Clearly it can be seen that using a phase shifter at a low fractional bandwidth can result
in the beam shifting away from the desired direction; this is beam squint. It can be seen
that over the same bandwidth, a larger beam squint is exhibited at the lower operating
frequency, and a TTD is necessary. However, it can be seen that the beam squint
exhibited at 32 GHz is negligible. For this reason, at the Ka-band operating with a 5
GHz instantaneous bandwidth, a phase shifter can be appropriate. For large arrays, a
TTD at the subarray level can provide a compromise to restore performance [3].
Fig. 2.1.2. Array factor for (a) high fractional bandwidth (b) low fractional bandwidth
18
2.2 Design of broadband passive components
Power combining and dividing techniques are readily found in many of today’s
cutting edge technology. With the introduction of GaN devices, broadband power
combing to achieve high output powers is very attractive [4]. It is also useful in many RF
receiver architectures to divide the in-phase and quadrature channels [5]. Many state-ofthe-art microwave hybrid designs are complex with multi-layers. System integration is
critical and much research work has been done to show that LCP is an ideal candidate for
the integrated millimeter wave needs for the future.
The Wilkinson power divider/combiner was first conceived by Wilkinson [6] and
later broadband characterization was done by Cohn [7]. In recent works, researchers have
fabricated a single stage Wilkinson combiner on LCP in the V-band and W-band [8].
Their designs are broadband by default because of the high fractional bandwidth
associated with a high center frequency. Many applications, such as UWB (3.1-10.6
GHz) require wide bandwidth, but at a much lower center frequency, a more difficult
task. This can only be done by cascading sections, as described by Cohn. In this chapter,
two state-of-the-art broadband Wilkinson power combiners with integrated resistors are
presented that operate from 2-18 GHz covering from the S-band to the Ku-band.
In Cohn’s original paper, bandwidth is enhanced by cascading quarter wavelength
transmission lines optimized by using the Chebyshev polynomial. The Chebyshev
multisection matching transformer optimizes the bandwidth at the cost of ripples in the
passband. In this work, a binomial multisection matching transformer is used to ensure a
maximally flat passband and also to simply calculations. As shown in Figure 2.2.1, the
19
2.2
N=1
N=3
N=5
N=7
2
VSWR
1.8
1.6
1.4
1.2
1
0
5
10
15
20
Frequency (GHz)
Fig. 2.2.1. Comparison of VSWR for differing number of cascading sections across frequency of interest.
Fig. 2.2.2. Photographs of the designed 2-18 GHz Wilkinson power combiner on LCP
cascading of quarter wavelength sections can increase the bandwidth dramatically. A 7stage planar Wilkinson is first demonstrated. Figure 2.2.2 shows the designed planar
Wilkinson on LCP.
Values for the resistors affect both the isolation between output ports and the
output return loss. Network theory can be applied to solving for appropriate values for the
resistors. For values of N=2, an analytical approach can be taken, but greater than that, an
iterative approach is necessary. For this work, the optimization function in Agilent’s
Advanced Design System (ADS) was used to solve for the value of the resistors to
20
optimize the output return loss. Equally, optimization of the isolation between output
ports could also be applied. Table I. shows calculated values for each cascading section
and the optimized values for the resistors used.
Test structures for resistor characterization were also manufactured. The primary
goal for the resistor test structures was to measure and insure broadband response of the
thin film resistors. The measured response of a 100 Ohm test resistor is shown in Figure
2.2.3 and confirms a broadband response and shows no more than a 1% deviation from
the design value throughout the frequency band.
Table 2.1.
Wilkinson Power Combiner Parameters
S(1,1)
100 Ohm Resistor
freq (2.000GHz to 20.00GHz)
Fig. 2.2.3. Measured frequency response of thin film 100 Ohm resistor
21
In the frequency band of 2-18 GHz, a quarter wavelength transmission line at the
center frequency of 10 GHz is 30 mm. Clearly size reduction becomes an important issue.
Using a traditional linear architecture such as in [9], the parasitics involved with
connecting the common mode resistors can become problematic. Techniques have been
suggested in [10] to overcome the transmission line effects of the resistor. Rather,
circular sections as suggested in [11] are utilized. To the authors knowledge this is the
first implementation using a cascaded topology on microstrip with integrated thin film
resistors.
Integrated thin film resistors can also be beneficial for microwave and millimeter
wave designs. Surface mount components can add unexpected parasitic from both the
packaging and the solder necessary to mount the devices. If a high sheet resistance is
chosen for the thin film resistors, the size of the resistors can be kept to a minimum. In
this design a 100 Ω/ NiCr resistor from Ticer Technologies was used.
The designed Wilkinson power combiner was manufactured on a 12mil thick LCP
substrate with 0.5-oz. copper. Since the device is a 3-port network, either a 4-port
network analyzer can be used, or the 3rd port needs to be terminated with a broadband 50
Ω load, the latter of which was used. Figure 2.2.4 shows the physical test setup used to
make measurements. A SOLT calibration was used in conjunction with an Agilent
E8361A Vector Network Analyzer for all measurements.
Simulation results for loss,
input and output return loss, and isolation were done using Sonnet EM software and
Ansoft HFSS, as shown in Fig. 1. The measured results from the 2-18 GHz Wilkinson
power combiner are shown in Figure 2.2.5.
22
Fig. 2.2.4. Photograph of physical test setup used to make measurements
Fig. 2.2.5. Simulation and measurement results for S11. (b) S21, S23. (c) S22. (d) Measured magnitude
imbalance between output ports.
23
As can be seen, the designed Wilkinson combiner has a broadband response. The
measured input and output return loss all exhibit broadband characteristics having a
VSWR of better than 1.6:1. The measured excess insertion loss never exceeds 1.1dB
across the full band. The isola
isolation is also better than -14dB,
14dB, however, suffers some
deviation from simulation. This is due to the variability of resistors values when the
feature sizes approach the minimum line widths that a manufacture can etch. A simple
solution would be to constrain resistor sizes to double that value to ensure consistent
resistor values. Finally, the measured magnitude imbalance was better than 0.46 dB
across the entire bandwidth of interest, from 22-18 GHz.
Although the previous design demonstrates the ability to bu
build
ild microwave circuit
components on LCP with good performance, oftentimes miniaturization is equally
important. A folded Wilkinson power divider proposed in [[12]] was implemented on a
GaAs substrate with a bandwidth ratio of 3:1, from 15 to 45 GHz. A second design is
demonstrated utilizing the multi
multi-layer
layer ability of LCP. This design is a folded compact
Wilkinson divider that is the most compact Wilkinson divider with the highest bandwidth
ratio to date. Figuree 2.2.6 shows a cross section of the multilayer design
sign as well as a 3D
rendition of the folded Wilkinson.
Fig. 2.2.6. Diagram of a folded Wilkinson power combiner on a multi
multi-layer
layer LCP substrate
24
As shown in Figure 1, four circular sections of quarter wavelength transmission
lines are printed on top of the LCP substrate and the other three sections on the bottom
side. A metal sheet is inserted in the center of the LCP board to provide a ground plane
for the top and bottom circular sections. We employed Ansoft’s High Frequency
Structure Simulator [13]] to design and optimize the vias used to connect sections of the
folded structure. The vias are 8 mil in diameter with 15 mil annular pads.
Figure 2.2.7a,b show the folded multi
multi-layered
layered power combiner top side and
bottom
tom side respectively. Figure 2.1.9c shows a equivalent planar implementation. The
planar Wilkinson power divider is 16.3 mm long while the folded circuit is only 9.3mm
long, resulting in a size reduction of 43%. Port 2 and 3 of the three port circuit is denoted
by its symmetry.
Fig. 2.2.7. Photograph
ograph of folded multi
multi-layer
layer Wilkinson power combiner (a) top side (b) bottom side (c)
Photograph of equivalent planar implementation
25
Figures 2.2.8a,b shows a measured VSWR for both input and output ports of
better than 1.6:1 from 2 to 18 GHz. The measured excess insertion loss, shown in Figure
2.1.10c, never exceeds 1.6dB and the measured isolation, shown in Figure 2.1.10d, is also
better than -12dB. Simulation results for insertion loss, input and output return loss, and
isolation were obtained using Ansoft’s HFSS software. The measured input and output
return loss closely matches with the simulation data. The insertion loss also exhibits a
close comparison to the simulation data. The isolation, however, suffers some deviation
from simulation. This is due to the variability of the resistor values when the feature sizes
approach the minimum line widths.
Fig. 2.2.8. Simulation and Measurement results for S11 (b) S22 (c) S21 (d) S23
26
2.3 Design of passive 8-element Antenna Array
In this design, an anti-podal tapered slot antenna is used because of its unique
combination of multi-octave bandwidth [14], [15], [16] and directive radiation pattern.
Figure 2.3.1 shows the designed antenna with its associated dimensions. It is well known
that the tapered slot antenna demonstrates optimal performance when the effective
substrate thickness, - √/0 1, is between 0.005λo and 0.03λo [17]. In this
design, a thickness of 12 mils was used, which meets this criterion. A 3 sectioned
Chebyshev transformer was used at the input of the antenna so that the coupled line balun
could be as streamlined as possible. Measured normalized radiation patterns are shown in
Figure 2.3.2. As can be seen, there is a slight beam tilt to the pattern at 32 GHz which can
cause an asymmetrical amplitude taper across an array aperture, mainly due to the
unequal geometry of the slot-line balun [18]. This can be remedied by introducing
asymmetries in the antenna fin to compensate.
Fig. 2.3.1. Designed Tapered Slotted Antenna Element
27
(a)
(b)
Fig. 2.3.2. Measured azimuthal radiation pattern of individual antenna at (a) 32 GHz (c) 36 GHz
The design of the feed network is critical for beamforming arrays. The
T Wilkinson
power combiner is especially well suited because it has good channel to channel
isolation, preventing re-radiation
radiation from the receiver. In this design, a multiple sectioned
Wilkinson power combiner with integrated thin film resistors was used. Using
U
the
corporate feed network, a passive tapered slot antenna array was designed, manufactured,
and measured. To satisfy the criterion for no grating lobes, an antenna element spacing of
185 mils was used, which corresponds to 0.5
0.5λ at 32 GHz and 0.58λλ at 37 GHz. This
corresponds to a maximum scan angle with isotropic sources of 800 at the 32 GHz and
28
46o at 37 GHz. The designed passive antenna array and test setup is shown in Figure
2.3.3 and the measured radiation patterns are shown in Figure 2.3.4. The measured return
loss is shown in Figure 2.3.5. and shows a VSWR of better than 1.9:1 from 30-50 GHz.
Radiation pattern measurements were made inside a compact range, and gain
measurements were made with a standard gain horn antenna. Simulations were completed
in HFSS. As can be seen, the measured sidelobes are asymmetric, an indication of phase
or amplitude imbalance gradient across the aperture. The antipodal tapered slot antenna is
especially sensitive to the balanced input transition. The measured HPBW is about 28o
and the maximum measured gain is 11.5 dBi at 38 GHz.
Fig. 2.3.3. Designed Tapered Slotted Antenna Element
29
15
32 GHz
34 GHz
36 GHz
38 GHz
Simulation at 34 GHz
10
0
-5
-10
-15
-20
-80
-60
-40
-20
0
20
Degree
40
60
80
Fig. 2.3.4. Measured Return Loss of Passive Array
0
-5
RETURN LOSS (dB)
Magnitude (dBi)
5
-10
-15
-20
-25
-30
-35
-40
30
32
34
36
38
40
42
44
FREQUENCY (GHz)
Fig. 2.3.5. Measured radiation pattern and gain
46
48
50
30
2.4 LCP Manufacturing Process
Many modern phased array modules use a tile- based array architecture. Essential
to reducing the complexity of the tile-based array is the integration of all control
distribution networks onto the subarray itself. For this, a multilayer board is paramount
and the advantages of using a cost effective multilayer organic substrate is clear.
LCP is a great candidate primarily because its lamination process lends itself well
to commercially available PCB (printed circuit board) processes requiring no additional
special processing steps. LCP is commercially available through Rogers in two forms, as
a core layer and as a bondinglayer. ULTRALAM 3850 forms the core layer and has 0.5
Oz. of double clad copper. ULTRALAM 3908 forms the bonding layer. It is important to
note that although a pre-preg is used, the substrate remains fully homogenous.
Additionally, in this design, the top core layer has an additional 270 Å layer of Ticer
resistive foil. As shown in Figure 2.4.1, the stackup is arranged in an interleaved fashion,
alternating between core and bonding layers. This is done to ensure uniform lamination.
Figure 2.4.2 shows the side view cross-section of the LCP stackup. Both through vias and
blind vias are utilized in this design. Vias have a drill diameter of 8 mils and a annular
pad of 16 mils.
Hard lamination techniques are used, where optimal bonding is achieved through
uniform temperature gradient across the board during the high temperature portion. The
temperature must be controlled very precisely as the lamination is sensitive to
temperature fluctuations. Typically, two temperature plateaus are utilized to achieve high
temperature uniformity. First the LCP is slowly ramped up to the first temperature
plateau, a nominal 260o C. It is kept at this temperature for at least 30 minutes. Then the
31
Fig. 2.4.1
2.4.1. Layer buildup of the LCP before lamination
Fig. 2.4.2. Side view cross section of the multi
multi-chip
chip phased array module
LCP is slowly ramped up to the final bonding temperature of 282o C. It is kept at this
temperature for 30 minutes. Then, the LCP board is slowly cooled down to room
temperature. Concurrently, while the temperature is applied, gas pressure must also be
applied to ensure good contact between LCP layers.
32
In the millimeter
millimeter-wave
wave frequency regime, inductance from the bondwires can
severely degrade performance. Multiple parallel bondwires can be used to mitigate the
inductance seen by the chip. Therefore, compensation of the bondwire is necessary. This
is done throughh the use of capacitative pads that compensate the effects of the bondwire.
Figure 2.4.3 shows the simulation model for the dual bondwire transition with the
inductive
ctive compensation pads. Figure 2.4.4 shows the predicted simulated results which
show a VSWR of better than 2:1 in the Ka
Ka-band
band and a maximum insertion loss of 0.5 dB.
Fig. 2.4.3. Bondwire compensation model
5
0
0
-0.1
-5
-0.2
-0.3
-10
-0.4
-15
-0.5
-20
-0.6
-25
-0.7
-30
-0.8
-35
-0.9
-40
-1
2
10
18
26
34
Insertion Loss (dB)
Return Loss (dB)
33
42
Frequency (GHz)
Fig. 2.4.4. Predicted simulation results of bondwire compensation
2.5 Active Array System Design and Interface
Many times, the low frequency control signal distribution can be a much greater
challenge than the RF circuit portion. For example, a typical phase shifter will have 5 bits
of control. For an 8 element array, that is 40 control lines. For a 100 element array, that is
500 control lines. The routing and distribution can be a challenge itself. In many modern
phased array module architectures, a tile-based array is used. Essential to reducing the
complexity of the tile-based array is the integration of all control distribution networks
onto the subarray itself. For this, a multilayer board is paramount and the advantages of
using a cost effective multilayer organic substrate is clear. In this design, the top layer is
used for passive circuit fabrication and active circuit MMIC assemblies. The bottom layer
was used for surface mounted digital control chips used to interface with the 5-bit phase
34
shifter. An intermediate layer is used for control line routing purposes. Finally a ground
plane layer is also used for microstrip structur
structures
es on the top layer and to shield from the
noisy digital circuits. Since the phase shifters have 55-bits
bits of control, and there are a total
of 8 elements, routing control lines would be very difficult. Using a serial to parallel shift
register would simplify the bus controller interface significantly. The Hittite
HMC677LP5 is a 6 bit serial to parallel controller with a built in level shift to interface
with GaAs components. A 3 wire SPI interface is possible when multiple chips are
cascaded. An FPGA with cus
custom
tom verilog was used to interface with the controller chip to
control the phase setting. Figure 2.5.1 demonstrates the serial to parallel interface. The
verilog code used to control the FPGA is shown in the appendix.
A system level schematic of the 88-element
nt active phased array module is shown
in Figure 2.5.2. The completed wideband phased array receiver module along with chip
assembly
ly photographs are shown in Figure 2.5.3
2.5.3.
Fig. 2.5.1. Serial to Parallel Interface
35
The MMICs were attached in a chip
chip-on-board
rd assembly using Diemat DM5030P silversilver
loaded polymeric epoxy which is cured at 200o C for 30 minutes. In this system, the
Hittite ALH369 LNA chip with a nominal gain of 22 dB covering the bandwidth from
24-40
40 GHz was used. The phase shifter MMIC is Triqu
Triquint’s TGP-2102EPU
2102EPU with 5-bits
5
of
control, a nominal insertion loss of 7 dB, with 3.5o of RMS phase error at the center
frequency, covering the frequency span from 32
32-37
37 GHz. Finally, to simplify the control
routing, the Hittite HMC677LP5 6 bit serial to pa
parallel
rallel controller with a built in level
shift to interface with GaAs components was interfaced with the phase shifter MMIC. A
3 wire SPI interface is possible when multiple chips are cascaded. An FPGA with custom
Verilog was interfaced with the controller chip to control the phase setting.
Fig. 2.5.2
2.5.2. System level schematic of linear phased array
36
Fig. 2.5.3. Phased array module top side (top left), phased array module bottom side (top right), LNA
assembly (bottom left), phase shifter assembly (bottom right)
2.6 Measurement Results
For radiation pattern measurements, the phased array antenna receiver module
was tested using the setup shown in Figure 2.6.1, with a HP8510C Network Analyzer in
conjunction with a computer controlled azimuth rotators. Calibration was achieved using
a standard gain horn antenna. The dynamic range of the system was increased using a
power amplifier on the transmit side, as well as a low noise amplifier (LNA) on the
backend of the receive side. Finally, a computer interfacing with a FPGA controlled the
phase states for proper beam steering. A frequency step of 1 GHz was used with 1o steps
37
Fig. 2.6.1. Test Setup
in the azimuth angle. Figure 2.6.2 shows the measured radiation patterns at 34 GHz with
the beam steered from ±30o in 15o steps. From the measured data, it is observed that the
amplitude with each beam steering angle is dissimilar. This is due to the asymmetric
beam pattern of the tapered slot radiator [[18],
], and can be corrected with asymmetric flare
compensation in the fin-line
line of the antenna.
As can be seen, at 34 GHz the pattern is well defined and there is less than 2o of
beam shift from the intended beam angle. This is because at the center frequency the
phase relationship is proportional to the frequency, and the beam squint is minimal. 3-D
3
representations of the radiation pattern can be useful in observing the beam pattern over
frequency, shown in Figure 2.6.3. When the active array is at boresight, the radiation
38
pattern isfairly constant, and does not shift over frequency. However, when the beam is
steered towards 15o, there is a slight beam squint at 32 and 37 GHz, the edges of the
operating frequency of the array. When the beam is steered towards 30o, the beam squint
becomes a little more obvious, and it can be seen that at the lower end of the frequency
spectrum, at 32 GHz, the observed beam squint is less than 5o. At the high end of the
frequency spectrum, 37 GHz, the observed beam squint is less than 6.75o.
Also observed from measurements is the presence of high sidelobes when the
beam is steered to -15o and -30o regardless of frequency. This suggests a high phase error
in this particular phase state. Imperfections in the measured array factors are due to
numerous reasons: the limited dynamic range caused by the frequency limitations of test
instruments, mutual coupling between antenna elements, reflection of power due to
mismatches, non-linear phase error from the phase shifter, and finally phase and
amplitude imbalance from antenna array. Degradation in the pattern due to phase errors
can be remedied by proper channel calibration. The addition of an attenuator for each
channel would allow for this.
Finally, calibrated gain measurements of the active array were also taken across
frequency; this is shown in Figure 2.6.4. Calibrations were taken with a WR-28 standard
gain horn antenna. The measured gain variation at boresight is between +3 dBi from 32–
37 GHz. When the array is steered towards 15o the gain fluctuation is +2 dBi. Finally,
when the array is steered towards 30o, a maximum gain variation of 4 dBi was measured.
Measurements show that the gain variation across frequency with different beam steering
directions is moderately small. Gain variations can also be remedied by proper channel
calibration through the use of weighted attenuators and variable gain blocks.
39
15
10
Gain (dBi)
5
0
-5
-10
-15
-20
-90
-70
-50
-30 -10 10 30
Azimuth (Degree)
50
Fig. 2.6.2
2.6.2. Measured radiation pattern at 34 GHz
(a)
70
90
40
(b)
(c)
Fig. 2.6.3. Measured 33-D
D antenna pattern for scan angles at (a) boresight
bores
(b) 15o (c) 30o
41
14
12
Gain (dBi)
10
8
6
4
2
30
32
34
36
38
40
Frequency (GHz)
Boresight
+15 Degree
+30 Degree
Fig. 2.6.4. Calibrated Gain Measurements
2.7 Conclusions
An integrated wideband millimeter wave phased array receiver module built on a
multilayer light weight organic substrate is presented in this work. The 8 element phased
array with integrated digital controls and wideband antennas is targeted for the Ka-band
(32-37 GHz) and achieves a maximum measured scan angle of ±30o from boresight with
a maximum beam squint of 6.75o. The total size of the mm-wave module is 100 mm x
100 mm with a board thickness of 16 mils and a total weight of less than 12.3 grams. To
the author’s best knowledge, this work represents the most complex multilayer LCP
module reported to date, proving LCP to be an enabling technology for light weight, high
density, large scale phased array systems.
42
2.8 Acknowledgments
This chapter, in part, has resulted in the following publications:
[1: Submitted] J. S. Chieh, A. Pham, T. W. Dalrymple, D. G. Kuhl, B. B. Garber, K.
Aihara, " A Millimeter Wave Phased Array Multi-Chip Module on Liquid Crystal
Polymer," Components, Packaging and Manufacturing Technology, IEEE Transactions
on ,
[2] J. S. Chieh, A. Pham, T. W. Dalrymple, D. G. Kuhl, B. B. Garber, K. Aihara, "A
Light Weight 8-Element Broadband Phased Array Receiver on Liquid Crystal Polymer,"
in IEEE International Microwave Symposium(IMS), Anaheim, CA, May 2010
[3] J.S. Chieh, A.V. Pham, T. Dalrymple, K. Aihara, D. Kuhl, B. Garber, “A broadband
8-Element Phased Array Antenna Receiver on Liquid Crystal Polymer in the Ka-Band,”
in Proceedings of GOMAC, Reno, NV, March 2010
[4] J.S. Chieh and A.V. Pham, Development of a Wide Bandwidth Wilkinson Power
Divider on Multilayer Organic Substrates, Microwave Opt Technology Letters, July 2010,
Issue 52#7
[5] Chieh, J.-C.S.; Anh-Vu Pham; , "Development of a broadband Wilkinson power
combiner on Liquid Crystal Polymer," Microwave Conference, 2009. APMC 2009. Asia
Pacific , vol., no., pp.2068-2071, 7-10 Dec.2009
[6] H. Ta, M.J. Chen, K. Aihara, A. Chen, J.S. Chieh and A.V. Pham, “LCP for Passive
Components,” in “LCP for Microwave Packages and Modules”, Cambridge Press, June
25th 2012.
[7] M.J. Chen, K. Aihara, A. Chen, J.S. Chieh and A.V. Pham, “LCP for System Design,”
in “LCP for Microwave Packages and Modules”, Cambridge Press, June 25th 2012.
I would like to acknowledge and thank my co-author Prof. Anh-Vu Pham.
43
References
[1] Hajimiri, A.; Hashemi, H.; Natarajan, A.; Guan, X.; Komijani, A., "Integrated Phased
Array Systems in Silicon," Proceedings of the IEEE , vol.93, no.9, pp.1637-1655, Sept.
2005
[2] Schrank, H.; Hemmi, C., "Antenna designer's notebook-bandwidth of the array factor
for phase-steered arrays," Antennas and Propagation Magazine, IEEE , vol.35, no.1,
pp.72-74, Feb 1993
[3] Kwang-Jin Koh; May, J.W.; Rebeiz, G.M., "A Millimeter-Wave (40–45 GHz) 16Element Phased-Array Transmitter in 0.18-$mu$ m SiGe BiCMOS Technology," SolidState Circuits, IEEE Journal of , vol.44, no.5, pp.1498-1509, May 2009
[4] J.J.Xu, W. Yi-Feng, S.Heller, S.Heikman, U.K.Misra, R.A.York, “1-8-GHz GaN
based power amplifier using flip-chip bonding,” IEEE Microwaves and Guided Wave
Letters, vol. 9, no.7, pp. 277-279, Jul. 1999
[5] S. Shamsinejad, M. Soleimani, N. Komjani Novel Miniaturized Wilkinson Power
Divider for 3G Mobile Receivers,” Progress in Electromagnetics Research Letters,
vol. 3, pp. 9-16, 2008
[6] E.J. Wilkinson, “An N-way hybrid power divider,” IEEE Trans. Microw. Theory
Tech., vol. MTT-8, no. 1, pp.116-118, Jan. 1960
[7] S.B. Cohn, “A Class of Broadband Three-Port TEM-Mode Hybrids,” IEEE Trans.
MTT, Vol. MTT-16, pp. 110-116, Feb. 1968
[8] S. Horst, R. Bairavasubramanian, M.M. Tentzeris, J. Papapolymerou, “Modified
Wilkinson Power Dividers for Millimeter-Wave Integrated Circuits,” IEEE Trans.
Microw. Theory Tech., vol.55, no. 11, pp.2439-2446, Nov. 2007
[9] Xing-Ping Ou; Qing-Xin Chu, "A modified two-section UWB Wilkinson power
divider," Microwave and Millimeter Wave Technology, 2008. ICMMT 2008.
International Conference on , vol.3, no., pp.1258-1260, 21-24 April 2008
[10] S. Horst, R. Bairavasubramanian, M.M. Tentzeris, J. Papapolymerou, “Modified
Wilkinson Power Dividers for Millimeter-Wave Integrated Circuits,” IEEE Trans.
Microw. Theory Tech., vol.55, no. 11, pp.2439-2446, Nov. 2007
[11] C.Q. Li, S.H. Li, R.G. Bosisio, “CAD/CAE Design of an Improved, Wideband
Wilkinson Power Divider,” Microwave Journal, pp.125-130, Nov. 1984
44
[12] Yi Sun; Freundorfer, A.P., "Broadband folded Wilkinson power
combiner/splitter," Microwave and Wireless Components Letters, IEEE , vol.14, no.6, pp.
295-297, June 2004
[13]
Ansoft Corporation, Pittsburgh, PA
[14] Lee, K. F. and W. Chen, Advances in Microstrip and Printed Antennas, John
Wiley and Sons, 1997
[15] Langley, J.D.S.; Hall, P.S.; Newham, P., "Balanced antipodal Vivaldi antenna for
wide bandwidth phased arrays ," Microwaves, Antennas and Propagation, IEE
Proceedings - , vol.143, no.2, pp.97-102, Apr 1996
[16] D. Schaubert, E. Kolberg, T. Korzeniowski, T. Thungren, J. Johansson, K.
Yngvesson, “Endfire tapered slot antennas on dielectric substrates,” IEEE Trans.
Antennas and Propagation, vol. 33, no. 12, pp. 1392-1400, Dec. 1985
[17] Lu Yang; Domier, C.W.; Luhmann, N.C., "Ka-Band True Time Delay E-plane
Beam Scanning and Broadening Phased Array System using Antipodal EllipticallyTapered Slot Antennas," Antennas and Propagation Society International Symposium
2006, IEEE , vol., no., pp.2213-2216, 9-14 July 2006
[18] Langley, J.D.S.; Hall, P.S.; Newham, P., "Balanced antipodal Vivaldi antenna for
wide bandwidth phased arrays ," Microwaves, Antennas and Propagation, IEE
Proceedings - , vol.143, no.2, pp.97-102, Apr 1996
45
Chapter 3
A Bidirectional X-Band Antenna Array on LCP
Typical planar radiators such as the patch antenna, inverted-F antenna, dipole
antenna and others have a size that is proportional to the frequency of operation. All of
the aforementioned antennas are medium in size and have dimensions that vary from λ/4
– λ/2. Although it is difficult to generalize all scenarios, typical bandwidths of these
antennas are usually no greater than 10%. To be able to obtain larger bandwidths, this
usually entails enlarging the antenna. Table 3.1 shows a comparison between some
typical planar antennas from medium in size and narrow in bandwidth, to large in size
and broadband. Figure 3.1.1 illustrates some of these antennas.
Typical antennas have a size that is relative to the frequency, and antennas in the
lower GHz regime can be quite large. As mentioned previously, high density integration
is ever increasing, and the demand for smaller and smaller electronic components is
growing. Many times, the antenna is the bottleneck and can be the largest component.
46
Table 3.1.
Antenna Comparison
Fig. 3.1.1. Typical Planar Antennas
The promise of miniaturized antennas has been the Holy Grail for antenna
engineers. Realization of these small antennas is challenging and a field in it of itself.
Small antennas, however, are not without their limitations. Small antenna theory was
developed by Wheeler and Chu [1], and without diving into the theory, it can be said that
small antennas generally have very small gains, high quality factors, and small
bandwidths.
47
One method for antenna miniaturization that has recently gained a lot of attention
has been metamaterials. Metamaterials is generally defined as a material that does not
occur homogeneously in nature and has to be synthetically or artificially created.
Metamaterials is a broad category which include numerous sub-categories ranging from
photonic bandgap structures (PBG), electro-magnetic bandgap structure (EBG), slowwave structures, periodic structures, and composite right-left hand transmission lines
(CRLH).
Beamforming arrays with the ability to scan N-number of beams concurrently
typically require N-number of concurrent receivers or transceivers when using traditional
architectures such as the Rotmans lens [2], butler matrix [3], and phased array
beamformer [4]. These architectures allow for multiple concurrent beam scanning in the
forward direction orthogonal to the antenna plane. Through the use of a bidirectional
antenna element, we can extend the field of view to also include concurrent scanning in
the backward direction orthogonal to the antenna plane. This would be useful for
applications where differentiation in the polarity of the beams is immaterial; examples
would include microwave sensor networks and collision avoidance systems.
In this chapter we present a novel planar wideband polygon slot antenna using a
metamaterial coupling element. The antenna achieves an impedance bandwidth of 17.5%
at a center frequency of 11.4 GHz. The proposed antenna shows bidirectional radiation
patterns in both the E and H planes. The total size of the antenna is 2.5 x 2.0 cm2.
Utilizing this novel antenna element, we also present the design of a 16-element antenna
array which has near symmetric beamwidths in both the E- and H-planes. The
bidirectional antenna array achieves a gain of 14.9 dBi with a radiation efficiency of 59%
48
and an impedance bandwidth of 21%. The antennas are built on a LCP substrate, which is
lightweight, flame retardant, and near hermetic which makes it an ideal material for
microwave modules.
3.2 Overview of Meta-material Antennas
Many metamaterial antennas have been reported in literature, and the majority of
them utilize the CRLH structure. The CRLH structure is the most applicable to planar
antennas as the implementation lends itself to printed circuit board technology. Within
the CRLH antenna category, there are two major sub-categories. The first is a
metamaterial leaky wave (MTM LW) antenna where the radiation is based on a leaky
wave mechanism which is based on traveling-waves [5]. These types of antennas tend to
exhibit high directivity with a fan-beam pattern simply because the radiating aperture is
electrically long in the horizontal direction. For traditional antennas, such as the patch,
the size is proportional to the operating frequency so one cannot simply make the antenna
larger in order to increase the gain or directivity. Using traditional antennas, in order to
increase the effective aperture, an array may be used; however the spacing of the array is
critical and can often exceed λ/2, which makes the radiating aperture quite large. This is
main benefit of the MTM LW antenna, in order to increase the gain or directivity, one
can simply cascade more CRLH cells and the cells do not need to be spaced in multiples
of λ/2. However, since the MTM LW antennas tend to be quite large, that is not the focus
of this work. The second type of MTM antenna is the Zero-Order Resonance (ZOR)
antenna where the radiation is based on a resonance mechanism. The structure is similar
to that of the MTM LW antenna except that the antenna is terminated with either an open
49
or a shorted load. This creates the resonance mechanism whereby the radiating wave is a
standing wave that has broadside characteristics [6]. Figure 3.2.1 shows the equivalent
circuit of a CRLH transmission line. From the equivalent circuit, it can be derived that
the propagation constant is:
ϖ
γ = ZY = α + jβ = js(ϖ ) 
ϖ R
2
2
 ϖ L 
2
 + 
 − Kϖ L
ϖ

 
− 1 if ϖ < min(ϖ se ,ϖ sh ) LH
where s (ϖ ) = 
+ 1 if ϖ > max(ϖ se ,ϖ sh ) RH
(1)
range
range
The series resonance can be defined as:
ϖ se =
1
(2)
LR C L
The shunt resonance can be defined as:
ϖ sh =
1
LL C R
(3)
-LR: right-handed series inductance
-CL: left-handed shunt capacitance
-CR: right-handed shunt capacitance
-LL: left-handed series inductance
(all per unit length)
Fig. 3.2.1. Equivalent CRLH transmission line model
50
The dispersion diagram is shown in Figure 3.2.2 and as can be seen, in the left handed
regime, the absolute value of the propagation constant ϐ is much larger than in the
traditional right handed regime. This results leads to the observation that for a physical
size, the wavelength is larger in the left handed regime than in the right handed regime.
Typical transmission line resonators operate in when the electrical length is θ = βl
or a multiple thereof, θm = βml = mπ. However, it is observed from the dispersion
diagram that the CRLH transmission line can have negative resonances and is rather nonlinear. Especially peculiar is the mode when m=0 or when ϐ=0. Since the field
distribution is flat when ϐ=0, this suggests that the resonance is decoupled from the
physical length, and therefore arbitrarily small resonators can be created.
Figure 3.2.2. Dispersion diagram of CRLH transmission line
51
When the CRLH transmission line is terminated with an open the resonating frequency is
defined as:
open
= ϖ sh =
ϖ res
1
LL CR
(4)
When the CRLH transmission line is terminated in a short, the resonating frequency is
defined as:
short
ϖ res
= ϖ se =
1
LR C L
(5)
As mentioned before, in the zeroth-order mode, the size of the resonator is not dependent
on the physical size but only by the values of the distributed lumped elements. This
creates a unique opportunity for miniaturization of antennas. However, there are two
caveats associated with building small resonating antennas using the MTM CRLH
concept. The first is that because the resonators are miniaturized, the effective radiating
surface area is also greatly reduced, therefore the directivity of the antennas tend to be
very small. The directivity of the antenna is proportional to the physical size, and
therefore although electrically small antennas can be designed, they have all the
performance metrics of a small antenna, namely low directivity [7]. The second concern
is that the design equations from (1) – (5) assume that the CRLH transmission line is
effectively homogenous. This can only occur when an infinite number of unit cells are
used, which is not conducive to the miniaturization of antennas. Typical MTM zeroth-
52
order antennas reported in literature [8] – [13] use 1-5 unit cells, and therefore the MTM
transmission line can no longer be considered homogenous and the governing equations
from (1) – (5) no longer hold true. However, it does provide a good starting point for
predicting the resonant frequencies.
Of the different zeroth-order MTM antennas reported in literature, one type has
received much attention as it overcomes one of the major issues with MTM antennas,
which is lack of bandwidth. It was discovered in [14] that with a truncated ground plane,
the resonances such as m = -1 and m = -2 could be turned such that they were very close
to the resonance m = 0, and therefore the bandwidth could be extended. Since the ground
plane is truncated, the transmission line can no longer be considered homogenous [15],
however, as stated before, provides a good starting point for the design of electrically
small antennas. It is also noted that it could be argued that the miniaturization of antennas
through the use of distributed lumped elements to create the CRLH MTM resonator
antennas is simply the result of reactive loading, similar to that of the PIFA (planar
inverted F antenna). This approach would not be incorrect.
53
3.3 Design of Metamaterial Inspired Wideband Slot Antenna
The planar wide slot antenna is one such antenna that shows bidirectional
radiation characteristics [16]. These antennas have an inherent advantage over other
planar antenna types in that the wide slot antenna is less dependent on the size of the
ground plane, and therefore can achieve a smaller footprint when compared to similar
planar antennas such as a patch antenna. Several different geometries of wide slot
antennas have been reported including elliptical [17], square [18], and circular [19] slots
with various types of coupling structures between the feedline and the slot.
The proposed antenna is printed on a single layer 10 mil thick LCP substrate with
a dielectric constant, εr ≈ 2.94 and a loss tangent δ ≈ 0.002. Figure 3.3.1a shows the
bottom metal layer of the proposed antenna while Figure 3.3.1b shows the top metal
layer. The respective dimensions are presented in Table 3.1. As can be seen, a polygon
slot was implemented in this design, with the widest portion of the slot being ~ λ/2.
Optimization of the slot was achieved parametrically in CST Microwave Studio
simulation software. For a wide slot antenna, coupling between the feedline and the slot
is essential and can affect both the impedance bandwidth and radiation pattern. Typical
coupling mechanisms include using a monopole tuning stub. Recently, researchers have
shown that small folded monopole antennas can be realized through the use of reactive
loading inspired by metamaterials [20]. Typical monopole radiators require a quarter
wavelength transmission line to operate, however, the same effect can be achieved
through metamaterial inspired reactive loading. Figure 3.3.2 shows the equivalent lumped
element model of the designed reactive tuning stub. At the designed resonance frequency,
this reactive circuit allows RF current to flow, thereby emulating a monopole antenna. As
54
(a)
(b)
Fig. 3.3.1. Geometry of the proposed antenna (a) bottom conductor (b) top conductor
Table 3.2.
Proposed Antenna Dimensions
Dimension
(mm)
Dimension
(mm)
W1
20
L1
25
W2
0.42
L2
3.53
W3
0.2
L3
4.4
W4
0.6
L4
10.5
W5
0.2
L5
4.2
W6
16
L6
3
Gap1
0.12
55
Fig. 3.3.2. Equivalent lumped element model of coupling element
can be seen, the first series inductor L1 is realized from the inductive strip at the end of
the microstrip feed, indicated by the width W3. The capacitor C2 is formed from the gap
capacitance that is seen between the V-shaped structure and the square patch. The
inductor L2 is formed from a combination of the inductance from the through via and the
inductive strip on the bottom conductor which is indicated by the width W5. Finally the
capacitance C1 is realized by the fringing fields from the square patch on the top metal
and the slotted ground plane on the bottom side. This provides an intuitive understanding
of how the reactive coupling element behaves, at the same time; it is acknowledged that
at microwave frequencies many parasitic effects can also play a significant role.
Since both the wide slot and coupling mechanism is fairly decoupled from the
substrate, this antenna can be realized on a variety of substrate materials with an
additional freedom to choose the substrate thickness, without really sacrificing the
performance. Fig. 3.3.3 shows the simulated co- and cross-polarization patterns at 11.5
GHz for a 5, 10, 15, 20 mils LCP substrate thickness. The reactive coupling element and
the wide slot dimensions are kept the same while the microstrip feedline was slightly
modified so that the antenna resonates at 11.5 GHz at each case. The calculated gains
56
using Ansoft-HFSS are 5.0 dBi, 4.78 dBi, 4.88 dBi, and 5.15 dBi, with an associated
maximum cross-polarization level of -7 dB, -7.1 dB, -6.7 dB, and -5.36 dB, respectively
for substrate thicknesses of 5, 10, 15, and 20 mils. It can be seen that the radiation pattern
is quite independent of the substrate height.
(a)
(b)
57
(c)
(d)
Fig. 3.3.3. Simulated Co- and Cross-polarization radiation patterns at 11.5 GHz on (a) 5 mil (b) 10 mil (c)
15 mil (d) 20 mil LCP substrates
58
The wide slot antenna is fairly insensitive to the ground plane size in regards to
frequency detuning as the elements contributing to the resonance are the slotted ground
plane and the coupling element. Three ground plane sizes were chosen arbitrarily from
small to large to investigate the effects; they are shown in Fig. 3.3.4a. The input return
losses of the three simulated antennas are shown in Fig. 3.3.4b., and it is apparent that
with an increase in ground plane size, the bandwidth of the antenna decreases slightly and
the frequency of operation is shifted down 1 GHz. The main reason for this behavior is
because of the lower ground inductance associated with a larger ground plane.
The radiation patterns are also simulated and analyzed shown in Fig. 3.3.5. The
simulated peak gains for the antennas are 4.78 dBi, 4.94 dBi, and 5.05 dBi for the small,
medium, and large ground planes respectively. It is noted that the cross polarization in
both planes are quite high and is largely due to the strong horizontal components of the
surface current which can lead to increased cross polarization levels [18]. It is seen that as
the ground plane size increases, the cross-polarization levels also increase. In the H-plane
we see approximately a 4 dB at increase at 45o in cross polarization levels from the
antenna with the small ground plane to the large ground plane. In the E-plane we observe
an 8.7 dB increase at -90o in cross polarization levels. As the size of the ground plane
increases, the surface area for horizontal components of the surface current also increase,
therefore degrading cross-polarization performance. It can be seen that a ground plane
size of 25 mm x 20 mm is quite ideal.
59
(a)
(b)
Fig. 3.3.4.. (a) Parametric study on ground plane size (b) simulated S11 for respective ground plane sizes
Fig. 3.3.5. Co- and Cross-polarization patterns for various ground plane sizes
60
The fabricated prototype antenna used for measurements is shown in Figure 3.3.6.
Simulations were completed in CST Microwave Studio. An 8mil through via is used. The
return loss of the antenna was measured by using an E8363B vector network analyzer. A
single port SOLT calibration was used. The measured and simulated input return loss
curves are plotted in Figure 3.3.7. As can be seen, there is good agreement between
simulation and measurement results. The measured antenna shows a 10 dB bandwidth of
2 GHz or equivalently a fractional bandwidth of approximately 17.5%.
The radiation patterns of the prototype antenna were measured inside an anechoic
chamber. As shown in Figure 3.3.8, the normalized measured patterns at 11.5 GHz agree
well with simulation. The E-plane (y-z plane) pattern is clearly bidirectional, showing
near equivalent beamwidth and magnitude in the forward and backward directions. The
H-plane (x-z plane) pattern is more omnidirectional, but the maxima still lies in the
forward and backward directions. It is noted that the cross polarization in both planes are
quite high and is largely due to the strong horizontal components of the surface current
which
Fig. 3.3.6. The fabricated prototype antenna (a) Front side (b) back side
61
Fig. 3.3.7. Measured and simulated return loss curves for proposed antenna
Fig. 3.3.8. Measured and simulated normalized radiation pattern in the E and H planes
62
Fig. 3.3.9. Simulated surface current distribution at 11.5 GHz
can lead to increased cross polarization levels [18]. The surface current distribution is
shown in Figure 3.3.9.
3.4 Design of Bidirectional X-Band Antenna Array
The mutual coupling between two antennas, in both E-plane and H-Plane
orientations was measured using the antenna layout shown in Fig. 3.4.1. A two port
SOLT calibration was used. The antennas were fabricated with a center to center spacing
of approximately 0.8λo at 11.5 GHz. The measurements shows a mutual coupling < -25
dB in the H-Plane orientation from 7 – 15 GHz. In the E-Plane orientation, the measured
mutual coupling is < -20 dB from 5 – 15 GHz. [Fig. 3.4.1(a) and (b)].
63
(a)
(b)
Fig. 3.4.1. Measured and simulated mutual coupling in the (a) H-Plane (b) E-Plane orientation. Inset shows
fabricated test structures for each orientation
64
Building on the bidirectional novel antenna structure proposed in the previous
section, a 16 element antenna array was modeled and simulated in CST Microwave
Studio. A T-junction power combiner is used for the corporate feed network. The model
includes an edge connected coaxial adapter from Southwest Microwave. Figure 3.4.2
shows the 3-dimensional model of the antenna array. An inter-elemental spacing of 0.8λ
was used.
Fig. 3.4.2. 3D model of 16-element concurrent dual beam antenna array
The simulated E and H plane radiation patterns at 11 GHz are shown in Figure 3.4.3. The
simulated E plane beamwidth is 17.3o while the H plane beamwidth is 16.9o. In both the
E and H planes, it is clear that radiation patterns are bidirectional and symmetric in both
magnitude and beamwidth. The simulated gain is approximately 15 dBi at 11 GHz with
65
an efficiency of 67%. It is noted that the sidelobe levels are asymmetric and is attributed
to the asymmetry contributed by the feedline, which can also contribute to radiation.
Since this antenna array is bidirectional, with the introduction of a progressive phase shift
between antenna elements, it becomes clear that the array’s forward and backward beams
can be steered simultaneously; although not independently. This would be useful for
applications where differentiation in the polarity of the beams is immaterial; examples
would include microwave sensor networks and collision avoidance systems.
(a)
(b)
Fig. 3.4.3. Simulated radiation patterns in the (a) E-Plane and (b) H-Plane
66
3.5 Measurement Results
A 4x4 two dimensional antenna array was fabricated on a 10 mil thick LCP
substrate. A coaxial adapter from Southwest Microwaves was used to interface with the
microstrip antenna array launch. The fabricated antenna array is shown in Figure 3.5.1.
The return loss was measured with an Agilent E8361C performance network analyzer
using the Agilent 85056A 2.4mm standard coaxial calibration kit, using a 1 port SOLT
calibration. The measured and simulated return loss graph is shown in Figure 3.5.2 and
the measurement results correspond well with the simulation results. The measured 10dB
return loss bandwidth is approximately 2.8 GHz, which corresponds to a fractional
bandwidth of 25% at a center frequency of 11.1 GHz.
Fig. 3.5.1. Fabricated 4x4 concurrent dual beam antenna array
67
Fig. 3.5.2
3.5.2. Measured and simulated return loss for 4x4 antenna array
It was found that simulating the antenna array with the coaxial adapter made for better
correlation between measured and simulated results.
Radiation
ion pattern measurements were made inside the anechoic chamber at UC
Davis. An Agilent 8510C network analyzer is used along with standard gain horn
antennas to obtain gain measurements. Custom labview software is used in order to rotate
the platform in the anechoic chamber in both the elevation and azimuthal planes and also
capture the data from the network analyzer. Figure 3.5.3 shows the measured radiation
pattern of the 4x4 antenna array in the elevation plane at 10 GHz and 12 GHz
respectively. The measured
red and simulated radiation patterns match extremely well. Major
sidelobes are more than 10dB down from the main lobe. The front
front-back
back ratio is 1:1. As
expected, the radiation pattern is bidirectional and symmetric. It is noted that the sidelobe
68
(a)
(b)
Fig. 3.5.3. Measured radiation pattern in the elevation plane at (a) 10 GHz (b) 12 GHz
levels are asymmetric and is attributed to the asymmetry contributed by the feedline,
feed
which can also contribute to radiation.
As expected, the radiation pattern is bidirectional and symmetric. As the
frequency increases, the beamwidth of the array decreases and this is expected as the
physical aperture scales with frequency. This beh
behavior
avior is better illustrated using nn
dimensional plots. Figure 3.
3.5.4 shows a 2D plot of the normalized magnitude of the
beams in the elevation pl
plane across frequency. Figure 3.5.5 shows the 3D plot of the
normalized magnitude of the beams in the elevation pplane
lane across frequency and angle. It
is significant to note that there are two major beams and they are symmetric in magnitude
and beamwidth.
width. As mentioned previously, as the frequency increases, the beamwidth of
both beams decrease. It is apparent that with the introduction of a progressive phase shift
between antenna elements, the array’s forward and backward beams can be steered
69
Fig. 3.5.4. 2D Plot of the radiation pattern in the elevation plane
Fig. 3.5.5. 3D Plot of the radiation pattern in the elevation plane
70
(a)
(b)
Fig. 3.5.6. Measured radiation pattern in the azimuth plane at (a) 10 GHz (b) 12 GHz
simultaneously; although not independently. This may not be practical
pract
for many
applications; however, for applications where the polarity of the beam is immaterial, this
type of antenna array can provide additional coverage in the backwards direction.
Interestingly, the front-to
to-back ration is 1:1.
Similarly, radiation pattern measurements were also taken in the azimuth plane.
Since the antenna array has equal number of elements in the horizontal and vertical
directions, it is expected that the beamwidth in the azimuth plane should be nearly equal
to the beamwidth in the elevation plane. Figure 3.5.6 shows the measured and simulated
radiation patterns in the azimuth plane at 10 GHz and 12 GHz respectively. The sidelobes
are down by at least 13 dB from the main lobe in both instances.
Figure 3.5.7 shows the 2D normalized radiation pattern and Figure 3.5.8
3.
shows
the 3D normalized radiation pattern in the azimuth plane. It can be seen that also in the
71
Fig. 3.5.7. 2D Plot of the radiation pattern in the azimuth plane
Fig. 3.5.8. 3D Plot of the radiation pattern in the azimuth plane
72
azimuth plane, as the frequency increases, the beamwidth decreases. It is clear from both
figures that both the front and back beams are symmetric in beamwidth and magnitude.
The front-to-back ratio in the azimuth plane is also 1:1.
Finally Figure 3.5.9 shows the calibrated gain measurements across frequency. In
simulation, the gain at 11 GHz is 15 dBi showing an efficiency of 67%. Measurements
show a gain of 14.9 dBi at 11 GHz indicating an efficiency of 65%, a 2% drop in
efficiency. This can be easily explained as in simulation, surface roughness was not
accounted for. In reality, in the microwave regime, surface roughness can contribute
heavily to the total loss. Additionally, surface waves in antenna arrays can also contribute
to losses. The loss due to these surface waves are often not available to observe in EM
(electro-magnetic) field solvers and therefore unaccounted for. Despite that fact, a 65%
efficiency for a 16-element antenna array, including a corporate feeding network is quite
Gain (dBi)
excellent.
15.75
15.50
15.25
15.00
14.75
14.50
14.25
14.00
13.75
13.50
10
10.5
11
11.5
Frequency (GHz)
Measurement
Simulation
Fig. 3.5.9. Calibrated gain measurement
12
73
3.6 Conclusions
A novel polygon wide slot antenna with a metamaterial inspired coupling
structure has been proposed and successfully implemented. The proposed antenna
occupies and area of 2.5 x 2.0 cm2 and achieves a fractional bandwidth of 17.5%. The
antenna has bidirectional radiation characteristics. A 16-element antenna array utilizing
the novel antenna is designed and characterized through simulation. The antenna array
has concurrent dual beams in the forward and backward directions. A 4x4 antenna array
is fabricated and characterized in an anechoic chamber. Both radiation patterns in the
elevation and azimuth planes show bidirectional characteristics. A beamwidth of 17o is
measured for both the elevation and azimuth plane at 10 GHz. A maximum gain of 14.9
dBi is measured at 10 GHz showing an efficiency of greater than 65%. The 4x4 antenna
array shows a measured fractional bandwidth of 25%.
It is apparent that with the introduction of a progressive phase shift between
antenna elements, the array’s forward and backward beams can be steered
simultaneously; although not independently. This may not be practical for many
applications; however, for applications where the polarity of the beam is immaterial, this
type of antenna array can provide additional coverage in the backwards direction.
74
3.7 Acknowledgments
This chapter, in part, has resulted in the following publications:
[1: To Submit] Chieh, J.S.; Pham, A.V.;, "A Bidirectional Microstrip X-Band Antenna
Array on Liquid Crystal Polymer for Beamforming Applications," Antennas and
Propagation, IEEE Transactions on
[2] Chieh, J.S.; Pham, A.V.; , "A Bidirectional X-Band Antenna Array on Liquid Crystal
Polymer," Antennas and Propagation Society International Symposium (APSURSI), 2011
IEEE , vol., no., pp.1-4, 3-8 July 2011
I would like to acknowledge and thank my co-author Prof. Anh-Vu Pham.
References
[1] L. J. Chu, "Physical Limitations of Omni-Directional Antennas,"J. Appl. Phys., 19,
December 1948, pp. 1163-1175.
[2] Rotman, W.; Turner, R.; , "Wide-angle microwave lens for line source applications,"
Antennas and Propagation, IEEE Transactions on , vol.11, no.6, pp. 623- 632, Nov 1963
[3] J. Butler and R. Lowe "Beam forming matrix simplifies design of electronically
scanned antennas", Electron. Design, pp.170 1961 .
[4] Dong-Woo Kang; Kwang-Jin Koh; Rebeiz, G.M.; , "A Ku -Band Two-Antenna FourSimultaneous Beams SiGe BiCMOS Phased Array Receiver," Microwave Theory and
Techniques, IEEE Transactions on , vol.58, no.4, pp.771-780, April 2010
[5] C. Caloz and T. Itoh, Electromagnetic Metamaterials: Transmission Line Theory and
Microwave Applications. New York: Wiley, 2004.
[6] C. Lee, K. M. H. Leong, and T. Itoh, "Composite Right/LeftHanded Transmission
Line Based Compact Resonant Antennas for RF Module Integration," IEEE Transactions
on Antennas andPropagation, AP-54, 8, August 2006, pp. 2283-229
[7] J. Volakis, C. Chen, K.Fujimoto, Small Antennas: Miniaturization Techniques &
Applications, New York: McGraw Hill, 2010
[8] Jae-Gon Lee; Jeong-Hae Lee; , "Zeroth Order Resonance Loop Antenna," Antennas
and Propagation, IEEE Transactions on , vol.55, no.3, pp.994-997, March 2007
75
[9] Lai, A.; Leong, K.M.K.H.; Itoh, T.; , "Infinite Wavelength Resonant Antennas With
Monopolar Radiation Pattern Based on Periodic Structures," Antennas and Propagation,
IEEE Transactions on , vol.55, no.3, pp.868-876, March 2007
[10] Jiang Zhu; Eleftheriades, G.V.; , "A Compact Transmission-Line Metamaterial
Antenna With Extended Bandwidth," Antennas and Wireless Propagation Letters, IEEE ,
vol.8, no., pp.295-298, 2009
[11] Jiang Zhu; Antoniades, M.A.; Eleftheriades, G.V.; , "A Compact Tri-Band
Monopole Antenna With Single-Cell Metamaterial Loading," Antennas and Propagation,
IEEE Transactions on , vol.58, no.4, pp.1031-1038, April 2010
[12] Jaehyurk Choi; Sungjoon Lim; , "Circular polarized small antenna based
metamaterial coplanar-waveguide (CPW) transmission line," Microwave Conference
Proceedings (APMC), 2010 Asia-Pacific , vol., no., pp.2044-2047, 7-10 Dec. 2010
[13] Taehee Jang; Jaehyurk Choi; Sungjoon Lim; , "Compact Coplanar Waveguide
(CPW)-Fed Zeroth-Order Resonant Antennas With Extended Bandwidth and High
Efficiency on Vialess Single Layer," Antennas and Propagation, IEEE Transactions on ,
vol.59, no.2, pp.363-372, Feb. 2011
[14] C. J. Lee, K. M. H. Leong, and T. Itoh, "Broadband Small Antenna for Portable
Wireless Application," International Workshop on Antenna Technology: Small Antennas
and Novel Metamaterials, iWAT 2008, March 4-6, 2008, pp. 1O- l 3.
[15] Cheng-Jung Lee; Wei Huang; Gummalla, A.; Achour, M.; , "Small Antennas Based
on CRLH Structures: Concept, Design, and Applications," Antennas and Propagation
Magazine, IEEE , vol.53, no.2, pp.10-25, April 2011
[16] E. A. Soliman, S. Brebels, E. Beyne, P. Delmotte, and G. A. E. Vandenbosch,
"Brick-wall antenna in multilayer thin-film technology", Microw. Opt. Technol. Lett.,
vol. 19, pp.360 -365 1998
[17] Pengcheng Li; Jianxin Liang; Xiaodong Chen; , "Study of printed elliptical/circular
slot antennas for ultrawideband applications," Antennas and Propagation, IEEE
Transactions on , vol.54, no.6, pp. 1670- 1675, June 2006
[18] Horng-Dean Chen; , "Broadband CPW-fed square slot antennas with a widened
tuning stub," Antennas and Propagation, IEEE Transactions on , vol.51, no.8, pp. 19821986, Aug. 2003
[19] Soliman, E.A.; Ibrahim, M.S.; Abdelmageed, A.K.; , "Dual-polarized
omnidirectional planar slot antenna for WLAN applications," Antennas and Propagation,
IEEE Transactions on , vol.53, no.9, pp. 3093- 3097, Sept. 2005
76
[20] Jiang Zhu; Antoniades, M.A.; Eleftheriades, G.V.; , "A Compact Tri-Band
Monopole Antenna With Single-Cell Metamaterial Loading," Antennas and Propagation,
IEEE Transactions on , vol.58, no.4, pp.1031-1038, April 2010
[21] Wei Huang; Nan Xu; Pathak, V.; Poilasne, G.; Achour, M.; , "Composite Right-Left
Handed Metamaterial ultra-wideband antenna," Antenna Technology, 2009. iWAT 2009.
IEEE International Workshop on , vol., no., pp.1-4, 2-4 March 2009
77
Chapter 4
W-Band antennas for Phased Array applications on
LCP
As mentioned previously, the demands for bandwidth is ever increasing, however
the allotted spaces is very limited and regulated by the FCC (Federal Communication
Commission). For wireless applications that require the transfer of information over
medium to long ranges, the low GHz regime is preferable because of the low oxygen
attenuation. This is the main reason why cellular communications takes place between
900 MHz and 1.8 GHz. Figure 4.1.1 shows the effects of oxygen absorption in various
frequency bands. As this space has been designated by the FCC for cellular use,
frequency regimes elsewhere must be appropriated for large bandwidth communications.
For this reason, many have proposed high bandwidth wireless backhauls to operate in the
E-band and W-band [1], where the allocated bandwidth is much higher and thus support
high data-rates. The main benefit of moving these high bandwidth systems to the W-band
is that there exists a window around 94 GHz, where the oxygen absorption is a minimum,
78
Fig. 4.1.1. Oxygen absorption at various frequencies
allowing for potential long distance wireless communications. Also beneficial is that by
moving to the higher millimeter wave frequency regime, components such as antennas
can be largely miniaturized simply by scaling with the frequency. So, for the same
physical aperture size, the performance of an antenna in the higher millimeter wave
regime as compared to the lower GHz frequency regime is improved drastically.
Therefore, there exists a push towards the design of wireless communication systems,
especially phased arrays, in the W-band. Figure 4.1.2 illustrates the concept of using
millimeter wave high bandwidth wireless backhauls or Fiber Extensions.
At the same time as wireless technology improves, there exists a demand for low
cost highly integrated systems. Planar antennas that can be implemented on standard PCB
79
Fig. 4.1.2. Potential Millimeter Wave High Bandwidth Backhaul
manufacturing processes then become increasingly important. Waveguides are the
transmission medium of choice at high frequencies having qualities such as low loss and
high power handling. However, in the W-band, waveguides are not compatible with the
cost and high density integration needs of modern wireless systems. Moreover, as the
frequency of operation increases and the wavelength shrinks, the accuracy of
manufacturing becomes highly important. Antenna arrays at 94 GHz have been
demonstrated through the use of micromachining [2], but it is costly and difficult to
integrate with other microwave components.
80
4.2 Materials Considerations
When designing components in the millimeter-wave regime, and more
specifically in the W-Band the choice of substrate materials becomes increasingly
important. One of the most important parameters when choosing an appropriate substrate
is the loss associated with the material. PTFE materials are typically used in the
microwave regime, however, in the millimeter wave regime, the losses associated are too
large to be considered a practical material.
In the millimeter wave regime, three materials stand out and are most commonly
used to build components. The first is Fused-Silica Quartz. Fused-Silica has a low
permitivity of around εr = 3.8 and a loss tangent of around δ = 0.0025 [3]. Quartz wafers
come in a variety of thicknesses anywhere from 5 – 20 mils. Most millimeter wave
designs benefit from a thin substrate to reducing multi-moding issues. Microfabrication
techniques can be used to process the Quartz wafers, however, the main drawback of this
material is that for thicknesses less than 10 mils, the material is extremely brittle. This not
only limits the size of the components that can be built, but also the density of vias that
can be manufactured, as cracking becomes more prevalent.
Another material candidate is ceramics. There is a plethora of ceramics comprised
of Low-Temperature-Co-Fired (LTCC) and High-Temperature-Co-Fired (HTCC)
varieties. Ceramic materials come in a wide range of permitivities but generally range
anywhere from εr = 7 – 10 with loss tangents around δ = 0.0002 – 0.002. Because the
permitivity is high with the ceramic material, component sizes can be miniaturized. One
advantage of ceramics when compared to Quartz is that multi-layered designs can be
achieved with Ceramics. The two main drawbacks of ceramics are its high permitivity,
81
which leads to high Q structures. When considering components such as antennas, this
leads to narrow bandwidths and low radiating efficiencies. Another drawback is that
ceramics are dense and therefore components can be heavier. When considering large
antenna arrays, the added weight can be very detrimental.
A third and final material to be considered as a candidate for millimeter wave
substrates is LCP. LCP has a low permitivity of around εr = 2.94 and a loss tangent of
around δ = 0.002 which is stable up to 110 GHz [4]. Also, because of its polymer nature,
it is naturally flexible and mechnically sturdy, even when thickness are in the range of 2
mils. This means that components are not limited in size due to fabrication. In addition,
because of it’s low permitivity, it’s is an ideal material to build wideband high efficiency
antennas on. Finally, it has the ability to support multi-layered designs. Essentially it has
all the main benefits of both Ceramic and Quartz without any of the drawbacks.
4.3 T-Junction Design
As already stated in previous chapters, power combiners are one of the main
components in antenna arrays. The ability to combine signals coherently from each
antenna element is what makes antenna arrays so powerful. In many cases, the Wilkinson
combiner is preferable because of its high isolation. However, the isolation comes at a
cost of additional resistors. In the W-band, surface mount resistors simply add too many
parasitic to be practical. Although NiCr foil resistors are an option, this can add
complexity. Simply because of this reason, the antenna arrays in the subsequent sections
use T-Junction power combiners. Although the isolation between the output channels
82
tends to be low, for passive antenna arrays, this is typically not a problem. The TT
Junction used is a 3-port
port passive structure. Figure 4.3.1 showss the schematic of the TT
Junction that was designed and used for the W
W-band
band antenna arrays. Ports 1, 2, and 3 are
simply 50 Ω transmission lines. At the junction where ports 2 and 3 meet, a quarter wave
impedance transformer is used to match port 1. The designed T-Junction
Junction resides on a
single 4 mil LCP dielectric layer.
Fig. 4.3.1. T-Junction power combiner on LCP
83
The simulated results of the T
T-Junction
Junction power combiner is shown in Figure 4.3.2. As can
be seen, the input has a VSWR of better than 1.6:1 across the full W
W-band
band spectrum. The
excess insertion loss never exceeds 1.4 dB with a nominal value of 0.7 dB at 95 GHz.
The isolation between ports is poor reaching only -5
5 dB. Finally, the output ports show a
VSWR of better than 2.5:1 across the full band. Simulations were completed in CST
Microwave Studio.
Fig. 4.3.2
.3.2. Simulated S-Parameters of the T-Junction combiner
84
4.4 Substrate Integrated Waveguide Slot Antenna on LCP
As wireless technology improves, there exists a demand for low cost highly
integrated systems. Planar antennas that can be implemented on standard PCB
manufacturing processes then become increasingly important. Waveguides are the
transmission medium of choice at high frequencies having qualities such as low loss and
high power handling. However, in the W-band, waveguides are not compatible with the
cost and high density integration needs of modern wireless systems. Antenna arrays at 94
GHz have been demonstrated through the use of micromachining [2], but it is costly and
difficult to integrate with other microwave components.
The substrate integrated waveguide is one such transmission modality that shows
great promise especially in the millimeter wave regime as it maintains waveguide
characteristics such as low loss, and at the same time is compatible with the requirements
of low cost and ease of integration [5]. It has been shown in [6] that LCP is an ideal
substrate material showing low loss even in the W-Band. Unlike comparable materials
such as fused-silica quartz, LCP is not brittle. When working with fused-silica, the size of
the substrate, number of vias, and the pitch of the vias are limited by the mechanical
stability. However, LCP because of its polymer nature is not limited by any of those
circumstances, making it an excellent material to build SIW antennas on.
Work in the W-band, however, has been limited to antenna arrays which utilize
specialized manufacturing processes. For example a single W-Band slot antenna has been
demonstrated in [7]; however it requires the use of costly ceramic materials which suffers
from high sensitivity to the slot placement. Another antenna array is demonstrated in [8]
at 94 GHz, and although shows good performance, it uses a very advanced PCB
85
manufacturing process which is not commercially available. Finally, researchers have
demonstrated a SIW array at 79 GHz [9], utilizing nickel nano-wires to form the SIW
sidewalls in order to reduce energy leakage. This however, is not a standard PCB
manufacturing process and cannot accommodate volume applications.
In this section, a W-band 64 element SIW antenna array on LCP utilizing standard
PCB manufacturing processes is presented. Simulation results show a 10 dB return loss
bandwidth of 3.56 GHz, a gain of 18.95 dBi with a radiation efficiency of 57%. The
measured beammwidth is approximately 13.2o with a maximum sensitivity of
121mV/mW at 96 GHz. To the author’s knowledge this is the first time an unbalanced
millimeter wave antenna is measured utilizing a diode detector circuit.
Figure 4.4.1 shows the proposed single SIW slot antenna with its corresponding
dimensions. The antenna resides on a 4 mil thick LCP substrate with a dielectric constant
εr ≈ 2.94 and a loss tangent δ ≈ 0.002 and with 0.5 oz. double clad rolled copper foil. The
antennas operating frequency is determined by the length of the resonant slot, which is
approximately 0.64λg. The impedance match of the antenna is determined by the slot’s
offset. The distance from the end of the slot to the end of the waveguide is three quarter
of the guide wavelength.
The plated vias around the slot behave as a waveguide sidewall, confining
propagation between the sidewalls. The vias are 8 mils in diameter with a pitch of 24
mils. A tapered microstrip transition is employed to enhance continuity between the
feedline and the radiator reducing impedance mismatch. Figure 4.4.2 shows the simulated
return loss of the single slot antenna.
86
Fig. 4.4.1. Schematic of the SIW single slot antenna. Dimensions are: L = 3.4 mm, LTrans = 0.98 mm, WTrans
= 1.34 mm, LSlot = 1.2 mm, WSlot = 0.127 mm, WOffset = 0.722 mm, LOffset = 1.66 mm, DVia = 0.2032 mm,
GVia = 0.6096 mm
Fig. 4.4.2. Return loss of SIW single slot antenna
The designed SIW based eight-by-eight slot antenna array is shown in Figure
4.4.3. The geometries of the slots follow that from the dimensions in Figure 4.4.1. A slot
spacing of approximately half a guided wavelength is used. It can be seen that this array
is comprised of 8 longitudinal SIW antenna arrays. The spacing between the longitudinal
87
arrays is half a wavelength at a center frequency of 95 GHz to emulate phased array
applications where grating lobes need to be minimized. The corporate feed network
consists of seven identical 3dB T-Junction power combiners. Elliot [10] describes a
closed form equation relating the resonant frequency with the slot length and offset.
2!3'
24,567"8
92.09
=6 ,
=' >
?cos ?
0.464E='
=6
2
F cos0.464EF G *sin
EJ
,
+
2
Where Gslot is conductance of the slot, Gwaveguide is the conductance of the waveguide, ‘a’
and ‘b’ are the large and small dimensions of the waveguide, and x is the slot offset. This
equation describes the normalized slot conductance. This provides a practical starting
point where optimization routines in HFSS are used to fine tune the design.
Fig. 4.4.3. Schematic of the SIW eight-by-eight slot antenna array with corporate feed network
88
Manufacturing tolerances start to become an issue in the W-band where the
wavelength is so short. The primary concern in regards to the SIW antenna is etching of
the slots. A sensitivity analysis was conducted on the SIW array changing only the slot
width from 3 mils to 5 mils. Figure 4.4.4 shows the simulated return loss of the SIW slot
array varying the slot widths. Table I shows the simulation results and as can be seen, the
effects on antenna performance are only nominal. To ensure the etching of the slots, a 5
mils slot width was selected for the actual manufacturing of the antenna array. Simulation
shows a 10 dB return loss bandwidth of 3.56 GHz with a Radiation Efficiency of 57%
and a directivity of 18.95 dBi.
Since the height of the SIW is nominal, the dominant propagating mode should be
the TE10 mode. Figure 4.4.5 shows the 3D E-Field distribution for one longitudinal slice
of the antenna array at 96 GHz and it can be seen that the array indeed supports the
dominant TE10 mode.
Fig. 4.4.4. Return loss of the SIW array varying the slot widths
89
Fig. 4.4.5. 3D E-Field distribution for a single longitudinal slice
90
4.5 Series fed patch array on LCP
The series fed patch array is another topology that shows much promise. The
patch antenna itself is one of the most ubiquitous antennas found in almost every wireless
application both commericial and defense related. The patch antenna is a resonant
antenna whereby its operating frequency is dependent on the dimension of the rectangular
metal patch. The series fed patch array is essentially a 1-dimensional linear array with a
sequential linear feed line. Because of this reason, the radiation pattern of one series fed
patch array has a fan shaped beam pattern, wide in the plane along the array and narrow
in the plane perpendicular to the array. The series fed patch array is primarily useful for
beamforming applications where the beam is scanned in only one axis, either vertical or
horizontal [11]. Recent developments of the series fed patch array has yielded designs on
LCP that function up to the V-band (50 – 75 GHz) [12].
In this section, a W-band 64 element series fed patch array on LCP utilizing
standard PCB manufacturing processes is presented. Simulation results show a 10 dB
return loss bandwidth of 12 GHz, a gain of 20 dBi. The measured beammwidth is
approximately 13.2o with a maximum sensitivity of 250mV/mW at 95 GHz.
Figure 4.5.1 shows the proposed single series patch antenna array with its
corresponding dimensions. The antenna resides on a 4 mil thick LCP substrate. The
antennas operating frequency is determined by the length and width of the rectangular
patch, which is approximately 0.28λo x 0.35 λo, as well as the width of the feedline that
connects the antenna together. Simulations were completed in Sonnet EM simulation
software.
91
Fig. 4.5.1.. W-band
band 4 element series fed patch antenna array on LCP
Fig. 4.5.2.. Simulated return loss of 4 element series fed patch array
Figure 4.5.2
.5.2 shows the simulated input return loss of the antenna and it can be seen that
the 10 dB return loss bandwidth is 8 GHz. Figure 4.5.3
.5.3 shows the simulated radiation
patterns from Sonnet in both in the E- and H-planes (yz- and xz- axis respectively).
respectively) As
92
Fig. 4.5.3. Simulated radiation pattern of 4 element series fed patch array
observed, the radiation pattern has a fan-like beam shape, with a peak gain of 10.8 dBi.
From this linear 4-element array, a 16-element 2D array was designed with
interelemental spacings of approximately λo. The 16-element array is shown in Figure
4.5.4. A total of 3 T-junction combiners were used. Figure 4.5.5 shows the simulated
return loss and Figure 4.5.6 shows the simulated radiation patterns at 95 GHz. The larger
array still maintains a wide impedance bandwidth and as expected, the gain improves by
approximately 6 dB as the aperture size is quadrupled. It is observed that the E- and Hplane patterns are slightly assymetric in terms on beamwidth and sidelobes. The
assymetry in the beamwidth is attributed to the fact that the patch antennas are
rectangular, and therefore have assymetric radiation properties. The assymetry in the
sidelobe levels is attributed to the leakage radiation from the electrically large feed
network, which can degrade the pattern in the E-plane.
93
Fig. 4.5.4. W-band 16-element series fed patch array
Fig. 4.5.5. Simulated return loss of W
W-band 16-element
element series fed patch array
94
Fig. 4.5.6. Simulated radiation pattern of W-band 16-element series fed patch array
Finally a 64-element array was designed. The same inter-elemental spacing was
used and a total of 7 T-junction combiners were used. Not only does the electrically large
feed network contribute to unwanted radiation, the losses induced from just the
transmission lines can degrade the efficiency and gain. The 64-element series fed patch
array is shown in Figure 4.5.7. Figure 5.5.8 shows the simulated return loss and Figure
4.5.9 shows the simulated radiation patterns at 96 GHz. Symmetrical boundary conditions
were used to reduce the computation time required. It can be seen that the return loss
bandwidth actually improves with the feeding network. This is mainly due to the fact that
the T-Junction combiner provides a broadband impedance match. The simulated 10 dB
95
Fig. 4.5.7. W-band 64-element series fed patch array
Fig. 4.5.8. Simulated return loss of W
W-band 64-element
element series fed patch array
96
Fig. 4.5.9 Simulated radiation pattern of W-band 64-element series fed patch array
return loss bandwidth is approximately 12 GHz. The radiation patterns show a maximum
gain of 22 dBi, which corresponds to a 6 dBi improvement from the 16-element array.
This intuitively makes sense as the aperture size increased by four times. It is noted,
however, that Sonnet EM does not account for conductor losses, which can be quite
significant in the W-band. Moreover, it is also noted that since Sonnet EM is a 2.5D field
solver, the far field patterns are constructed through extrapolations, which are less
accurate. However, the computation time required as opposed to a full 3D field solver is
very much reduced.
97
4.6 Measurement Scheme using Zero-Bias Diode
Antenna measurements in the W-Band can be very cumbersome. Researchers
tend to adopt two methods; the first utilizing customized probe station setups in
conjunction with millimeter-wave extender modules. The second method is to utilize a
high frequency diode detector circuit, which provides a DC voltage output, to measure
the radiation patterns. Typical implementations of this method generally utilize self
complimentary or balanced antennas [13], where the diode can be placed directly in the
center without much complication. However, the SIW antenna array and series fed patch
array are singled ended antennas and therefore requires a specialized detector circuit [14].
The fully assembled SIW antenna array and series fed patch array with the detector
circuits are shown in Figure 4.6.1a,b.
The zero bias Schottky diode used in this work is the MZBD-9161GaAs beam
lead diode from Aeroflex, with typical device electrical parameters under forward bias of
Isat = 12 µA, Rs = 50 Ω, ideality factor, η = 1.2, and junction capacitance, Cjo = 0.03 pF.
The beam lead diode is mounted onto the antenna using the combination of an ultrasonic
wedge bonder and silver epoxy. In order to provide a DC return, a high impedance λ/2
open stub is placed at the anode, which is bonded from the middle (λ/4) to ground. The
bondwire acts as a RF Choke and in this instance a longer bondwire can actually be
advantageous.
In order to suppress RF signals from propagating to the output a low pass filter
(LPF) is employed. Typical distributed LPFs require a large size to obtain wide stopband
rejection. Therefore, a photonic bandgap structure (PBG) was used in order to obtain
wide stopband rejection with a minimal size [15].
98
(a)
(b)
Fig. 4.6.1. Fully assembled W-band antenna array detectors (a) SIW array (b) series fed patch array
99
(a)
(b)
Fig. 4.6.2. (a). Schematic of PBG LPF. Dimensions are: L1 = 0.71 mm, L2 = 0.81 mm, W1 = 0.76 mm, L3 =
0.71 mm, W2 = 1.37 mm, L4 = 0.81 mm, W3 = 0.15 mm (b). Sonnet simulated S-parameters
parameters
The designed LPF is shown in Figure 4.6.2a and the simulated S
S-parameter
parameter results from
Sonnet EM are shown in Figure 4.6.2b. As can be observed the PBG LPF has a rejection
of more than 50 dB at 95 GHz. This ser
serves
ves to improve the signal to noise (SNR) ratio of
the measurement system.
100
Fig. 4.6.3. Experimental measurement setup
Figure 4.6.3 shows the experimental setup block diagram for measuring the
radiation patterns. A back-wards-oscillator (BWO) is used as a W-band signal source for
the transmit horn antenna. A 3 dB WR-10 directional coupler attached to a HP W8486A
power sensor monitors the transmitted power level. A frequency meter serves to ensure
transmission at the correct frequency. Because the amplitude of the transmitted signal can
be quite low due to limitations of the BWO and losses incurred from free space, a lock-in
amplifier is employed in order to improve the sensitivity of the detection system. A 60 Hz
mechanical chopper is used to amplitude modulate the high frequency signal which is
necessary for the lock-in system. Finally, custom software interfaces the rotational
platform with a multimeter, which samples the voltage at various angles to measure the
radiation pattern of the antenna array under test. Figure 4.6.4 shows physical setup on the
101
transmit side. Figure 4.6.5 shows the physical setup on the receive side, which includes
the lock-in amplifier, oscilloscope for monitoring the 60 Hz transmitted and received
signal levels, and a multi-meter. Finally Figure 4.6.6 shows the SIW antenna array under
test inside of the anechoic chamber.
Fig. 4.6.4. Physical setup used for radiation pattern measurements on the transmit side
102
Fig. 4.6.5. Physical setup used for radiation pattern measurements on the receive side
Fig. 4.6.6. Antenna array under test inside the anechoic chamber
103
4.7 Measurement Results
LCP is commercially available through Rogers in two forms, as a core layer and
as a bonding layer. The proposed antenna array is fabricated using 4 mil thick
ULTRALAM 3850 from the Rogers corporation. The LCP material comes double clad in
0.5 Oz. copper foil. Laser penetration was used to form the 8 mil vias that form the SIW
side walls.
Measurements were taken inside an anechoic chamber. Figure 4.7.1 shows
measured radiation patterns of the 64-element SIW antenna array at 96 GHz. As can be
seen, the 3 dB beam width is approximately 13.2o and show symmetry in both the E and
H planes. The first major sidelobe is approximately 9 dB down from the main lobe in the
azimuth plane. In the elevation plane, the absence of clearly defined sidelobes is
indicative of an aperture distribution which is non-uniform. Due to motor limitations, the
antenna is scanned from ±60o. The simulated and measured radiation patterns match well.
(a)
104
(b)
Fig. 4.7.1. Measured and simulated radiation patterns of the SIW array in the (a) Azimuth (b) Elevation
planes
Antenna sensitivity measurements for the SIW array were made from 85 – 105
GHz, this is shown in Figure 4.7.2. The sensitivity of the antenna array detector is
defined as:
K#!""5"L M0N
OP0QNP
(1)
Where Vrms is the voltage output of the lock-in amplifier and Ptransmit is the incident RF
power. The antenna array with the detector has a peak sensitivity of 121mV/mW at
around 96 GHz. The narrow sensitivity characteristics correlate well with the simulated
return loss which has a minimum value also at 96 GHz and also shows a narrow
bandwidth.
105
Fig. 4.7.2. Measured sensitivity of SIW antenna array with diode detector
Figure 4.7.3 shows the measured radiation patterns of the 64-element series fed patch
array at 95 GHz. The beamwidth in the azimuth and elevation planes are near symmetric
with a measured beamwidth of approximately 13o. It is clear that in the elevation plane,
the feed network is distorting the radiation pattern. This can be remedied by either
creating an enclosure for the feed network or by placing absorbing foam in front of the
feed network to prevent radiation. Sensitivity measurements were also taken for the patch
array; this is shown in Figure 4.7.4. It can be seen that this array has a larger range in
which the sensitivity is high. This corresponds well with the simulated return loss which
shows a very broadband response. It is also noted that the peak sensitivity is much higher
in this array as compared to the SIW array. This is attributed to the fact that the series fed
patch array has a better impedance match to the zero-bias detector diode than the SIW
array. Sensitivity measurements were taken inside an anechoic chamber with a distance
of 2.5 meters from the transmit antenna to the receive antenna.
106
(a)
(b)
Fig. 4.7.3. Measured and simulated radiation patterns of the series fed patch array in the (a) Azimuth (b)
Elevation planes
107
Fig. 4.7.4.. Measured sensitivity of patch antenna array with diode detector
4.8 Conclusion
Two 64-element
element W
W-band
band antenna array detectors have been presented in this
chapter. Both the SIW array and series patch array are implements on LCP utilizing
standard PCB manufacturing processes. Demonstrated also is a measurement method for
unbalanced millimeter wave antennas utilizing a diode detector circuit. Measurement
results show a beamwidth of appr
approximately 13o in the E- and H-planes
planes for both antenna
arrays. The maximum sensitivity of the SIW detector array is 121mV/mW at 96 GHz.
The maximum sensitivity of the series patch array detector is 250mV/mW at 95 GHz.
The total size of the antenna array wit
with
h detector circuitry is 4 cm x 3 cm. It is
demonstrated that LCP can be utilized to extend the electrical performance of the next
generation millimeter wave phased array systems.
108
4.9 Acknowledgment
This chapter, in part, has resulted in the following publications:
[1] Chieh, J.S.; Pham, B.; Pham, A.V.; Kannell, G.; Pidwerbetsky, A., “A W-Band 8 x 8
Series Fed Patch Array on Liquid Crystal Polymer,” Antennas and Propagation Society
International Symposium (APSURSI), 2012 IEEE
[2: To Submit] Chieh, J.S.; Pham, A.V.; Pidwerbetsky, A; Kannell, G., " A Low Cost 8 x
8 W-Band Substrate Integrated Waveguide Antenna Array Detector on LCP," Antennas
and Propagation, IEEE Letters on
References
[1] Lockie, D.; Peck, D.; , "High-data-rate millimeter-wave radios," Microwave
Magazine, IEEE , vol.10, no.5, pp.75-83, August 2009
[2] Gauthier, G.P.; Raskin, J.-P.; Katehi, L.P.B.; Rebeiz, G.M.; , "A 94-GHz aperturecoupled micromachined microstrip antenna," Antennas and Propagation, IEEE
Transactions on , vol.47, no.12, pp.1761-1766, Dec 1999
[3] Shu Chen; Nguyen, K.N.; Afsar, M.N.; , "Complex Dielectric Permittivity
Measurements of Glasses at Millimeter Waves and Terahertz Frequencies," Microwave
Conference, 2006. 36th European , vol., no., pp.384-387, 10-15 Sept. 2006
[4] Smith, S.L.; Dyadyuk, V.; , "Measurement of the dielectric properties of Rogers
R/flex 3850 liquid crystalline polymer substrate in V and W band," Antennas and
Propagation Society International Symposium, 2005 IEEE , vol.4B, no., pp. 435- 438 vol.
4B, 3-8 July 2005
[5] Xiao-Ping Chen; Ke Wu; Liang Han; Fanfan He; , "Low-Cost High Gain Planar
Antenna Array for 60-GHz Band Applications," Antennas and Propagation, IEEE
Transactions on , vol.58, no.6, pp.2126-2129, June 2010
[6] Thompson, D.; Kirby, P.; Papapolymeron, J.; Tentzeris, M.M.; , " W-band
characterization of finite ground coplanar transmission lines on liquid crystal polymer
(LCP) substrates," Electronic Components and Technology Conference, 2003.
Proceedings. 53rd , vol., no., pp. 1652- 1655, May 27-30, 2003
[7] Stephens, D.; Young, P.R.; Robertson, I.D.; , "W-band substrate integrated waveguide
slot antenna," Electronics Letters , vol.41, no.4, pp. 165- 167, 17 Feb. 2005
109
[8] Garland, P.; Aguirre, J.; Hsueh-Yuan Pao; Hung-Sheng Lin; O'Neill, D.; Horton, K.; ,
"Manufacturing challenges for a W-band laminated waveguide phased array," Antennas
and Propagation Society International Symposium, 2008. AP-S 2008. IEEE , vol., no.,
pp.1-4, 5-11 July 2008
[9] Shi Cheng; Yousef, H.; Kratz, H.; , "79 GHz Slot Antennas Based on Substrate
Integrated Waveguides (SIW) in a Flexible Printed Circuit Board," Antennas and
Propagation, IEEE Transactions on , vol.57, no.1, pp.64-71, Jan. 2009
[10] Elliott, R.; , "An improved design procedure for small arrays of shunt
slots," Antennas and Propagation, IEEE Transactions on , vol.31, no.1, pp. 48- 53, Jan
1983
[11] Woosung Lee; Jaeheung Kim; Young Joong Yoon; , "Compact Two-Layer Rotman
Lens-Fed Microstrip Antenna Array at 24 GHz," Antennas and Propagation, IEEE
Transactions on , vol.59, no.2, pp.460-466, Feb. 2011
[12] Il Kwon Kim; Pinel, S.; Laskar, S.; Jong-Gwan Yook; , "Circularly & linearly
polarized fan beam patch antenna arrays on liquid crystal polymer substrate for V-band
applications," Microwave Conference Proceedings, 2005. APMC 2005. Asia-Pacific
Conference Proceedings , vol.4, no., pp. 3 pp., 4-7 Dec. 2005
[13] Lei Liu; Hesler, J.L.; Haiyong Xu; Lichtenberger, A.W.; Weikle, R.M.; , "A
Broadband Quasi-Optical Terahertz Detector Utilizing a Zero Bias Schottky
Diode," Microwave and Wireless Components Letters, IEEE , vol.20, no.9, pp.504-506,
Sept. 2010
[14] Moyer, H.P.; Bowen, R.L.; Schulman, J.N.; Chow, D.H.; Thomas, S.; Lynch, J.J.;
Holabird, K.S.; , "Sb-Heterostructure Low Noise W-Band Detector Diode Sensitivity
Measurements," Microwave Symposium Digest, 2006. IEEE MTT-S International , vol.,
no., pp.826-829, 11-16 June 2006
[15] Taesun Kim; Chulhun Seo; , "A novel photonic bandgap structure for low-pass filter
of wide stopband," Microwave and Guided Wave Letters, IEEE , vol.10, no.1, pp.13-15,
Jan 2000
110
Chapter 5
Dual Polarized Scalable W-band Phased Arrays on
LCP
In the last chapter it is evident that wireless systems operating in the E-band and
W-band benefit in terms of bandwidth and signal transmission clarity. Antennas and
antenna array architectures were presented, however, they were limited to scanning in
only one direction as the antenna arrays were linear. There exists, also, many noncommercial applications where advanced phased arrays in the W-band are needed.
Examples would include monopulse radars for tracking [1] – [3], polarimetric radars for
weather observation [4] – [8], and satellite communications (SATCOM) [9] – [11]. In
these instances the main benefit of operation in the W-band is the scaling of the physical
aperture size with frequency. In other words, for a given amount of physical space, the
beamwidth narrows as the frequency of operation increases. In addition, the channel
throughput can be doubled by utilizing dual concurrent polarizations. The challenge with
this is to achieve an acceptable cross polarization level where isolation between the
polarizations is large. This is particularly attractive for phased array systems because it
opens the possibility of obtaining a massively large high performance phased array with
hundreds of thousands of elements where real estate is limited.
111
There are challenges and they are multi-faceted and only recently has technology
matured to make this a real possibility. One of the main limitations is integration
scalability. In the past, most passive W-band components have been waveguide in nature
and active components utilizing vacuum tubes, both of which are bulky. In order to
achieve a truly scalable phased array, an architecture must be adopted where small fully
integrated tiles can be combined in order to build a larger array. As mentioned earlier in
previous sections, in order to avoid grating lobes in phased arrays, an inter-elemental
spacing of λ/2 must be used. As an example in the W-band, a 2 x 2 tile could not exceed
3.15 mm in length or width, precluding the use of any waveguides or vacuum tube
components.
Integrated circuit technologies, however, have matured in recent years and Wband circuits have been demonstrated on Silicon Germanium (SiGe) [12] – [15], Indium
Phosphide (InP) [16] – [18], and most recently on Gallium Nitride (GaN) [19], [20]. The
availability of millimeter-wave monolithic circuits in the W-band has truly opened the
doors to the possibility of building massively large scalable phased array antennas.
Although now possible, there exist two schools of thought on how this level of scalability
can be accomplished. In the first proposed method the active circuitry and antenna
elements are all integrated in a wafer-scaled package [21] – [23]. The benefit of this
method is that antenna elements are small enough at these frequencies to be integrated
without a large penalty and interconnect parasitics can be minimized. The main drawback
is the wafer-scaled packaging complexity which is both expensive and an immature
technology at the present. Another method which is more traditional, however has not
been explored much in literature is utilizing chip-on-board packaging using an organic
112
substrate for the antenna elements. This second method will be the primary topic of this
chapter.
5.2 Dual Polarized Antenna Topology
Dual polarized antennas offer the benefit of essentially adding an additional entire
channel. In effect one can think of a dual polarized antenna behaving as a diplexer,
however instead of multiplexing in the frequency domain, it is done through the
polarizations of the EM wave. This offers a huge benefit in that since the frequency of
operation can be re-used, transceiver complexity is reduced because components need to
only work at one frequency. For example a 2 x 2 dual polarized array tile could have both
transmit and receive capabilities concurrently, with a polarization dedicated to each
respectively. This however, comes with one large caveat, and that is the cross
polarization levels needs to be quite low in order for this assumption to hold.
For very large phased array applications, the tile architecture is the most popular.
In the tile arrangement, the antennas and associated circuitry are stacked vertically.
Figure 5.2.1 shows conceptually a W-band 2 x 2 array tile with the chip-on-board
package. As can be seen, it is important for the associated circuitry to reside on the
backside of the array. The 2 x 2 tiles can then be arrayed further to create a scalable and
large phased array system. The details regarding the phased array MMIC will not be
discussed as it goes beyond the scope of this dissertation. However, it is acknowledged
that the output frequency of the phased array MMIC will probably be at a much lower IF
(intermediate frequency) frequency. This is because designing a feed network in the W-
113
Fig. 5.2.1. Conceptual diagram showing a W-band 2 x 2 antenna array tile with a chip-on-board assembly
for the MMIC.
band to combine all the tiles would simply be too lossy and cancel any benefits from
beamforming.
In the W-band, the frequency is high and the wavelength is small. In the waferscaled packaging scheme, fine lithography and small vias in the order of microns is
available. When designing components on a printed circuit board, these fine resolutions
are simply not available and this creates a serious limitation. Because of this limitation,
an antenna feed utilizing a feedthrough via was neglected. Instead, a more elegant and
mechanically stable electromagnetic feed was used. The aperture coupled patch antenna
was first described by Pozar [24] and basically uses a slot in the ground plane to couple
energy from the microstrip feedline to the radiating patch. Countless permutations of this
114
simple
imple design has been reported in literature and some of the most significant
contributions include bandwidth improvement through the use of stacked patches [25]
and air cavities [26], achieving low cross polarization by using a differential feed [27],
and has been demonstrated in the millimeter wave regime in the V
V-band
band (57 – 67 GHz)
on LTCC [28] and even on LCP [29].
5.3 Low Isolation Design
Figure 5.3.1 shows the first of two dual polarized aperture fed stacked
stac
patch
antenna on LCP. Figure 5.3.2 shows the associated antenna geometries for each metal
layer. This first design utilizes two identical orthogonal slots in order to feed each
individual polarization. A stacked patch topology was adopted in order to extend the
bandwidth. In addition, an air cavi
cavity
ty was also used in order to lower the dielectric and
extend the bandwidth.
width. Figure 5.
5.3.3
.3 shows the LCP cross section used for the antenna.
Fig. 5.3.1.. Dual polarized aperture fed stacked patch on LCP/Kapton
115
Fig. 55.3.2. Antenna Geometries for various layers
Fig. 5.3.3. LCP/Kapton Stackup for Dual Polarized Aperture Fed Stacked Patch Antennas
116
As can be seen from Figure 5.3.2 and Figure 5.3.3, the antenna utilizes 4 metal
layers with dual dielectrics, both LCP and Kapton. As mentioned previous, LCP is a
stable dielectric that has been characterized up to 110 GHz [30], [31]. Therefore, LCP
was chosen as the substrate material used for the microstrip feed on the top metal layer. A
2 mil thick LCP substrate was chosen in order to both minimize loss and trace widths for
the feedlines. The next metal layer forms the ground plane for the microstrip feedline as
well as the aperture slots that couple energy from the feedline to the metal patches. To
achieve dual polarizations in both the vertical and horizontal orientations, orthogonal
slots were designed and optimized. A “dog bone” shape was adopted in order to improve
the coupling efficiency [32]. The length, width and position of the slot all impact the
bandwidth. Generally speaking, when the slot is positioned at the center of the patch, it
achieves a much larger bandwidth. However, because dual orthogonal slots are necessary,
the slots were positioned near the edges of the patch offset from the center. Recursive
routines in HFSS were used to optimize the size and position of the slot for the largest
bandwidth. Beneath the aperture/ground metal layer is another 2 mil thick LCP substrate
for which the driven patch resides. The driven patch is approximately 30 mils x 30 mils.
Beneath the driven patch layer is a superstrate layer for the air cavity. It has been widely
reported that the bandwidth of the patch antenna is proportional to the height of the
dielectric substrate and the dielectric constant of the substrate. The relationship is as
follows:
R4 ∝
T-UTP
√ VW
(1)
For this reason, low dielectric materials are preferred, in fact many times standoffs are
used in order to suspend the patch elements in air, which has a εr = 1 [33]. For millimeter
117
wave designs, standoffs are not appropriate and therefore a superstrate with a drilled air
cavity is preferred. In this design, we have adopted an interleaved Kapton and LCP
dielectric material for the superstrate layer. The reason being that during lamination, LCP
has a tendency to flow, therefore filling the drilled cavities. In order to circumvent this,
Kapton adhesive was used to adhere the superstrate layer to the LCP layers. Kapton has a
much lower lamination temperature and has the added benefit of having very similar
dielectric properties with a εr ~ 3.2 and a loss tangent of δ ~ 0.0004. The height of the air
cavity is 8 mils. Finally, beneath the superstrate layer is the parasitic patch layer. The size
of the parasitic patch is 0.95 mm x 0.95 mm. A thin 2 mil Kapton dielectric layer acts as a
mechanical support membrane layer for the parasitic metallic patch. It was found that
thicknesses in excess of 2 mils would degrade the electrical performance of the antenna.
Figures 5.3.1 and 5.3.2 show the dual polarized antenna with a T-Junction in order to
receive both polarizations concurrently. In order to investigate the performance of the
polarizations independently, an alternative configuration was also used, shown in Figure
5.3.4 where the feedlines are separate. From henceforth, the x-axis will be referred to as
the vertical polarization and the elevation plane and the y-axis will be referred to as the
horizontal polarization and the azimuth plane.
The simulated return loss is shown in Figure 5.3.5 and as can be seen, the
response is symmetric between the two polarizations. Although the return loss creeps
slightly above -10 dB at around 100 GHz, the 2:1 VSWR bandwidth is 35 GHz,
corresponding to a fractional bandwidth of about 37%. The strategy in improving the
bandwidth comes primarily from the use of the stacked patches. As can be seen, there are
two major dips in the return loss plot. The resonance associated with the first dip is
118
Fig. 5.3.4. HFSS simulation model with independent polarization feedlines
Fig. 5.3.5. Simulated return loss for both vertical (red) and horizontal (purple) polarizations
119
attributed to the parasitic patch, which is larger in size. The resonance associated with the
second dip is attributed to the driven patch. When the two resonances are placed close to
each other, they can overlap, extending the overall bandwidth of the antenna. The
impedance matching is affected primarily by several parameters, the width and length of
the aperture slot, and the length of the open stub feedline. Although the slot length can be
tuned, the width is set by the minimum tolerance achievable by manufacturing. The slot
length and open stub length were optimized in HFSS through recursive algorithms.
The isolation of this design isn’t very high because the orthogonal feeds are quite
close, and therefore there exists a high level of coupling. Figure 5.3.6 shows the
simulated isolation between the two orthogonal ports. As can be seen, it achieves better
than 12 dB isolation across the full W-band, however this is generally not low enough.
Radiation pattern simulations were also completed in HFSS. Figure 5.3.7 shows
the simulated radiation patterns for the vertical polarization in both planes. As can be
seen, the beamwidth is approximately 70o and the cross-polarization level is down by 11
dB from the co-polarized pattern in both the azimuth and elevation planes. It is noted that
the asymmetric location of the aperture slot feed contributes greatly to the high crosspolarization levels. The asymmetry enhances radiating effects from the “non-radiating”
edges of the patch antenna, introducing cross-polarization. Figure 5.3.8 shows the
simulated radiation patterns for the horizontal polarization in both planes and the results
are very similar as expected. The simulated gain for both polarization orientations at 95
GHz is 7.65 dBi. For polarimetric radar applications, studies have indicated that the bare
minimum polarization isolation should be around -15 dB while the target should be
around -30 dB for good performance [35].
120
XY Plot 2
HFSSDesign1
ANSOFT
-5.00
Curve Info
dB(St(Rectangle104_T1,Rectangle104_T2))
dB(St(Rectangle104_T1,Rectangle104_T2))
Setup1 : Sw eep
-7.50
-10.00
-12.50
-15.00
-17.50
-20.00
-22.50
70.00
80.00
90.00
100.00
110.00
Freq [GHz]
Fig. 5.3.6. Simulated isolation between orthogonal polarization ports
(a)
(b)
120.00
121
Fig. 5.3.7. Simulated vertical co- and cross-polarized radiation patterns in the
(a) azimuth (b) elevation planes
(a)
(b)
Fig. 5.3.8. Simulated horizontal co- and cross-polarized radiation patterns in the
(a) azimuth (b) elevation planes
5.4 High Isolation Design
As seen in the last section, although that design achieves a large bandwidth and
high gain, the isolation is low and the cross polarization levels are high. In order to
improve on the isolation and cross polarization levels, an improved design was completed
which utilizes a very simple method to achieve feed polarization isolation. The simplest
way to increase the isolation between the horizontal and vertical polarizations is to use a
differential feed for one of the polarizations [36], [37]. In order to realize this, two
coupling slots are used for the horizontal polarization on opposite sides of the patch.
122
Since the coupling slots
lots are on opposite sides, they are 180º out of phase with each other.
In order to recover thee phase for the horizontal orientation, a T-junction
junction combiner is used
where one side has an additional λ/2
/2 transmission line, in effect acting as a simple balun.
When there is coupling from the vertical orientation, because of the symmetry and the
balun feed,
ed, effects from the coupling are cancelled. Figure 5.4.1 shows a 3D depiction of
the improved design. The cross
cross-section
section used remains the same, and the only difference is
in the aperture layer and the microstrip feed layer. Figure 5.4.2 shows the associated
associate
antenna geometries for each layer.
Fig. 5.4.1. High isolation dual polarized aperture fed stacked patch on LCP/Kapton
123
Fig. 55.4.2. Antenna Geometries for various layers
Figure 5.4.3 shows intuitively how the balun feed improves the performance of
the patch antenna. As shown in Figure 5.4.3a, since the aperture slot positions are
symmetric, the amount of coupling from the vertical polarization feed to each of the two
horizontal polarization feeds are the same. By implementing a 180º phase shift, the
effects of the coupling are out of phase and when combined, cancel entirely. This is the
mechanism whereby isolation of the feed ports can be improved. Figure 5.4.3b,c shows a
cavity model of the patch antenna when two different modes are excited. The TM010
mode is the dominant mode, and as can be seen, the edges parallel to the aperture slots
are the radiating edges and are opposite in field polarity. The orthogonal edges are
referred to as the non-radiating
radiating edges because the fields are 180º out of phase
pha and so any
corresponding radiation cancels each other out. In order to radiate effectively, a 180º
124
phase shift must be introduced at the radiating edges so that the contributions combine inin
phase. Higher modes also can be excited, and are the primary re
reason
ason for high crosscross
polarization. As an example, Figure 5.4.3c shows one higher order mode, the TM002
mode. When this mode is excited, the radiating edge is now orthogonal to the slots,
inciting reception of cross
cross-polarized
polarized signals. This, however, can be mitigated
m
also
through the introduction of a 180º shift. As can be seen, since the contributions from the
higher order modes are in
in-phase,
phase, after the 180º phase shift, the contributions are
combined out-of-phase
phase and the effects are cancelled.
Fig. 5.4.3. Schematic of balun fed patch (a) isolation (b) primary mode (c) higher order mode
125
Fig. 5.4.4. Simulated return loss for both vertical (purple) and horizontal (red) polarizations
Once again, simulations were completed in HFSS and Figure 5.4.4 shows the
simulated return loss for each polarization. The return loss response is no longer
symmetric, and this is no surprise, as the feeds are asymmetric. In the horizontal
polarization, the 10 dB return loss bandwidth is 37 GHz corresponding to a fractional
bandwidth of 39%. In the vertical polarization the antenna achieves a 10 dB return loss
bandwidth of 21 GHz, corresponding to a fractional bandwidth of 22%.
More importantly, we expect that the isolation will be improved by adopting this
topology. Figure 5.4.5 shows the simulated isolation between the two polarizations. As
can be seen, the isolation is better than 50 dB at 100 GHz, and better than 20 dB from 87
– 112 GHz. The reason the isolation has a trough response and why the cancellation isn’t
more broadband
adband is because of the type of balun used. The T
T-junction
junction along with the half
126
Isolation
Optimized_Iso_Patch
ANSOFT
0.00
-10.00
Y1
-20.00
-30.00
-40.00
-50.00
Curve Info
dB(St(Signal2_4_1_T1,Rectangle151_T1))
Setup1 : Opt_Sw eep
dB(St(Rectangle151_T1,Signal2_4_1_T1))
Setup1 : Opt_Sw eep
-60.00
50.00
62.50
75.00
87.50
100.00
Freq [GHz]
112.50
125.00
137.50
Fig. 5.4.5. Simulated isolation between orthogonal polarization ports
wave phase shift typically exhibits a 10% bandwidth, and therefore with a more
sophisticated balun, one could possibly achieve a more broadband cancellation.
With the increased isolation, we also expect that the cross-polarization levels will
decrease. Figure 5.5.6 shows the simulated radiation patterns for the vertical polarization
in both planes. As can be seen, the beamwidth remains approximately 70o however the
cross-polarization level is down by more than 25 dB from the co-polarized pattern in both
the azimuth and elevation planes. Figure 5.4.7 shows the simulated radiation patterns for
the horizontal polarization in both planes and the results are very similar as expected. The
simulated gain for both polarization orientations at 95 GHz is 7 dBi. Utilizing the
differential feed, both the isolation and the cross-polarization levels are improved. The
main penalty of using this topology is the increased area due to the addition of the balun.
For phased arrays where the inter-elemental spacing can be greater than λ/2, this design
would apply. A more sophisticated balun could also be used to improve the bandwidth
decrease the size; however that is left for future work.
127
(a)
(b)
Fig. 5.3.6. Simulated vertical co- and cross-polarized radiation patterns in the
(a) azimuth (b) elevation planes
(a)
(b)
Fig. 5.3.7. Simulated horizontal co- and cross-polarized radiation patterns in the
(a) azimuth (b) elevation planes
128
5.5 2 x 2 Antenna Array Tile Concept
For many phased array applications, there exists a need to steer the beam at a
maximum angle from broadside. In order to achieve this without the detrimental effects
of grating lobes, the inter-elemental spacing must be less than λ/2. As mentioned earlier,
at the W-band, this corresponds to a spacing of 1.57 mm, which leaves little room for
both chip-on-board assembly and signal trace routing. Although we have seen the
benefits of using a differential feed, the T-junction balun occupies a large area and unless
implemented differently, would be impractical. There exists, however, a very elegant
method to suppress the cross-polarization levels when the antennas are arranged in an
array format. Previous sections have focused on reducing the cross-polarization on the
individual antenna element level. However, the same methods taking advantage of field
symmetry can also be used in an array configuration to achieve cross-polarization
cancellation.
The first instance of array symmetry being used to achieve cross-polarization
suppression is reported in [38] and [39]. Various orientations were chosen, with zero,
single, and dual axis of symmetry for a 2 x 2 tile array. Figure 5.5.1 shows the base line
model that has no symmetry and a model with two axes of symmetry. The method for
which cross-polarization cancellation is achieved is the same as with the individual
antenna element. When there is an axis of symmetry, the vertically polarized fields from
two separate antennas are 180º out of phase and require a balun to recover the phase.
However, because the horizontal fields are in phase, all contributions are cancelled after
the 180º phase shift through the balun. Figure 5.5.2 illustrates this concept pictorially.
129
(a)
(b)
Fig. 5.5.1. 2 x 2 tile array (a) base line model (b) dual axis of symmetry model
Fig. 5.5.2. Illustration of higher mode cancellation through symmetry (a) dominant (b) high order mode
130
As can be seen, in the dominant TM010 mode, the fields at the radiating edges are 180º out
of phase. At the non-radiating edges, the fields cancel and therefore no radiation is
emitted from these sides. In reality, because of imperfections in manufacturing, perfect
symmetry is not attainable, and so therefore cross-polarization due to these “nonradiating” edges are actually finite, although minimal. Through the 180º phase shift, the
main components are combined in-phase. Alternatively, when higher modes are excited,
such as the TM002 mode, the fields on all sides are in-phase. Through the addition of the
180º phase shift, effects of these higher order modes are cancelled out.
However, because this technique utilizes symmetry at the array level, the
introduction of phase shifts to steer the beam would degrade the amount of suppression
achievable. For this reason, most literature using this technique has used it for fixed beam
arrays, where beam steering is not necessary. If, however, suppression was accomplished
at the individual antenna level, the introduction of phase shifts would not degrade the
suppression level. Regardless, the configuration with dual axis of symmetry lends itself
nicely to the packaging of a 2 x 2 array tile. The main benefit is that since the tile is
symmetrical, signal routing becomes much easier.
Simulations in HFSS were completed in order to validate the benefits of utilizing
a symmetrical tile configuration and to investigate the amount of degradation experienced
by steering the beam away from broadside. Figure 5.5.3 shows the HFSS simulation
model for the base line model and Figure 5.5.4 shows the simulation model for the dual
symmetric model. In this study, only one polarization was analyzed as the orthogonal
polarization is completely symmetrical.
131
(a)
(b)
Fig. 5.5.3. HFSS simulation model for non-symmetric 2 x 2 tile (a) top (b) bottom side
(a)
(b)
Fig. 5.5.4. HFSS simulation model for dual symmetric 2 x 2 tile (a) top (b) bottom side
132
Figure 5.5.5 shows the simulated radiation pattern in the elevation plane for the
vertical polarization at broadside for both the base tile and the dual symmetric tile. As can
be seen, the cross-polarization for the base tile is only down by 11 dB whereas for the
dual symmetric tile, the cross-polarization is down by more than 50 dB, a significant
difference. The gain, front to back ratio, and beamwidth remain the same for both
models. Figure 5.5.6 shows the simulated patterns when both arrays are steered 10º off
broadside. In the case of the base tile, the cross-polarization is still around 11 dB down
whereas for the dual symmetric tile, the cross-polarization is down by more than 40 dB.
Although the performance is still good, 10 dB of degradation is suffered. Figure 5.5.7
shows the simulated patterns when both arrays are steered 30º off broadside. In this
instance, the base tile still maintains a cross-polarization level of 11 dB down from the
primary polarization. In the dual symmetric tile, the cross-polarization continues to
degrade to around 20 dB down from the co-polarized pattern. Finally, Figure 5.5.8 shows
the simulated patterns when both arrays are steered 45º off broadside. For the base tile,
the cross-polarization levels are down by 9 dB from the co-polarized pattern. For the dual
symmetric tile, the cross-polarization levels are down by 16 dB from the co-polarized
pattern. The trend is that as the beam is steered farther and farther away from broadside,
the cancellation gets worse. This intuitively makes sense because as the beam is steered
farther from broadside, the progressive phase shifts between elements get larger and
larger, adding to the phase imbalance. This scheme relies heavily on phase symmetry,
and so any imbalance would degrade the amount of cross-polarization cancellation
achievable. However, that said, in all cases, the dual symmetric tile still shows superior
performance in terms of cross-polarization suppression.
133
(a)
(b)
Fig. 5.5.5. Simulated radiation pattern at broadside (a) base line model (b) dual symmetric model
(a)
(b)
Fig. 5.5.6. Simulated radiation pattern at 10º (a) base line model (b) dual symmetric model
134
(a)
(b)
Fig. 5.5.7. Simulated radiation pattern at 30º (a) base line model (b) dual symmetric model
(a)
(b)
Fig. 5.5.8. Simulated radiation pattern at 45º (a) base line model (b) dual symmetric model
135
5.6 Antenna Prototypes and Measurement Results
As discussed in section 5.4, the high isolation design is more superior when size
and antenna spacing is not an issue. For this reason, the high isolation design was chosen
for prototyping. As mentioned previously, the antennas were prototyped using the multilayer board stackup from Figure 5.3.3 and with standard PCB manufacturing techniques.
As the operating frequency increases, the tolerances of the manufacturing processes
become increasingly important. For this specific antenna, alignment of the driven patch to
the parasitic patch is critical and can greatly affect the impedance bandwidth. Figure 5.6.1
shows the PCB process used to construct the multi-layer LCP/Kapton board. Step 1 was
to laminate and etch the feedline layer, aperture layer, and driven patch layer. These all
reside on a homogenous LCP stackup and so were processed together. Step 2 was to
laminate and laser drill the air cavities for the superstrate layer. The superstrate was
processed as a sub-assembly because for the laser drill step, a solid metal plane is needed
at the bottom of the cavity. Step 3 was to etch the parasitic patch layer. Finally, Step 4
was to laminate all the sub-assemblies together with the Kapton adhesive. This was done
because the Kapton adhesive cures at a much lower melting point than LCP, keeping the
LCP from melting and filling the air cavities. Figure 5.6.2 shows a close-up top view of a
single antenna element prototype. As can be seen, the embedded air cavity is clearly
depicted from the charred edges from the laser, and it can be seen that the parasitic patch
is not perfectly aligned. Figure 5.6.3 shows an 8-element linear array that is severely
misaligned, taken from a corner edge of the PCB panel. This just reinforces the fact that
manufacturing tolerances need to be controlled precisely.
136
Fig. 5.6.1. PCB manufacturing process for LCP/Kapton antenna prototypes
Fig. 55.6.2. Top view of a single antenna element prototype
137
Fig. 5.6.3. Top view of a 88-element
element linear antenna array that has been misaligned
As discussed in section 4.6, making antenna measurements in the W-band
W
can be
quite challenging, and therefore a diode detector method for measuring the antennas
patterns was adopted. Figure 5.6.
5.6.4 shows the test boards for a single antenna element in
both the horizontal and vertical polarizations. Immediately following the antenna feedline
is a diode detector assembly with a LPF at the output. The zero bias Schottky
Schottk diode used
in this work is the HSCH
HSCH-9161GaAs beam lead diode from Agilent,, with typical device
electrical parameters under forward bias of Isat = 12 µA, Rs = 50 Ω,, ideality factor, η =
1.2, and junction capacitance, Cjo = 0.035 pF. The beam lead diode iss mounted onto the
antenna using the combination of an ultrasonic wedge bonder and silver epoxy. In order
to provide a DC return, a high impedance λ/2
/2 open stub is placed at the anode, which is
138
(a)
(b)
Fig. 5.6.4. Test boards for the single antenna element for both the (a) horizontal (b) vertical polarizations
bonded from the middle ((λ/4)
/4) to ground. The bondwire acts as a RF Choke and in this
instance a longer
er bondwire can actually be advantageous.
A measurement setup was previously discussed in section 4.6. In that setup,
measurements were taken inside an anechoic chamber. This was possible because the
antenna under test (AUT) had a moderately sized equivalent
nt aperture, compensating for
the path loss in the 3 meter chamber. However, for a single antenna element, with a
effective aperture size less than 2mm, a 3m path loss would degrade the SNR so much so
that even a lock-in
in amplifier could not recover the sig
signal.
nal. Furthermore, alignment of the
AUT inside a chamber would also be more difficult. For this reason, a bench-top
bench
setup is
preferred, where the distance between the transmit antenna and receive antenna can be
adjusted. Figure 5.5.5 shows the block diagram of the measurement setup. In this case,
the 60 Hz mechanical chopper was replaced by directly modulating the BWO with a 1.8
kHz square wave. Figures 5.6.
5.6.6 and 5.6.7 show the actual measurement laboratory setup.
139
Fig. 55.6.5. Block diagram of measurement setup
Fig. 5.6.6. Physical measurement setup
140
Fig. 5.6.7. Close-up of measurement setup
Figure 5.6.8 shows the measured H-plane radiation patterns for the single antenna
element in the vertical polarization at 95, 80, and 105 GHz. At 95 GHz, the
t center
frequency, the single antenna in the vertical polarization achieves a beamwidth of
approximately 90º, which correlates well simulation results. The measured crosscross
polarization levels are down by 25 dB from ±90º. It is observed that at angles greater
grea than
65º, the radiation pattern has a steep drop
drop-off.
off. This is mainly attributed to scattering
effects from various metal surfaces that are present in the table
table-top
top setup. It can be
observed that as the frequency increases, the beamwidth decreases. Figure
Figu 5.6.9 shows
the measured H-plane
plane radiation patterns for the single antenna element in the horizontal
polarization at 95, 80, and 105 GHz. At 95 GHz, the single antenna in the horizontal
141
(a)
(b)
142
(c)
Fig. 5.6.8. Measured and simulated H-plane radiation patterns in the vertical polarization at (a) 95 (b) 80
(c) 105 GHz
(a)
143
(b)
(c)
Fig. 5.6.9. Measured and simulated H-plane radiation patterns in the horizontal polarization at (a) 95 (b) 80
(c) 105 GHz
144
(a)
(b)
Fig. 5.6.10. Test boards for an 8-element
element array in both the (a) horizontal (b) vertical polarizations
es a beamwidth of 90º, and again correlates well with simulation. The
polarization achieves
measured cross-polarization
polarization levels are down by 20 dB from ±90º.
In addition to the single antenna element, an 88-element
element linear array was also
prototyped. Figure 5.6.10 shows the test boards for the 8-element
element linear array in both the
horizontal and vertical polarizations. Figure 5.6.11 shows the measured H-plane
H
radiation
patterns in the vertical polarization at 95, 80, and 105 GHz. Measured results at 95 GHz
show a beamwidth of 13º wi
with a cross-polarization
polarization level of less than 28 dB for ±90º. The
sidelobes are also clearly defined and below the main lobe by more than 15 dB. The
beam patterns are also very well defined at 80 and 105 GHz. Figure 5.6.12 shows the
measured H-plane
plane radiation patterns in the horizontal polarization at 95, 80, and 105
GHz. Measured results at 95 GHz show a beamwidth of 13º with a cross-polarization
cross
level of less than 20 dB for ±90º. The sidelobes are clearly defined and below the main
lobe by more than 13 dB. A
All
ll pattern measurements of the linear array correlate well with
simulation results and further validate the operation of the single antenna element.
element
145
(a)
(b)
146
(c)
Fig. 5.6.11. Measured and simulated H-plane radiation patterns in the vertical polarization at (a) 95 (b) 80
(c) 105 GHz
(a)
147
(b)
(c)
Fig. 5.6.12. Measured and simulated H-plane radiation patterns in the horizontal polarization at (a) 95 (b)
80 (c) 105 GHz
148
5.6 Conclusion
We have presented the design and development of wideband dual polarized
antennas on a LCP Kapton hybrid substrate. An aperture fed stacked patch antenna with
an air cavity is utilized to extend the bandwidth. Two designs are presented, a low
isolation design and a high isolation design. Both designs achieve a simulated gain of
greater than 7 dBi with an simulated impedance bandwidth of greater than 21 GHz. A
study was conducted for a 2 x 2 phased array tile concept. A methodology for crosspolarization suppression was proposed and simulation results were presented. Finally
fabricated prototypes using standard PCB manufacturing processes were presented.
Radiation pattern measurements were conducted at 80, 95, and 105 GHz for a single
antenna element as well as an 8-element linear array. Measurements were compared to
simulation results and correlate well. Measurements indicate a beamwidth of 90º and 13º
for the single antenna element and the 8-element linear array respectively.
149
References
[1] Storkus, W.L.; , "Design techniques for compact monopulse antenna feeds for Wband radar systems," Microwave Symposium Digest, 1990., IEEE MTT-S International ,
vol., no., pp.805-808 vol.2, 8-10 May 1990
[2] Toulios, P.; Yong-Hui Shu; Navarro, J.; Knox, R.; , "W-band integrated monopulse
radar transceiver," Microwave Symposium Digest, 1995., IEEE MTT-S International ,
vol., no., pp.423-426 vol.2, 16-20 May 1995
[3] Raman, S.; Barker, N.S.; Rebeiz, G.M.; , "A W-band dielectric-lens-based integrated
monopulse radar receiver ," Microwave Theory and Techniques, IEEE Transactions on ,
vol.46, no.12, pp.2308-2316, Dec 1998
[4] Holliday, R.; Rhys-Roberts, M.; Wynn, D.A.; , "A lightweight, ultra wideband
polarimetric W-band radar with high resolution for environmental applications," Radar
Conference, 2006. EuRAD 2006. 3rd European , vol., no., pp.194-197, 13-15 Sept. 2006
[5] Sung-Hyun Kim; Jin-Taek Seong; Hyuk Park; Ho-Jin Lee; Yong-Hoon Kim; Valeriy,
O.; Alexander, D.; , "W-band 2-D Scanning Fully Polarimetric Radiometer System for
Remote Sensing Applications," Geoscience and Remote Sensing Symposium, 2008.
IGARSS 2008. IEEE International , vol.2, no., pp.II-645-II-648, 7-11 July 2008
[6] Frasier, S.J.; Venkatesh, V.; Orzel, K.; Hartley, T.; Salazar, J.; Medina, R.; Knapp, E.;
Ibe, O.; Bluestein, H.B.; Snyder, J.; Tanamachi, R.; , "X- and W-band mobile Doppler
radar observations from VORTEX2 and current developments," Radar Conference
(RADAR), 2011 IEEE , vol., no., pp.774-777, 23-27 May 2011
[7] Aydin, K.; Singh, J.; , "Identification of cloud ice crystals using a 95 GHz
polarimetric radar," Geoscience and Remote Sensing Symposium, 2002. IGARSS '02.
2002 IEEE International , vol.5, no., pp. 2814- 2816 vol.5, 2002
[8] Pazmany, A.L.; Mead, J.B.; Galloway, J.; McIntosh, R.E.; Kelly, R.D.; Vali, G.; , "A
95 GHz airborne radar for high resolution polarimetric cloud measurements," Geoscience
and Remote Sensing Symposium, 1994. IGARSS '94. Surface and Atmospheric Remote
Sensing: Technologies, Data Analysis and Interpretation., International , vol.1, no.,
pp.424-426 vol.1, 8-12 Aug 1994
[9] Sacchi, C.; Musso, M.; Gera, G.; Regazzoni, C.; De Natale, F.G.B.; Jebril, A.;
Ruggieri, M.; , "An efficient carrier recovery scheme for high-bit-rate W-band satellite
communication systems," Aerospace Conference, 2005 IEEE , vol., no., pp.1379-1390, 512 March 2005
150
[10] De Luise, A.; Paraboni, A.; Ruggieri, M.; , "Satellit
"Satellitee communications in W-band:
W
experimental set-up
up for channel characterization," Aerospace Conference, 2004.
Proceedings. 2004 IEEE , vol.1, no., pp. 6 vol. (xvi+4192), 6-13
13 March 2004
[11] De Fina, S.; Ruggieri, M.; Bosisio, A.V.; , "Exploitation of the W-band
W
for high
capacity satellite communications," Aerospace and Electronic Systems, IEEE
Transactions on , vol.39, no.1, pp. 82
82- 93, Jan. 2003
[12] May, J.W.; Rebeiz, G.M.; , "High
"High-performance W-band
band SiGe RFICs for passive
millimeter-wave
wave imaging," Radio Frequency Integrated Circuits Symposium, 2009. RFIC
2009. IEEE , vol., no., pp.437
pp.437-440, 7-9 June 2009
[13] Shih, S.E.; Duan, D.W.; Fordham, O.; Tornquist, K.; Zeng, X.; Chang-Chien,
Chang
P.;
Tsai, R.; , "A W-Band
Band 44-Bit
Bit Phase Shifter in Multilayer Scalable Array
Arra
Systems," Compound Semiconductor Integrated Circuit Symposium, 2007. CSIC 2007.
IEEE , vol., no., pp.1-4,
4, 14
14-17 Oct. 2007
[14]] Sarkas, I.; Khanpour, M.; Tomkins, A.; Chevalier, P.; Garcia, P.; Voinigescu, S.P.; ,
"W-band 65-nm
nm CMOS and SiGe BiCMOS trans
transmitter
mitter and receiver with lumped I-Q
I
phase shifters," Radio Frequency Integrated Circuits Symposium, 2009. RFIC 2009.
IEEE , vol., no., pp.441-444,
444, 77-9 June 2009
[15] Rebeiz, G.M.; May, J.; Uzunkol, M.; Shin, W.; Inac, O.; Chang, M.; , "Towards
high-performance
mance > 100 GHz SiGe and CMOS circuits," Microwave Symposium Digest
(MTT), 2010 IEEE MTT--S International , vol., no., pp.1320-1323, 23-28
28 May 2010
[16] Nakasha, Y.; Sato, M.; Tajima, T.; Kawano, Y.; Suzuki, T.; Takahashi, T.;
Makiyama, K.; Ohki, T.; Hara, N.; , "
-band
band Transmitter and Receiver for 10-Gb/s
10
Impulse Radio With an Optical
Optical-Fiber Interface," Microwave Theory and Techniques,
IEEE Transactions on , vol.57, no.12, pp.3171
pp.3171-3180, Dec. 2009
[17] Elgaid, K.; McLelland, H.; Stanley, C.R.; Thayne, I.G
I.G.;
.; , "Low noise W-band
W
MMMIC amplifier using 50nm InP technology for millimeterwave receivers
applications," Indium Phosphide and Related Materials, 2005. International Conference
on , vol., no., pp.523-525,
525, 88-12 May 2005
[18] Grundbacher, R.; Lai, R.; Barsky, M.; Tsai, R.; Gaier, T.; Weinreb, S.; Dawson, D.;
Bautista, J.J.; Davis, J.F.; Erickson, N.; Block, T.; Oki, A.; , "0.1 µm
m InP HEMT devices
and MMICs for cryogenic low noise amplifiers from X
X-band
band to W-band,"Indium
W
Phosphide
de and Related Materials Conference, 2002. IPRM. 14th , vol., no., pp. 455455 458,
2002
[19] Masuda, S.; Ohki, T.; Makiyama, K.; Kanamura, M.; Okamoto, N.; Shigematsu, H.;
Imanishi, K.; Kikkawa, T.; Joshin, K.; Hara, N.; , "GaN MMIC amplifiers for W-band
W
transceivers," Microwave Integrated Circuits Conference, 2009. EuMIC 2009.
European , vol., no., pp.443
pp.443-446, 28-29 Sept. 2009
151
[20] Micovic, M.; Kurdoghlian, A.; Shinohara, K.; Burnham, S.; Milosavljevic, I.; Hu,
M.; Corrion, A.; Fung, A.; Lin, R.; Samoska, L.; Kangaslahti, P.; Lambrigtsen, B.;
Goldsmith, P.; Wong, W.S.; Schmitz, A.; Hashimoto, P.; Willadsen, P.J.; Chow, D.H.; ,
"W-Band GaN MMIC with 842 mW output power at 88 GHz," Microwave Symposium
Digest (MTT), 2010 IEEE MTT-S International , vol., no., pp.237-239, 23-28 May 2010
[21] Atesal, Y.A.; Cetinoneri, B.; Chang, M.; Alhalabi, R.; Rebeiz, G.M.; , "MillimeterWave Wafer-Scale Silicon BiCMOS Power Amplifiers Using Free-Space Power
Combining," Microwave Theory and Techniques, IEEE Transactions on , vol.59, no.4,
pp.954-965, April 2011
[22] Byung-Wook Min; Rebeiz, G.M.; , "W-band low-loss wafer-scale package for RF
MEMS," Gallium Arsenide and Other Semiconductor Application Symposium, 2005.
EGAAS 2005. European , vol., no., pp.589-592, 3-4 Oct. 2005
[23] Shih, S.E.; Duan, D.W.; Fordham, O.; Tornquist, K.; Zeng, X.; Chang-Chien, P.;
Tsai, R.; , "A W-Band 4-Bit Phase Shifter in Multilayer Scalable Array
Systems," Compound Semiconductor Integrated Circuit Symposium, 2007. CSIC 2007.
IEEE , vol., no., pp.1-4, 14-17 Oct. 2007
[24] Targonski, S.D.; Waterhouse, R.B.; Pozar, D.M.; , "Design of wide-band aperturestacked patch microstrip antennas," Antennas and Propagation, IEEE Transactions on ,
vol.46, no.9, pp.1245-1251, Sep 1998
[25] Targonski, S.D.; Waterhouse, R.B.; Pozar, D.M.; , "Design of wide-band aperturestacked patch microstrip antennas," Antennas and Propagation, IEEE Transactions on ,
vol.46, no.9, pp.1245-1251, Sep 1998
[26] Lamminen, A.; Saily, J.; Vimpari, A.R.; , "60-GHz Patch Antennas and Arrays on
LTCC With Embedded-Cavity Substrates," Antennas and Propagation, IEEE
Transactions on , vol.56, no.9, pp.2865-2874, Sept. 2008
[27] Shi-Gang Zhou; Tan-Huat Chio; , "Dual linear polarization patch antenna array with
high isolation and low cross-polarization," Antennas and Propagation (APSURSI), 2011
IEEE International Symposium on , vol., no., pp.588-590, 3-8 July 2011
[28] Duixian Liu; Akkermans, J.A.G.; Ho-Chung Chen; Floyd, B.; , "Packages With
Integrated 60-GHz Aperture-Coupled Patch Antennas," Antennas and Propagation, IEEE
Transactions on , vol.59, no.10, pp.3607-3616, Oct. 2011
[29] Kam, D. G.; Liu, D.; Natarajan, A.; Reynolds, S. K.; Floyd, B. A.; , "Organic
Packages With Embedded Phased-Array Antennas for 60-GHz Wireless
Chipsets," Components, Packaging and Manufacturing Technology, IEEE Transactions
on , vol.1, no.11, pp.1806-1814, Nov. 2011
152
[30] Smith, S.L.; Dyadyuk, V.; , "Measurement of the dielectric properties of Rogers
R/flex 3850 liquid crystalline polymer substrate in V and W band," Antennas and
Propagation Society International Symposium, 2005 IEEE , vol.4B, no., pp. 435- 438 vol.
4B, 3-8 July 2005
[31] Thompson, D.; Kirby, P.; Papapolymeron, J.; Tentzeris, M.M.; , " W-band
characterization of finite ground coplanar transmission lines on liquid crystal polymer
(LCP) substrates," Electronic Components and Technology Conference, 2003.
Proceedings. 53rd , vol., no., pp. 1652- 1655, May 27-30, 2003
[32] Pozar, D.M.; Targonski, S.D.; , "Improved coupling for aperture coupled microstrip
antennas," Electronics Letters , vol.27, no.13, pp.1129-1131, 20 June 1991
[33] Young-Mio Jo; , "Broad band patch antennas using a wedge-shaped air dielectric
substrate," Antennas and Propagation Society International Symposium, 1999. IEEE ,
vol.2, no., pp.932-935 vol.2, Aug 1999
[34] A.Elhawil, L. Zhang, J. Stiens, C. D. Tandt, N. A. Gotzen, G. V. Assche, and R.
Vounckx, "A quasi-optical free-space method for dielectric constant characterisation of
polymer materials in the mm-wave band," in Symposium of IEEE-LEOS-Benelux, 2007,
pp. 187 - 190.
[35] Yanting Wang; Chandrasekar, V.; , "Polarization isolation requirements for linear
dual-polarization weather Radar in simultaneous transmission mode of
operation," Geoscience and Remote Sensing, IEEE Transactions on , vol.44, no.8,
pp.2019-2028, Aug. 2006
[36] Sim, C.-Y.D.; Chun-Chuan Chang; Jeen-Sheen Row; , "Dual-Feed Dual-Polarized
Patch Antenna With Low Cross Polarization and High Isolation," Antennas and
Propagation, IEEE Transactions on , vol.57, no.10, pp.3321-3324, Oct. 2009
[37] Xiu-Yin Zhang; Quan Xue; Bin-Jie Hu; Sheng-Li Xie; , "A Wideband Antenna With
Dual Printed $L$-Probes for Cross-Polarization Suppression," Antennas and Wireless
Propagation Letters, IEEE , vol.5, no.1, pp.388-390, Dec. 2006
[38] Woelder, K.; Granholm, J.; , "Cross-polarization and sidelobe suppression in dual
linear polarization antenna arrays," Antennas and Propagation, IEEE Transactions on ,
vol.45, no.12, pp.1727-1740, Dec 1997
[39] Granholm, J.; Woelders, K.; , "Dual polarization stacked microstrip patch antenna
array with very low cross-polarization," Antennas and Propagation, IEEE Transactions
on , vol.49, no.10, pp.1393-1402, Oct 2001
153
Chapter 6
Conclusion and Future Work
This dissertation presented various antenna arrays, from the X- to W-band
implemented in LCP technology for beam forming applications. To maintain scalability,
what is needed is a material that light weight, low in density, has the ability to support
multi-layers with a vast assortment of interconnect options, and low in loss up to the
millimeter wave regime. This thesis has demonstrated the realization of this on LCP, to
provide lightweight structures for potentially large arrays.
Chapter 2 presents for the first time a phased array antenna receiver module on
LCP operating in the Ka-band. The 8 element phased array with integrated digital
controls and wideband antennas is targeted for the Ka-band (32-37 GHz) and achieves a
maximum measured scan angle of ±30o from boresight with a maximum beam squint of
6.75o. The total size of the mm-wave module is 100 mm x 100 mm with a board
thickness of 16 mils and a total weight of less than 12.3 grams. To the author’s best
knowledge, this work represents the most complex multilayer LCP module reported to
154
date, proving LCP to be an enabling technology for light weight, high density, large scale
phased array systems.
Chapter 3 presents a bidirectional antenna array for beamforming applications in
the X-band. A novel polygon wide slot antenna with a metamaterial inspired coupling
structure has been proposed and successfully implemented. A 16-element antenna array
utilizing the novel antenna is designed and characterized through simulation. The antenna
array has concurrent dual beams in the forward and backward directions. A beamwidth of
17o is measured for both the elevation and azimuth plane at 10 GHz. A maximum gain of
14.9 dBi is measured at 10 GHz showing an efficiency of greater than 65%. The 4x4
antenna array shows a measured fractional bandwidth of 25%.
Chapter 4 presents for the first time antenna arrays designed on LCP operating in
the W-band. Both a SIW array and a series patch array are implements on LCP utilizing
standard PCB manufacturing processes. Demonstrated also is a measurement method for
unbalanced millimeter wave antennas utilizing a diode detector circuit. Measurement
results show a beamwidth of approximately 13o in the E- and H-planes for both antenna
arrays. The maximum sensitivity of the SIW detector array is 121mV/mW at 96 GHz.
The maximum sensitivity of the series patch array detector is 250mV/mW at 95 GHz.
The total size of the antenna array with detector circuitry is 4 cm x 3 cm. It is
demonstrated that LCP can be utilized to extend the electrical performance of the next
generation millimeter wave phased array systems.
Chapter 5 presents the design and development of scalable W-band dual polarized
antennas for phased array applications. Simulation indicates the single antenna element
achieves a gain of greater than 7 dBi with an impedance bandwidth of greater than 21
155
GHz. A methodology for cross-polarization suppression in two dimensional array tiles
was presented. Finally fabricated prototypes using standard PCB manufacturing
processes were presented. Radiation pattern measurements were conducted at 80, 95, and
105 GHz for a single antenna element as well as an 8-element linear array. Measurements
were compared to simulation results and correlate well. Measurements indicate a
beamwidth of 90º and 13º for the single antenna element and the 8-element linear array
respectively.
This dissertation has demonstrated LCP as a viable material for future millimeterwave antenna needs. Future work would include the design of an integrated 2 x 2 W-band
phased array tile. This would include a thorough investigation of the interface between
the antenna substrate with a phased array MMIC and power distribution networks. We
are at an exciting time where advances continue in the millimeter to sub-millimeter wave
regime, which will enable technologies we have only dreamed of.
156
Appendix A Custom Verilog for FPGA
________________________________________________________________________
module SPI_Interface
(
CLOCK_50,
reset,
switch_input,
latch_enable,
LE1,
LE2,
LE3,
LE4,
LE5,
LE6,
LE7,
dout,
dout1,
dout2,
dout3,
dout4,
dout5,
dout6,
dout7,
out_clk,
state,
);
157
input reset;
input switch_input;
input CLOCK_50;
output latch_enable;
output LE1;
output LE2;
output LE3;
output LE4;
output LE5;
output LE6;
output LE7;
output dout;
output dout1;
output dout2;
output dout3;
output dout4;
output dout5;
output dout6;
output dout7;
output out_clk;
output state;
reg dout;
reg dout1;
reg dout2;
reg dout3;
reg dout4;
158
reg dout5;
reg dout6;
reg dout7;
reg latch_enable;
reg LE1;
reg LE2;
reg LE3;
reg LE4;
reg LE5;
reg LE6;
reg LE7;
reg [28:0] clock_count;
reg [3:0] state, c_state;
reg disable_clock;
assign out_clk = ~disable_clock & clock_count[3];
parameter RESET = 3'd0;
parameter STATE1 = 3'd1;
parameter STATE2 = 3'd2;
parameter STATE3 = 3'd3;
parameter STATE4 = 3'd4;
parameter STATE5 = 3'd5;
parameter STATE6 = 3'd6;
parameter STATE7 = 3'd7;
always @ (posedge CLOCK_50)
begin
159
clock_count = clock_count + 1'd1;
end
always @(reset or state or switch_input or clock_count)
begin
//c_state = state;
if(reset)
begin
c_state = RESET;
end
case(state)
STATE1: //Bit6
begin
disable_clock = 1'b0;
dout = 1'b0;
dout1 = 1'b0;
dout2 = 1'b0;
dout3 = 1'b0;
dout4 = 1'b0;
dout5 = 1'b0;
dout6 = 1'b0;
dout7 = 1'b0;
latch_enable = 1'b0;
LE1 = 1'b0;
LE2 = 1'b0;
LE3 = 1'b0;
LE4 = 1'b0;
LE5 = 1'b0;
LE6 = 1'b0;
LE7 = 1'b0;
160
c_state = STATE2;
end
STATE2:
begin
disable_clock = 1'b0;
dout = 1'b0;
dout1 = 1'b0;
dout2 = 1'b0;
dout3 = 1'b0;
dout4 = 1'b0;
dout5 = 1'b0;
dout6 = 1'b0;
dout7 = 1'b0;
latch_enable = 1'b0;
LE1 = 1'b0;
LE2 = 1'b0;
LE3 = 1'b0;
LE4 = 1'b0;
LE5 = 1'b0;
LE6 = 1'b0;
LE7 = 1'b0;
c_state = STATE3;
end
STATE3:
begin
disable_clock = 1'b0;
dout = 1'b0;
dout1 = 1'b0;
dout2 = 1'b0;
dout3 = 1'b0;
dout4 = 1'b0;
161
dout5 = 1'b0;
dout6 = 1'b0;
dout7 = 1'b0;
latch_enable = 1'b0;
LE1 = 1'b0;
LE2 = 1'b0;
LE3 = 1'b0;
LE4 = 1'b0;
LE5 = 1'b0;
LE6 = 1'b0;
LE7 = 1'b0;
c_state = STATE4;
end
STATE4:
begin
disable_clock = 1'b0;
dout = 1'b0;
dout1 = 1'b0;
dout2 = 1'b0;
dout3 = 1'b0;
dout4 = 1'b1;
dout5 = 1'b1;
dout6 = 1'b1;
dout7 = 1'b1;
latch_enable = 1'b0;
LE1 = 1'b0;
LE2 = 1'b0;
LE3 = 1'b0;
LE4 = 1'b0;
LE5 = 1'b0;
LE6 = 1'b0;
162
LE7 = 1'b0;
c_state = STATE5;
end
STATE5:
begin
disable_clock = 1'b0;
dout = 1'b0;
dout1 = 1'b0;
dout2 = 1'b0;
dout3 = 1'b0;
dout4 = 1'b1;
dout5 = 1'b1;
dout6 = 1'b1;
dout7 = 1'b1;
latch_enable = 1'b0; //6-bit
LE1 = 1'b0;
LE2 = 1'b0;
LE3 = 1'b0;
LE4 = 1'b0;
LE5 = 1'b0;
LE6 = 1'b0;
LE7 = 1'b0;
c_state = STATE6;
end
STATE6: //pad1
begin
disable_clock = 1'b0;
latch_enable = 1'b0;
LE1 = 1'b0;
LE2 = 1'b0;
LE3 = 1'b0;
163
LE4 = 1'b0;
LE5 = 1'b0;
LE6 = 1'b0;
LE7 = 1'b0;
dout = 1'b0;
dout1 = 1'b0;
dout2 = 1'b0;
dout3 = 1'b0;
dout4 = 1'b0;
dout5 = 1'b0;
dout6 = 1'b0;
dout7 = 1'b0;
c_state = STATE7;
end
STATE7: //pad2
begin
latch_enable = 1'b1;
LE1 = 1'b1;
LE2 = 1'b1;
LE3 = 1'b1;
LE4 = 1'b1;
LE5 = 1'b1;
LE6 = 1'b1;
LE7 = 1'b1;
disable_clock = 1'b1;
dout = 1'b0;
dout1 = 1'b0;
dout2 = 1'b0;
dout3 = 1'b0;
164
dout4 = 1'b0;
dout5 = 1'b0;
dout6 = 1'b0;
dout7 = 1'b0;
c_state = RESET;
end
RESET:
begin
//if(switch_input == 1'b0) begin
// c_state = STATE1;
//end
if (clock_count[28] == 1'b1) begin
c_state = STATE1;
end
else begin
c_state = RESET;
end
dout = 1'b0;
dout1 = 1'b0;
dout2 = 1'b0;
dout3 = 1'b0;
dout4 = 1'b0;
dout5 = 1'b0;
dout6 = 1'b0;
dout7 = 1'b0;
latch_enable = 1'b0;
LE1 = 1'b0;
LE2 = 1'b0;
LE3 = 1'b0;
LE4 = 1'b0;
LE5 = 1'b0;
165
LE6 = 1'b0;
LE7 = 1'b0;
disable_clock = 1'b0;
end
endcase
end
always @ (posedge clock_count[3])
begin
state <= c_state;
end
endmodule
Документ
Категория
Без категории
Просмотров
0
Размер файла
74 560 Кб
Теги
sdewsdweddes
1/--страниц
Пожаловаться на содержимое документа