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A steerable array antenna using controllable microwave dielectric slab phase shifters on a coplanar waveguide

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© Copyright 2006
Jun Ho Cha
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Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
A Steerable Array Antenna Using Controllable Microwave
Dielectric Slab Phase Shifters on a Coplanar Waveguide
Jun Ho Cha
A dissertation submitted in partial fulfillment o f the
requirements for the degree of
Doctor of Philosophy
University o f Washington
2006
Program Authorized to Offer Degree:
Department o f Electrical Engineering
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UMI N um ber: 3241886
Copyright 2006 by
Cha, Jun Ho
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University o f Washington
Abstract
A Steerable Array Antenna Using Controllable Microwave Dielectric Slab Phase
Shifters on a Coplanar Waveguide
Jun Ho Cha
Chair of the Supervisory Committee:
Professor Yasuo Kuga
Department o f Electrical Engineering
A simple steerable array antenna is designed and developed using a movable dielectric
phase shifter. The change of the effective dielectric constant at different dielectric slab
positions on a coplanar waveguide is used as the phase shifter. The impedance matching
and desired phase shift conditions are satisfied at two slab heights, and the reflection is
designed to be minimized at these slab positions. A low-loss dielectric material is used
as the dielectric slab and is placed close to a coplanar transmission line with an airgap.
Numerical simulations using Ansoft HFSS and Designer are conducted. In order to
verify the validity o f this phase shifter, we fabricate a CPW for testing purposes at 6
GHz. The 4x1 and 4x2 steerable array antennas at 5.8 GHz are fabricated and measured
for a wireless network. A 4x4 steerable array antenna with 3-bit dielectric slab phase
shifters is designed and fabricated at 20 GHz. Finally, the 8x4 and 8x7 steerable array
antennas with 4-bit dielectric slab phase shifters are designed and fabricated at 24 GHz.
The simulated and measured E- and H-plane radiation patterns of the proposed array
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antennas are illustrated. The H-plane radiation patterns are measured at different phase
shift positions and compared with the expected simulation results.
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TABLE OF CONTENTS
List of Figures...................................................................................................................... iii
List o f Tables........................................................................................................................ ix
Chapter 1. Introduction........................................................................................................1
Chapter 2. Phase Shifter Based on a Movable Dielectric Slab........................................... 6
2.1. Impedance Mismatch and Possible Solutions...................................................... 9
2.2. Approximate Analytic Formulas for a CPW ...................................................... 12
2.2.1. Without Dielectric Materials......................................................................12
2.2.2. With Dielectric Materials......................................................................... 15
2.3. Verification of a CPW Phase Shifter...................................................................19
Chapter 3. A Continuously Steerable Array Antenna Using Movable Dielectric
Slabs on a Coplanar waveguide...................................................................... 23
3.1. Continuous Phase Shifter Based on a Movable Dielectric Slab.....................23
3.2. Design of the Feed Network Using the Preset Delay Lines............................ 32
3.3. Verification o f a 4-Element Array Antenna...................................................... 34
3.4. Fabrication of a 4-Element Steerable Array Antenna at 5.8 GHz................... 39
Chapter 4. A 20 GHz Steerable Array Antenna Using 3-bit Dielectric Slab Phase
Shifters on a Coplanar Waveguide................................................................ 51
4.1. Phase Shifter Based on Movable 3-bit Dielectric Slab...........................
.53
4.1.1. Impedance Mismatch and Possible Solutions......................................55
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4.2. 4x4 Array Antenna with Phase Shifter at 20 GHz.............................................56
4.3. Simulated and Measured E- and H-Plane Radiation Patterns o f a 4x4
Array Antenna.....................................................................................................65
Chapter 5. A Steerable Phased-Array Antenna Using Mechanically Controllable
4-bit Dielectric Slab Phase Shifter on a Coplanar Waveguide................... ....73
5.1. Phase Shifter Based on Movable 4-bit Dielectric Slab................................... 75
5.2. 8x7 Tapered Array Antenna with Phase Shifter at 24 G Hz............................. 78
5.3. 8x4 Array Antenna with Phase Shifter at 24 GHz............................................93
5.4. A Mechanically Steerable Array Antenna Using Controllable
Dielectric Phase Shifters for 77 GHz Automotive Radar System.................. 103
Chapter 6 . Conclusions and Future Work.........................................................................113
BIBLIO G RAPHY.............................................................................................................. 116
Appendix A: Measurement Processing............................................................................. 124
Appendix B: HFSS and Designer Simulations................................................................ 127
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LIST OF FIGURES
Figure Number
Page
2.1 Phase shifter. A 3-D schematic of a ground-signal-ground (G-S-G) CPW with
a movable dielectric slab. The width o f the signal and ground traces are S and Gw ,
respectively. The gap between the ground and the signal is G and the substrate
thickness is h. The length of the dielectricmaterial is / ................................................ 8
2.2 Reflection and Transmission Coefficients o f aLayered Structure............................. 8
2.3 TL model o f a 4-bit phase shifter................................................................................ 10
2.4 Frontal view o f ground-signal-ground (G-S-G) coplanar waveguide with finite
lateral ground planes.....................................................................................................13
2.5 Coplanar Waveguide structure with dielectric materials, (a) frontal view o f a
ground-signal-ground (G-S-G) CPW when the dielectric material is attached to the
substrate, (b) frontal view of G-S-G CPW when the dielectric material is above the
substrate, in other words, the dielectric material (s rA) is not attached to the substrate
and is a distance o f 0 </*3<infmity from the substrate................................................. 14
2.6 Atop view o f the test circuit layout which includes the CPW and microstrip TL.
This circuit a 360 degree phase shift at d = 0 mm. The bottom layer o f the
CPW section does not have a ground plane and via is used for connecting
the ground..................................................................................................................... 18
2.7 Phase shifter without dielectric constant..................................................................... 21
2.8 Phase shifter with a dielectric constant....................................................................... 21
2.9 The measured phase with the 3-bit dielectric slab combinationsfor the desired
eight phase conditions(0°, -45 °, -90 °, -135 °, -180 °, -225 °, -270 °, and -315 °).......22
3.1 Block diagram of a 4-element steerable array antenna...............................................25
3.2 Block diagram o f a 4-element steerable array antenna...............................................26
3.3 Phase shifter. A 3-D schematic o f a ground-signal-ground (G-S-G) CPW with
a movable dielectric slab. The width o f the signal and ground traces are
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S (2 mm) and Gw (6 mm), respectively. The gap between the ground and
the signal is G (1 mm) and the substrate thickness is h (1 .6 mm).
The length o f the dielectric material is / ...................................................................... 27
3.4 Effective dielectric constant vs height d from the substrate........................................30
3.5 Simulated S21 and Sn results from d = 0 to d =5 mm.................................................31
3.6 Phase difference among 4 elements from -105° to +105°..........................................33
3.7 A schematic o f a 4-element patch antenna.................................................................. 35
3.8 Simulated radiation patterns o f a 4-element array antenna.........................................35
3.9 Fabrication o f the 4x1 array antenna............................................................................36
3.10 4x1 measured H-Plane radiation patterns................................................................. 36
3.11 Fabrication of the 4x2 array antenna..........................................................................37
3.12 4x2 measured H-Plane radiation patterns................................................................. 37
3.13 Fabrication o f the 4x4 array antenna..........................................................................38
3.14 4x4 measured H-Plane radiation patterns................................................................. 38
3.15 Photo o f a 4x1 array antenna without dielectric slabs...............................................41
3.16 Photo o f a 4x1 array antenna with dielectric slabs....................................................42
3.17 Photo o f a 4x2 array antenna without dielectric slabs...............................................43
3.18 Photo o f a 4x2 array antenna with dielectric slabs....................................................44
3.19 Radiation patterns o f a 4x1 array antenna when d = 5 mm at 5.8 GHz.
(a) Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern...45
3.20 Radiation patterns o f a 4x1 array antenna when d = 0 at 5.8 GHz.
(a) Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern...46
3.21 Radiation patterns o f a 4x1 array antenna when d = 0.1 mm at 5.8 GHz.
(a) Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern...47
3.22 Radiation patterns o f a 4x2 array antenna when <7=5 mm at 5.8 GHz.
(a) Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern...48
3.23 Radiation patterns o f a 4x2 array antenna when <7= 0.1 mm at 5.8 GHz.
(a) Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern...49
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3.24 Radiation patterns of a 4x2 array antenna when d = 0 at 5.8 GHz.
(a) Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern...50
4.1 Block diagram o f a 4x4 steerable array antenna....................................................... 52
4.2. Phase Shifter. A-3D schematic of a ground-signal-ground (G-S-G) CPW with a
movable dielectric slab. The width o f the signal and ground traces are S (0.8 mm)
and Gw (2.5 mm), respectively. The gap between the ground and the signal is
G (0.4 mm) and the substrate thickness is h (0.508 mm). The height o f the
dielectric material is 2.5 mm and its length is / ........................................................... 54
4.3 TL model o f a 3-bit phase shifter.................................................................................54
4.4 Fabrication o f the 4x2 array antenna at 20 G Hz......................................................... 57
4.5 4x2 Measured radiation patterns (a) H-plane radiation pattern
(b) E-plane radiation pattern......................................................................................58
4.6 Fabrication o f the 4x4 array antenna at 20 G Hz......................................................... 59
4.7 4x2 Measured radiation patterns (a) H-plane radiation pattern
(b) E-plane radiation pattern......................................................................................60
4.8 Photo o f a 4x4 array antenna without dielectric slabs................................................63
4.9 Dielectric slab positions on a CPW for the 3-bit phase shifter.................................. 64
4.10 Measured Sn of a 4x4 array antenna without slabs as shown in Figure 4.8............64
4.11 Simulated and measured H-plane Radiation Patterns: No phase shift case.
Slab positions are (000,000,000,000)........................................................................ 67
4.12 Simulated and measured E-plane Radiation Patterns: No phase shift case.
Slab positions are (000,000,000,000).........................................................................68
4.13 Simulated and measured H-plane Radiation Patterns.
The phase shift among elements is -45° which corresponds to the beam
angle of 15°. Slab positions are (110,010,100,000)..................................................69
4.14 Simulated and measured H-plane Radiation Patterns.
The phase shift among elements is -90° which corresponds to the beam
angle o f 30°. Slab positions are (011,001,010,000)..................................................70
4.15 Simulated and measured H-plane Radiation Patterns.
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The phase shift among elements is -135° which corresponds to the beam
angle o f 135°. Slab positions are (000,110,011,100)................................................71
4.16 Actuators for a 3-bit phase shifter. Each rod actuates 4 slabs with the
same phase shift and all rods are synchronized. One bit pattern on the
top section shows the (110,010,100,000) case.......................................................... 72
4.17 Conceptual view o f a 2-D satellite tracking antenna based on our tunable design.
CPW and feeding network are on the bottom side o f PCB.......................................72
5.1 Figure 5.1 Block diagram of an 8x7 steerable array antenna.
(a) Front Feeding (b) Back Feeding.............................................................................74
5.2 Phase shifter. A 3-D schematic o f a ground-signal-ground (G-S-G) CPW with
a movable dielectric slab. The width o f the signal and ground traces are
S (0.8 mm) and Gw (4.6 mm), respectively. The gap between the ground and
the signal is G (0.4 mm) and the substrate thickness is h (0.508 mm).
The length o f the dielectric material is 1.......................................................................76
5.3 TL model o f a 4-bit phase shifter.................................................................................76
5.4 (a) Top view o f an 8x7 array antenna without dielectric slabs.
(b) Bottom view o f an 8x7 array antenna without dielectric slabs............................ 82
5.5 Dielectric slab positions on a CPW for the 4-bit phase shifter.................................. 83
5.6 Measured Sn o f an 8x7 array antenna without slabs as shown in Figure 5.4..........83
5.7 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna: No phase shift case. Slab positions are (000,000,000,000,000,000,
000,000).................................................................................... 86
5.8 Simulated and measured E-plane Radiation Patterns o f an 8x7 tapered array
antenna: No phase shift case. Slab positions are (000,000,000,000,000,000,
000,000)........................................................................................................................ 87
5.9 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna : The phase shift among elements is -22.5° which corresponds to the beam
angle o f 7.5°. Slab positions are (1110,0110,1010,0010,1100,0100,1000,0000).......88
5.10 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna: The phase shift among elements is -45° which corresponds to the beam
angle o f 15°. Slab positions are (0111,0011,0101,0001,0110,0010,0100,0000)...... 89
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5.11 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna: The phase shift among elements is -90° which corresponds to the beam
angle of 30°. Slab positions are (0011,0001,0010,0000,0011,0001,0010,0000)...... 90
5.12 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna : The phase shift among elements is -135° which corresponds to the beam
angle o f 45°. Slab positions are (0101,0010,0111,0001,0100,0011,0110,0000)...... 91
5.13 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna: The phase shift among elements is -157.5° which corresponds to the beam
angle o f 52.5°. Slab positions are (1000,0101,1100,0011,1010,0111,1110,0000). ...92
5.14 (a) Top view o f a 8x4 array antenna without dielectric slabs.
(b) Bottom view o f a 8x4 array antenna without dielectric labs............................ 95
5.15 Simulated and measured H-plane Radiation Patterns of a 8x4 array antenna: No
phase shift case. Slab positions are (0000,000,0000,0000,0000,0000,0000,0000)..96
5.16 Simulated and measured E-plane Radiation Patterns of a 8x4 array antenna: No
phase shift case. Slab positions are (0000,000,0000,0000,0000,0000,0000,0000)..97
5.17 Simulated and measured H-plane Radiation Patterns of a 8x4 array antenna:
The phase shift among elements is -22.5° which corresponds to the beam angle
o f 7.5°. Slab positions are (1110, 0110,1010,0010,1100,0100,1000, 0000).......98
5.18 Simulated and measured H-plane Radiation Patterns of a 8x4 array antenna:
The phase shift among elements is -45° which corresponds to the beam angle
o f 15°. Slab positions are (0111,0011,0101, 0001, 0110, 0010, 0100, 0000).......99
5.19 Simulated and measured H-plane Radiation Patterns of a 8x4 array antenna:
The phase shift among elements is -90° which corresponds to the beam angle
of 30°. Slab positions are (0011, 0001, 0010, 0000, 0011, 0001, 0010, 0000)......100
5.20 Simulated and measured H-plane Radiation Patterns of a 8x4 array antenna:
The phase shift among elements is -135° which corresponds to the beam angle
o f 45°. Slab positions are (0101, 0010, 0111, 0001, 0100, 0011, 0110, 0000)......101
5.21 Simulated and measured H-plane Radiation Patterns of a 8x4 array antenna:
The phase shift among elements is -157.5° which corresponds to the beam angle
o f 52.5°. Slab positions are (1000,0101,1100, 0011,1010, 0111,1110, 0000)....102
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5.22 (a) Block diagram of a 3x8 transmitting array antenna and
(b) a 7x8 receiving array antenna...........................................................................104
5.23 Utilization of dielectric slabs on a CPW ..................................................................106
5.24 Layout o f the whole antenna. (a)3x8 transmitting array antenna
(b)5x8 receiving array antenna (c)7x8 receiving array antenna............................ 107
5.25 Radiation patterns o f a 3x8 array antenna, (a) Radiation patterns occurring
when there are no delays lines and no phase shifters, (b) Radiation patterns
occurring when the dielectric slab (180°) is only inserted into the left CPW.
(c) Radiation patterns occurring when the dielectric slab (180°) is only
inserted into the right CPW ...................................................................................... 109
5.26 Radiation patterns o f a 5x8 array antenna, (a) Radiation patterns occurring
when there are no delays lines and no phase shifters, (b) Radiation patterns
occurring when the dielectric slab (360°and 720°) is only inserted into
the left CPW. (c) Radiation patterns occurring when the dielectric slab
(360° and 720°) is only inserted into the right CPW............................................... 110
5.27 Radiation patterns o f a 7x8 array antenna, (a) Radiation patterns occurring
when there are no delays lines and no phase shifters, (b) Radiation patterns
occurring when the dielectric slab (360°, 720° and 1080°) is only inserted into
the left CPW (c) Radiation patterns occurring when the dielectric slab
(360°, 720° and 1080°) is only inserted into the right CPW......................................I l l
5.28 Combined radiation patterns, (a) Combined radiation patterns with a 3x8
transmitting array antenna (Figure 5.25(b)) and a 5x8 receiving array antenna
(Figure 5.26(b)). (b) Combined radiation patterns with a 3x8 transmitting array
antenna (Figure 5.25(c)) and a 5x8 receiving antenna (Figure 5.26 (c)).
(c) Combined radiation patterns with a 3x8 transmitting array antenna (Figure
5.25(b)) and a 7x8 receiving array antenna (Figure 5.27(b)). (d) Combined
radiation patterns with a 3x8 transmitting array antenna (Figure 5.25(c))
and a 7x8 receiving array antenna (Figure 5.27(c))................................................ 112
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LIST OF TABLES
Table Number
Page
Table 2.1 Comparison with approximate analytic formulas and numerical results.........18
Table 3.1 Simulated results at d = 5 mm, d = 1.25 mm, and d = 0 .................................. 29
Table 3.2 Phase relationship o f a 4-element array at 3 phase (slab) positions:
-105°, 0, and+105°............................................................................
32
Table 4.1 Simulation results withe,. = 3.73 slab (slab length = 4.48 mm)....................... 62
Table 4.2 Simulation results withe, = 10.2 slab (slab length= 3.26 mm)........................ 62
Table 5.1 Simulation results w ithe, = 2.94 slab (slab length= 4.74 mm)...................... 79
Table 5.2 Simulation results w ithe, = 3.73 slab (slab length = 4.06 mm)....................... 79
Table 5.3 Simulation results withe, = 10.2 slab (slab length=2.71 mm)......................... 80
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ACKNOWLEDGEMENTS
First of all, I wish to express my gratitude to my academic advisor, Professor Yasuo
Kuga, for providing me with many worthwhile research ideas to pursue, for his sage
advice, and financial support during my doctoral program. He has always been my
inspiration as to how to grow and build myself as a researcher. His calm approach to
solving technical problems always inspires me to think critically and take another fresh
look at the problems I encounter. I am honored to have had the opportunity to not only
work with him as my advisor, but also to have him as a mentor. Professor Kuga has
always impressed me with his understanding and willingness to help students with their
problems. W ith his spirit, he forms a standard that I aspire to in my own life. I am also
very thankful to Professor Akira Ishimaru, Professor Minoru Taya, Professor Jenq-Neng
Hwang, and Professor Sermsak Jaruwatanadilok for serving as my committee members.
And I am thankful to Mrs. Noel Henry, Program Manager, for her gracious assistance
and generous guidance in all aspects of this journey. I am also very grateful for the
cooperation and motivation of my colleagues, as I can’t imagine working with a better
team. For me, the members of the Electromagnetics and Remote Sensing Laboratory
(ERSL) are like family. I appreciate their friendship, understanding, and care. A ll the
members of ERSL come from different parts o f the world, but there exists a friendship
and camaraderie among them that is quite rich. I thank Dr. John R. Thomas whose
continuous encouragement, valuable discussions and unlimited friendship have been
invaluable. Finally, I would like to acknowledge the continued encouragement and
support I have received from my mother, father-in-law, mother-in-law, and especially
from my wife, Hyunkyung Park.
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DEDICATION
To my father:
Seok il Cha, 1928-1983, B.A., Konkuk University, Seoul, Korea.
Requiescat in peace.
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1
Chapter 1
Introduction
Global communication systems play a significant role in commercial companies
and in the military. Existing satellite or airborne communications systems have some
serious limitations for global communications. Due to its practical applications in
diverse areas, a significant amount of research has been devoted to investigate this
problem. A low-cost steerable antenna is one of the missing links o f the future flexible
wireless communication systems. For example, the most flexible
satellite to
ground/airplane communication systems is based on electronic phased-array. antenna
technology [1],[2].
However, the cost o f a phased-array antenna is related to the
number of active elements, and thus the present systems are often too expensive for
many commercial and military applications. The antenna beam steering can also be done
by mechanically moving the reflector [3]-[6 ]. Although the mechanically steerable
antennas can be inexpensive, current antennas which use the electro-mechanical actuator
are usually bulky and prone to mechanical failure.
M y research has focused on the design and development o f a low-cost phasedarray antenna. This newly proposed phase shifter design is anticipated to be easy to
fabricate and w ill greatly reduce cost. Reducing the fabrication costs o f this antenna
opens possibilities for many exciting applications. It w ill, for instance, allow the antenna
to be suitable for Low Earth Orbit (LEO) satellite systems, W LAN ( Wireless Local
Area Network), automobile collision avoidance radar, and so on. These are just
applications o f many that could benefit from a low cost phased array antenna.
Many antenna system applications require that the main beam pointing direction
be changed with time, or scanned. A phased array is defined as an array antenna whose
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2
main beam maximum direction is controlled by varying the phase or time delay to the
elements. Array technology is moving toward the integration of transmit/receive
electronics and associated controllers. The antenna is then a subsystem rather than a
separate device. The term smart antenna has been coined for antennas that include
control functions such as beam scanning. Smart antennas are finding wide use in several
commercial and military applications [7]. A unique advantage offered by phased arrays
is the ability to form multiple main beams pointing in different directions simultaneously.
Why develop a phased array antenna? Contrary to dish or slotted array antennas,
which use physical shape and direction to form and steer the beam, phased array
antennas utilize the interference between multiple radiating elements to achieve beam
forming and beam steering [8 ]. By electronically adjusting the signal, each element
radiates and the combined radiation pattern can be scanned and shaped at high speeds.
This gives phased array antennas some unique advanced capabilities. Some advantages
o f phased array antennas (electronic scanning) over dish or slotted array antennas
(mechanical scanning) include the extreme beam agility, the reduction o f the antenna
radar cross section, and the advanced beam forming capabilities. The phased array
antennas have broad applicability for both commercial and military applications,
including military radars, cellular base stations, satellite communications, automotive
anti-collision radar, and ultrasound in medicine.
The cost o f a phased-array antenna is related to the number o f active elements,
and thus the present systems are often too expensive for many commercial and military
applications. Phased array per element costs is currently in the range of $50 to $150 at
20GHz and $80 to $250 at 44GHz. One goal for 5 years from now is to create a price
point o f $10 in at 20GHz and in 7 years, $20 at 44GHz. Phase shifters are also critical
element for electronically scanned phased array antennas, and typically account for a
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significant amount of the cost o f producing and antenna array. Phase shifters are the
devices in an electronically scanned array that allow the antenna beam to be steered in
the desired direction without physically re-positioning the antenna. It is not uncommon
for the cost o f the phase shifters to represent nearly half o f the cost o f the entire
electronically scanned array. This excessive cost has limited the deployment o f
electronically scanned antennas and restricted their use to expensive and specialized
systems, such as fighter aircraft radar and commercial systems such as cellular
telephone base stations. The proposed phase shifters in this dissertation have the
potential to reduce fabrication cost. One primary method that lowers cost is using an
inexpensive movable dielectric slab placed on a CPW (Coplanar waveguide), which can
be used as a phase shifter. In addition, patch antennas are fairly inexpensive, easy to
manufacture, and extremely low profile.
In my dissertation, a mechanically steerable antenna using a movable dielectric
phase shifter is designed and developed. We demonstrate that a movable dielectric slab
placed close to a CPW can be used as a phase shifter. The effective dielectric constant
and the characteristic impedance can be calculated as a function of slab height. The
impedance matching and desired phase shift conditions are satisfied at two slab
positions. The best dielectric material seem to be in the rage o f sr =3 to 10 in our case.
Because we are introducing the impedance mismatch section on TL, the elimination of
reflection is essential to design a phase shifter, and possible solutions for the impedance
mismatch problems are explored [9]. To minimize the reflection caused by the dielectric
slab and also obtain the desired phase shifter, the dielectric constant of the slab has to be
set to a specific value.
Many researchers have focused on developing a phase shifter [10], [11].
Recently, the different types o f microwave phase shifters including MEMS-based and
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ferroelectric-based, have been proposed for antenna applications [12]-[15]. In addition,
a phase shifter using a piezoelectric transducer controlled dielectric layer to perturb the
electromagnetic fields o f a CPW has been demonstrated [16], [17]. The proposed phase
shifter is to minimize the reflection with or without (ON-OFF position) at the designed
frequency. In addition, the proposed phase shifter is fairly inexpensive, easy to
manufacture, and extremely low profile.
In Chapter 2, we introduce a new concept o f a phase shifter using dielectric
slabs on a CPW with airgaps. We investigate two ideas o f phase shifters. The first idea is
that the dielectric slab continuously moves onto a CPW from d - 0 to d = 5 mm. The
second idea is that the dielectric slab on a CPW is binary states (d = 0 or d = 2 mm). The
performance o f the CPW-based phase shifter is fabricated and tested at 6 GHz.
In Chapter 3, a simple continuously steerable array antenna is designed and
fabricated using a movable dielectric phase shifter. The proposed array antennas consist
o f a patch array antenna, phase shifter, and feed network with a preset delay lines. The
effective dielectric constant and the characteristic impedance are calculated as a function
o f slab height. We compare the simulated radiation patterns and the measured radiation
patterns of the 4x1 and 4x2 steerable array antenna at 5.8 GHz.
In Chapter 4, a 20 GHz steerable array antenna using 3-bit dielectric slab phase
shifters on a CPW is designed and fabricated. The change of effective dielectric constant
at different dielectric slab positions on a CPW is used as the phase shifter. We use a 3-bit
phase shifter give by 45°, 90°, and 180° sections. Then the available phase shifts w ill be
eight states (0°, 45°, 90°, 135°, 180°, 225°, 210°, and 315°). The H-plane radiation
patterns o f the 4x4 steerable array antenna are measured at different phase shift positions
and compared with the expected results. By using a 3-bit phase shifter, the antenna beam
angle can be scanned from -45° to + 45 °.
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5
In Chapter 5, we have improved the steerable array antenna shown in Chapter
4. This array antenna is operating at 24 GHz. We want a 4-bit phase shifter given by
22.5 °, 45 °, 90 0 and 180 0 sections. Then the available phase shifts w ill be sixteen states
(0 °, 22.5°, 45°, 67.5°, 90°, 112.5°, 135°, 157.5°, 180°, 202.5°, 225°, 247.5°, 270°,
292.5 °, 315 °, and 337.5 °). In order to reduce the E-plane radiation patterns, we have
designed and fabricated the 8x7 tapered array antenna and placed both the CPW and the
feeding network section on the bottom side. The H-plane radiation patterns o f the 8x4
and 8x7 steerable array antenna are measured at different phase shift positions and
compared with the expected results. By using a 4-bit phase shifter, the antenna beam
angle can be scanned from -52.5° to + 52.5 °. In addition, we present the steerable array
antenna at 77 GHz. As the dielectric slab is inserted into the CPW in an alternating and
repetitive fashion, the beam angle is able to scan from - 2 0 ° to + 2 0 °.
In Chapter 6 , we conclude the dissertation and propose the future work.
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6
Chapter 2
Phase Shifter Based on a Movable Dielectric Slab
The phase shifters are a critical element for electronically scanned phased-array
antennas, and typically account for a significant amount o f the cost o f producing an
antenna array [18], [19]. The reduction o f fabrication cost opens possibilities for many
applications. Recently, the different types o f microwave phase shifters including
MEMS-based and ferroelectric-based, have been proposed for antenna applications. In
addition, a phase shifter using a PZT controlled dielectric layer to perturb the
electromagnetic fields of a CPW has been demonstrated. In this dissertation, we
introduce that a movable dielectric slab which is placed close to a coplanar waveguide
(CPW) with airgap can be used as a phase shifter. We minimize the refection with or
without dielectric material (On-OFF position) at the designed frequency. The proposed
array antenna consists o f the series-fed patch antennas, phase shifters, and the feeding
network as shown in Figure 2.1. It is noted that the most difficult part of our design,
simulation and fabrication processes is the phase shifter [20]. The desired phase shift
may not be achievable with all transmission line (TL) types. To obtain the optimum
combinations o f the TL structure and dielectric material, we have investigated several
TL structures. These include a microstrip TL, CPW TL without airgaps, and CPW TL
with airgaps. M y study shows that the microstrip TL and CPW TL without airgaps are
insensitive to the presence o f a dielectric slab on them and are eliminated from
consideration. In addition, it is found that a high- er material such as BaTiC>3 is not suited
for a phase shifter. It appears that the electromagnetic field perturbation caused by a
high- sr material is significant, and the structure does not work as a phase shifter. The
best dielectric materials seem to be in the range o f er=3 to 10 in our case. Because we
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7
are introducing the impedance mismatch section on TL, the elimination o f reflection is
essential to design a phase shifter, and possible solutions for the impedance mismatch
problems are explored [21]. To minimize the reflection caused by the dielectric slab and
also obtain the desired phase shifter, the dielectric constant of the slab has to be set to a
specific value. The performance o f the CPW-based phase shifter was tested at 6 GHz.
The basic concept of the phase shifter is illustrated in Figure 2.1. The CPW has
airgaps between the center signal line and ground lines. As the movable dielectric slab
moves closer or into the gap o f the CPW, the effective dielectric constant changes and is
given as a function of d for the given structure. In this dissertation, we assume that the
slab can be either attached on the substrate ( d = 0 or very small) or far away from the
substrate (d = oo or d > 5 mm in our case).
The effective dielectric constant can be calculated using the transmission
coefficient (7) o f a layered structure where T i is the reflection coefficient at the
boundary and 6 is the phase shift due to a change in the effective dielectric constant.
The transmission coefficient (7) o f a layered structure is illustrated in Figure 2.2. In
principle, it should be able to obtain the characteristic impedance Zs and seffective. This
structure can be modeled as an unmatched TL section (or a layered structure). Assuming
that the reflection is small (r ^ O .l), the transmission coefficient 7’ o f a layered structure
is given as [22], [23]
1 2/, =
* _^
= q-rfK
\
i - r 3r 2e~2je \ - r y 2je
T = W ne- * ,
_
(2 . 1)
where Tj is the reflection coefficient at the boundary due to a semi-infinite layer and
9 is the phase shift due to a slab. When 6 becomes mn where m is an integer, the
reflection from the slab diminishes (or |7] becomes 1). This is the impedance matching
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
■r ir >
Figure 2.1 Phase shifter. A 3-D schematic of a ground-signal-ground (G-S-G) CPW with
a movable dielectric slab. The width o f the signal and ground traces are S and Gw,
respectively. The gap between the ground and the signal is G and the substrate thickness
is h. The length o f the dielectric material is I.
r ,=
Zs
—
Z0
Zs + Z 0
Z0 Zs
Z 0+ Z s
T21
w
—
r2=
Tn
= -r>
w
M—
Tn
z0 zs = r 2 = - r 1
—
r 3=
Z 0+ Zs
r,
r 2
^
Zo
t 2 1 = i + r\
t ;2 = i + r 2 = i - r ,
e~J0 : delay
Figure 2.2 Reflection and Transmission Coefficients o f a Layered Structure.
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9
condition which can also be stated in terms o f the effective slab length of mX/2. The
slab length must satisfy this condition to minimize reflection. In addition, when d is
changed from d = oo to d = 0 , the phase shift o f the slab section w ill be related to the
effective dielectric constant of these two states [24]. Therefore, if we want to create a
desired phase shift without creating undesired reflection, we need to satisfy specific
conditions for each slab.
The phase change at the slab height d is with respect to that without a dielectric
slab ( d = oo), and it can be expressed as
(2.2)
where £ 0 is wavenumber in free space, Ld is the slab length, seff d=00 represents the
effective dielectric constant when the dielectric material is far enough away from the
substrate, and seff d represents the effective dielectric constant at the slab height d.
The phase is related to the operating frequency ( A0), CPW length, and effective
dielectric constant. Since the CPW length is fixed, the only effective dielectric constant
as a function of the dielectric slab position can be changed.
2-1. Impedance Mismatch and Possible Solutions
I f we change the slab height continuously, we can adjust the phase shift.
Unfortunately, this also changes the characteristic impedance of the CPW section and
introduces reflection. Despite this, we can eliminate the impedance mismatch problem
by using two positions (d = 0 and d = oo). To minimize reflection when the material is
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10
/.
o
o
o
o
o
o
o
o
o
o
Figure 2.3 TL model o f a 4-bit phase shifter.
added to TL, we can set the length o f the modified section to be X/2 (or mk/2 where m
is an integer). Suppose we want a 4-bit phase shifter given by 22.5 °, 45 °, 90° and 180°
sections as shown in Figure 2.3. Then the available phase shifts w ill be sixteen states (0°,
315 °, and 337.5 °). The fixed CPW section is denoted by Zo and nhwithout dielectric
slabs. The sections h, I2, h, and U w ill have dielectric slabs, and are given by (Z i, «i) for
the jr 18 section, (Z 2, ni) for the n/A section, (Z 3, ni) for the n j 2 , and (Z 4, n$) for the
n section where Z and n are the characteristic impedance and the index o f refraction of
each section. We hope to create conditions with no reflection for all states. We also aim
to obtain the lengths l\, h, h, and U as well as the index of refraction n\, ni, «3, and «4 in
terms of «b- This can be done by satisfying the following conditions:
1) For n 18 (22.5 °, bit 0), we need to satisfy
(«j
= n/% and nxk0ll = m l7z
where m\ is an integer. By eliminating /,, n\ is given by
(2.3)
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2) For njA section (45 °, bit 1), we need to satisfy
(n2 - nb) kQl2 = n/A and n2k0J2 = m2n
where mi is an integer. By eliminating l2, ni is given by
n, = nh
2
4m,
—
4m2 -1
I f we choose m, = 1, then
4
»2 =T«a
3
, ,
m2n 3 An
and l2 = i = o
h2£0
8 nb
3) For ;r/2 section (90 °, bit 2), we need to satisfy
( n3 ~ nb ) K h
=7t! 2 and «3^0/3 = m 2n
where m3 is an integer. By eliminating /3, « 3 is given by
2 m,
2 m3 - 1
For m3 = 1:
A
= 2nb and /3 = ——
4nb
_
_
4
, ,
3 An
Form3= 2: n3 = —nb and /3 = — 3
4 «A
4) For
71 section
(180 °, bit 3), we need to satisfy
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12
( « 4 - nb) k0l4 = n and n4k0l4 = m47c
where m\ is an integer. By eliminating /4, n\ is given by
(2 .10)
(2 .11)
(2 . 12)
2-2. Approximate Analytic Formulas fo r a CPW
2-2-1. Without Dielectric Materials
Coplanar waveguides are used for transmission lines where all the conductors
are on the same plane; namely, on the top surface o f the dielectric substrate. The
coplanar waveguide is initially proposed by Wen [25]-[28] in 1969. Figure 2.4 illustrates
a CPW with finite dielectric thickness and finite width ground planes [29].
Next we need to derive a method to obtain the effective dielectric constant of
CPW for a given d. The effective dielectric constant o f a simple CPW structure has been
studied extensively [30]. However, we are unable to find an existing analytical solution
for a CPW structure shown in Figure 2.1, and we have to rely on the numerical
technique. A fullwave analysis of a CPW has been reported using the Galerkin’s method
in the spectral domain [31]. Analytical formulas for the quasi-TEM parameters are
obtained from either exact or approximate conformal mapping techniques. A general
CPW with finite width ground planes o f the effective dielectric constant and the
characteristic impedance are given by [32]
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13
Figure 2.4 Fontal view o f ground-signal-ground (G-S-G) coplanar waveguide with finite
lateral ground planes
z _ 30n K'(k)
=
(2.13)
^
(2.14)
eeff = l + q{er - \ )
where q is called the filling factor
_ 1 Kjk^K 'jk)
q=
(2.15)
2 K'(kx) K{k)
k = a \ l - b 2/ c 2
(2.16)
b ] j l - a 2/ c 2
_ sinh{na/2h) 11 - sinh2 {nb / 2 h ) /sinh2 {nc / 2 h)
K = -
s in h (^ /2 /z) y 1 - sinh2 (na / 2 h)/ sinh2 {nc / 2 h)
*(*)_ 1
AT-fr)
K {k)
* '( * )
f
1+ V F
(2.17)
for 0.707 < £ < 1
(2.18)
for 0 < £ <0.707
(2.19)
* T l-V *
- In
;r
\ + 4k
\-4 k
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14
€„a - a i r
IV
6576
m
b
mi
s, = a ir
(a)
(b )
Figure 2.5 Coplanar Waveguide structure with dielectric materials, (a) frontal view o f a
ground-signal-ground (G-S-G) CPW when the dielectric material is attached to the
substrate, (b) frontal view o f G-S-G CPW when the dielectric material is above the
substrate, in other words, the dielectric material ( er4) is not attached to the substrate and
is a distance o f 0 </j3<infinity from the substrate.
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15
I f the parameters, a, b, c, h, and sr are given, the effective dielectric constant and the
characteristic impedance can be calculated.
2-2-2. With Dielectric Materials
Figure 2.5(a) shows the dielectric slab attached to the substrate. For this structure
(Figure 2.5(a) the effective dielectric constant can be expressed as [33]
£ eff
= Q \ £ r\ + ^ 2 £ r2
+ < J l £ r3
+^ 4 £ r4
(2 .20)
where qi,q2,q3, and q4 describe the filling factor for the various dielectric regions.
These were obtained by conformal mapping techniques and can be written, as follows:
C ; = 2s0
K(k.)
(i = I, II, III, and IV )
(2.21)
where
ki = T
(2 .2 2 a)
s in h fa /2 /iJ
~ sinh(«&/2/i2)
(2'22b)
sinh(aa/2ft,)
“ skih(xb/2h,)
tm h ( r n n h )
(Z22C)
^
Let us now determine the filling factors for the configuration in Figure 1.2 (a). In this
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16
case,
(2.23)
°K (k,)
Here C a is the capacitance o f the CPW structure with air as dielectric.
The filling factor for the second region is
C
^IIa
(2.24)
with
K (k„)
c a = 'I f.
(2.25)
K \K )
Similarly,
Ca
q,=—
3 ca
(2.26)
with
r 'a _ 9 c
'
'4 )
)
(2.27)
for the region IV, the filling factor is defined as
C a/ 2 - C j U
1
C
'- 'in
ca
(2.28)
Similarly, for the region I
Ca1 2 - Ca
u
1
q, = ------------- — = —
Ca
2
ca
ca
(2.29)
It can be verified that
=1
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(2.30)
17
The characteristic impedance can be determined from seff and C a
Z
°
=
<2'31)
^
where c is the speed o f light.
Similarly, the effective dielectric constant and the characteristic impedance o f the Figure
2.5 (b) structure can be expressed as
£ e ff
=
<h£ rl
+ < l2 £ r 2 +
c - = 2*o
<h£ *
+
+ ^ 5 £ r5
(2-32)
d = I, II, HI, IV , and V )
(2.33)
where
* /= 7
b
(2.34a)
sinh(^-a/2 ^ )
77
s in h ( ^ / 2 / 22)
(
_ s in h (^ / 2 /?3)
777 _ s in h (^ / 2 A3)
(234C)
_ sinh(^r/2 /?4)
7F 's in h (,ri> /2 /,4)
(2‘34d)
_ tanh(^a / 2 /*5)
~ tanh(;rf> / 2h5)
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(2,34e)
}
18
TABLE 2.1
Comparison with approximate analytic formulas and numerical results
Analytic Formulas
Parameters
Characteristic
impedance ( Q )
Numerical Results
(HFSS)
d=5vam
<7=0mm
if=5mm
<7=0mm
45.37
31.9
45.37
31.9
2.59
5.31
2.59
5.31
Effective dielectric
constant ( £eff)
Figure 2.6 A top view of the test circuit layout which includes the CPW and microstrip
TL. This circuit a 360 degree phase shift at d = 0 mm. The bottom layer o f the CPW
section does not have a ground plane and via is used for connecting the ground.
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19
Similarly, the filling factor can be expressed as
1
c°u
(2.35a)
q' = 2 ~ a
c IaT
ft-p -
(2.35b)
ca
ca
c IV
a
q3 = -JIL
= C°
aj^
_ U IL
5
(2.35c)
(2.35d)
s~ia
%^in
s~*a
i
JV _ 1
s-%a
'-'ffl
sia
^ IV
2
Ca
Ca
ca
(2.35e)
It can be verified that
0i + 0 2 + 0 3 + 0 4 =1
To check the accuracy o f this formula and also the correctness of our simulations, we
compared several cases with Ansoft HFSS simulations. Table 2.1 shows the comparison
o f the approximate analytic formulas given in Equations (2.20) and (2.31) with the data
obtained from HFSS numerical simulations. The agreement is reasonable and our
simulation seems to be working. A ll the following results are obtained using HFSS
simulations.
2-3. Verification o f a CPW Phase Shifter
In order to verify the validity o f this phase shifter, we fabricated the CPW on a
Duroid 5880 (e sub = 2.2) and measured eight different states. Duroid 5880 was chosen
because the dielectric constant is accurately known. Figures 2.7 and 2.8 show the test
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20
phase shifter circuits designed at 6 GHz with and without the dielectric slab. The
CPW with gap ( Z 0 = 98Q ) is matched to the mictrostrip TL with Z 0 = 50Q with XI4
transformer for testing purposes. The width of the signal trace S' is 2 mm and the width
o f the ground Gw is 6 mm. The gap G between the ground and the signal is 1 mm. For
Duroid 5880, we choose m\ = \ , m 2 = 2 , and /W3 = 4 so that we can use the same material
for all the dielectric slabs. The dielectric slabs must have
«, = n 2 = w3 = 4 / 3 n b
Since
(2.36)
lx=2>l%(X0lnb)
(2.37a)
/2 = 3 / 4 ( V " A)
(2.37b)
/3 = 3 / 2 ( V " a)
(2.37c)
= 1.23, the required dielectric constant o f the slab issr « 2.2. Polyethylene
which has sr = 2.25, is selected as a dielectric slab material. The lengths o f the dielectric
slabs which can be obtained using Equations (2.37a) through (2.37c) are 15.24 mm,
30.48 mm, and 60.97 mm for -45°,-90°, and -180° sections, respectively. The height of
the dielectric slab is 4 mm and the width is 19 mm. The bottom layer of the CPW
section does not have a ground plane and via is used for connecting to the ground.
Figure 2.9 shows the measured phase of S21 with the vector network analyzer (VNW A)
at seven phase shift positions created by the 3-bit dielectric slab combinations. The
measured phase is close to that o f the eight phase conditions (0, -45°, -90°, -135°, -180°,
-225°, -270°, and -315°). The reflections in all cases are small at 6 GHz. This
demonstrates that we can design the phase shifter using the dielectric slab on a CPW
without introducing impedance mismatch.
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Figure 2.7 Phase shifter without dielectric constant.
Figure 2.8 Phase shifter with a dielectric constant.
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22
S21 Phase M easurem ent (-45°, -90°, a n d -180° dielectric slabs)
-20
-40
-60
> - 88:2
-BO
£ -100
-160
-180
-200
5
5.2
5.4
5.6
5.8
6
6.2
6.4
6.6
6.8
7
Frequency (GHz)
S21 Phase Measurement (-135°,-225°,-270°,and -315° dielectric slabs)
-100
131.0
-150
-200
I—219.2
&
®
259.5
-250
^ -302.9'
-300
-350
5.2
5.4
5.6
5.8
6.2
6.4
6.6
6.8
Frequency(GHz)
Figure 2.9 The measured phase with the 3-bit dielectric slab combinations for the
desired eight phase conditions(0°, -45 °, -90 °, -135 °, -180 °, -225 °, -270 °, and -315 °).
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23
Chapter 3
A Continuously Steerable Array Antenna Using Movable Dielectric
Slabs on a Coplanar Waveguide
Microstrip Patch antennas are constructed using printed circuit fabrication
techniques which facilitate the construction o f a portion of the metallization layer that is
responsible for radiation [34]-[37]. Microstrip antenna patch elements, and arrays o f
patches, are the most common form o f printed antenna and were conceived in the 1950s
[38]. Extensive investigations of patch antennas began in the 1970s [39] and resulted in
many useful design configurations [40], [41]. Printed antennas are popular with antenna
engineers for several reasons: their low profile, the ease with which they can be
configured to specialized geometries, and their low cost when produced in large
quantities.
In the previous section, the phase shifter is discussed, and it is assumed that it is
working properly. It is widely known that a phase shifter is essential for a steerable
antenna. In addition, a phased-array antenna and feed network line are also significant
factors when designing the steerable antenna [42], [43]. In this Chapter, feed network
techniques using the preset delay line w ill be introduced. The impedance matching and
desired phase shift conditions are satisfied at two slab heights, and reflection is designed
to be minimized at all slab positions. A 4-element steerable array antenna at 5.8 GHz is
fabricated and measured for testing purpose.
3. 1. Continuous Phase Shifter Based on a Movable Dielectric Slab
In this Chapter, we introduce a movable dielectric slab placed close to a
coplanar waveguide (CPW) which can be used as a phase shifter for the array antenna.
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24
The proposed array antenna consists of a patch array antenna, phase shifter, and feed
network with a preset delay line as shown in Figure 3.1. The effective dielectric constant
and the characteristic impedance are calculated as a function o f slab height. We compare
the simulated radiation patterns and the measured radiation patterns of the steerable
array antenna at 5.8 GHz.
The basic concept o f the phase shifter is illustrated in Figure 3.3. The CPW has
airgaps between the center signal line and ground lines. As the movable dielectric slab
moves closer or into the gap o f the CPW, the effective dielectric constant changes and it
is given as a function o f d for a given structure. In this dissertation, we assume that the
slab can be moved continuously from the flash on the substrate ( d = 0 or very small) to
far away from the substrate ( d = oo or d > 2mm in our case). This structure can be
modeled as an unmatched TL section (or a layered structure). The transmission
coefficient T of a layered structure is given as
T=
1 - T ,e
-2 j6
(3.1)
where Tj is the reflection coefficient at the boundary due to a semi-infinite layer and
0 is the phase shift due to a slab. When 0 becomes an integer multiple of X/2, it is
well known that the reflection from the slab diminishes (or \T\ becomes 1). This is the
impedance matching condition. In addition, the phase shift of the slab section when d is
changed from d = oo to d = 0 , w ill be related to the effective dielectric constant of
these two states. Our primary concern w ill be a good impedance matching at all values
o f d. First, we set the matched impedance condition at d = 0 and d = oo. Since the
integer multiple of the X/2 condition must be satisfied at d = 0, this w ill determine the
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25
Microstrip
TL
Patch
Antenna
Coplanar
TL
Phase
Shifters
Preset Delay Lines
(Zo=103Q)
Microstrip
TL
^T ransform er
Zo=50Q
Input
Figure 3.1 Block diagram of a 4-element steerable array antenna.
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26
Preset Delay Lines
-315° delay
Phase Shifters Patch Antenna
*
Z<,=103Q
Zo=103 Q
-210“ delay
Microstrip Lines
50Q
Zo=103 O
-105" delay
Zo=103 O
0Udelay
Figure 3.2 Block diagram of a 4-element steerable array antenna.
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27
w
Figure 3.3 Phase shifter. A 3-D schematic o f a ground-signal-ground (G-S-G) CPW with
a movable dielectric slab. The width of the signal and ground traces are S (2 mm) and
Gw (6 mm), respectively. The gap between the ground and the signal is G (1 mm) and
the substrate thickness is h (1.6 mm). The length o f the dielectric material is /.
available length o f each slab. At d = oo or without slab, the impedance is always
matched. The main task, therefore, is to estimate the effective dielectric constant of
CPW with a slab on it. This process requires knowledge of the material characteristics,
TL structure, and a good 3-D simulation tool. Because we are designing a low-cost
antenna, we use a CPW structure fabricated on a FR-4 type substrate (IS620 from Isola)
which has a thickness o f 1.6 mm and esub = 3.73. The CPW has a signal trace width of
2 mm and a ground trace width o f 6 mm. The gap between the ground and the signal is 1
mm. Without the dielectric slab (d = 5 mm or effectivelyd = oo), the CPW has a
characteristic impedance o f Zo = 103 Q and an effective dielectric constant of seffective =
1.4. To simplify the fabrication process, we also used dielectric slabs without protruding
notches in the initial design. Therefore, when d = 0, the CPW gap is filled with air rather
than with the dielectric material. The dielectric constant of the slab must be much higher
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28
than that o f the substrate, but it cannot be too high. It is found that a high-^ material
such as BaTiC>3 creates a strong electromagnetic field perturbation and is not useful as a
slab in our design. We choose the Teflon-ceramic composite w ith fr =10.5 as a slab.
Using HFSS simulations, the desired effective dielectric constant is found to be about
^effect™ = 7.76 when d = 0. This w ill determine the length o f the X/2 section to be 9.32
mm, and the amount of phase shift with respect to the d = oo condition is 105°.
Therefore, if we want to introduce a 105° phase difference among four antenna elements,
the length should be 0, XI2, X, and 3X/2. Since we can’t find an analytical formula to
estimate Selective for d > 0, extensive HFSS simulations are conducted for 0 < d < 5 mm.
Because the slab length is set close to the integer multiple o f A /2 when d = 0, the
reflection from this slab w ill be small at d = 0. By setting e~2’e « 1 , we can approximate
T in Equation (3.1) asT « e~J0 . Then the phase change at the slab height d with respect
to that without a dielectric slab (d = °o) can approximately be expressed as
(3.2)
where k0 is the wavenumber in free space and
is the slab length. eeff_d=°° represents the
effective dielectric constant when the dielectric material is far enough away from the
substrate, and sef f j represents the effective dielectric constant when the slab height is d.
The effective dielectric constant for 0 < d < 5 mm is estimated from this phase. Table 3.1
presents numerical results for the G-S-G CPW with the center gap for cases with and
without the dielectric material. The characteristic impedance and the effective dielectric
constant change depending on the distance o f the dielectric material from the substrate.
As expected, the impedance is matched at d = 5 mm and d = 0. When d = 1.25 mm,
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
29
Table 3.1 Simulated results at d - 5 mm, d = 1.25 mm and d = 0 .
d=5
d = 1.25
d= 0
IS21 I
0.99
0.91
0.98
IS11I
0.01
0.36
0 .0 2
Characteristic impedance (Q )
103
83
51
Effective dielectric constant
1.4
3.73
7.76
Phase difference based on d - 5 mm
0°
49.5°
1O
—
H
O
Parameters (mm)
there is a slight impedance mismatch, although IS21I is still close to 1. Figure 3.4 shows
the effective dielectric constant as a function of distance d. Beyond d > 2 mm, the slab is
completely out o f the CPW and there is no change in eeffecttve- The effective dielectric
constant gradually changes from d = 2 mm to d = 0.25 mm and then increases rapidly at
d = 0. We chose the phase difference of -105° to + 105° between adjacent elements,
because the phase difference between d = 5 mm and d = 0 is 105° in Table3.1.
Figure 3.5 shows the magnitudes of S21 and Sn as a function of d. As expected, good
matching (small Sn) can be obtained at d = 0 and d > 2 mm, but the maximum value can
reach to 0.366 at d = 1.25 mm. The transmission coefficient, however, is always greater
than 0.9 at all positions of d. Although we cannot satisfy the impedance matching
condition at all values o f d for the continuous phase shifter, |Sn| is still less than -9 dB
(0.366) for this phase shifter.
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30
Effective dielectric constant v(e effective7
„ ^ ) as a function of slab height
0)
6u
a
SH
0
0.5
1
1.5
2
2.5
3.5
4.5
distance( mm) from the substrate
Figure 3.4 Effective dielectric constant vs height d from the substrate.
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S21 and S l l as a function of slab height(d)
0.9
0.8
S21
0.7
I
S >D-5
A
£ 0.4
S ll
0.3
0.2
0.5
1
1.5
2
2.5
3.5
4.5
distance( mm ) from the substrate
Figure 3.5 Simulated S21 and Sn results from d = 0 to d = 5 mm.
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32
3.2. Design o f the Feed Network Using the Preset Delay Lines
In order to reduce the number of phase shifters to N- 1 where N is the number of
arrays, the preset delay line is added to each feed line as shown in Figure 3.1. The basic
concept is illustrated in Figure 3.6. Suppose the antenna is designed with a phase
difference of -105° to +105° between adjacent elements, then the required phases at
different antenna elements are shown in Figure 3.6. The antenna element 1 which is
shown as S21 does not require an adjustable phase shifter, but it must have a -105° preset
phase with respect to element 2 (S31 in Figure 3.6). The largest phase shift o f 630° is
required for element 4 (S 51 in Figure 3.6). Therefore, by introducing the preset delay line,
we can eliminate one phase shifter.
The CPW has the characteristic impedance o f Z 0 = 103 Q with d = 5 mm and
an input impedance o f the patch antenna o f 103 Q . The main problem with the proposed
phase shifter is that the characteristic impedance also changes when the dielectric slab
position is changed from d = 5 mm (Zo = 103 Q ) to d = 0 (Z 0 = 51Q ). The lowest Zo
occurs at d = 0 , and thus the circuit must be designed to minimize the reflection at d = 0 .
Table 3.2
Phase relationship o f a 4-element array at 3 phase (slab) positions:-l 05°, 0 ,
and +105°.
Phase
Distance(mm)
S 21
S 31
S 41
S 51
Zo
-105°
d= 0
0°
210°
420°
630°
51
0
11
^3
0°
O
O
210°
315°
77
0°
0°
0°
0°
103
0°
+105°
d= 5
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Initial
condition
A ^ + 1 0 5
at </=5mm
+105° —
Element
Positions
AlP = 0°
at rf=0.5mm
A'P = -10^
at rf=0mm
Figure 3.6 Phase difference among 4 elements from -105° to +105°.
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34
In order to reduce the unwanted reflection, the effective length o f the CPW is set with
d = 0 to be mX 12 where m is an integer. Because a X 12 section does not create a
reflection, this w ill guarantee the impedance matching at d = 0. At other slab positions,
we estimate the characteristic impedance using HFSS. Table 3.2 shows the required slab
height d and the characteristic impedance Zo for the 4-element array antenna with the
phase difference at -105°, 0°, and +105°. Although the reflection is minimized at d = 0
and d - 5 mm, we have an impedance mismatch at d = 0.5 mm as shown in Table 2. For
the element phase differences of -105° to +105°, we can obtain the antenna beam scan
angle of -40° to +40°.
3.3. Verification o f a 4-Element Array Aantenna
A basic diagram o f the 4-element array antenna is shown in Figure 2.7. The
element separation is set to 22 mm which corresponds to 0.5/1. This w ill create a
grating mode at 180° phase difference, although a narrower beam can also be obtained
with only four elements. In order to remove the grating mode, the antenna separation
must be less than X 12.
The radiation pattern is plotted using Ansoft Designer. Ansoft Designer is based
on Method of Moment (M O M ) and is suitable for 2-D structure simulations. The
magnitude and the phase of the S21, S31, S41, and S51 channels are calculated
respectively. Each channel feeds the magnitude and phase simultaneously. Figure 2.7
illustrates the 4-element patch antennas’ schematic.
The initial phase is shifted to create a -90° phase difference at d= 0 mm (Figure 3.8 (a)).
At <7=5mm, the phase difference becomes +90° (Figure 3.8(c)) and at <7=0.5mm, the
phase difference becomes 0° (Figure 3.8(b)).
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35
S21
S31
S41
S51
Figure 3.7 A schematic of a 4-element patch antenna.
o
90
250
(a)
(b)
(c)
Figure 3.8 Simulated radiation patterns o f a 4-element array antenna.
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36
10mm
II
Figure 3.9 Fabrication o f the 4x1 array antenna.
o°
-30°
30°
-60°
60°
-90°
90°
20dB
-10dB
OdB
^ 120 °
120°
-150°
150°
±180°
Figure 3.10 4x1 measured FI-Plane radiation patterns.
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37
*'
-
Figure 3.11 Fabrication o f the 4x2 array antenna.
o
330
-32a Bm
300
-34dBm
270
90
240
120
210
150
180
Figure 3.12 4x2 measured H-Plane radiation patterns.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
Figure 3.13 Fabrication o f the 4x4 array antenna.
o
330
'dBm
300
90
240
120
210
150
180
Figure 3.14 4x4 measured H-Plane radiation patterns
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39
Figures 3.9, 3.11, and 3.13 show the fabrication o f the 4x1, 4x2, and 4x4 array
antenna, respectively. The Measured H-plane for a 4x1, 4x2 and 4x4 array antennas is
illustrated in Figures 3.10, 3.12 and 3.14, respectively. Before adding the phase shifter
component, or the CPW, we tested patch antennas with feed network parts. The
measured H-plane radiation patterns in Figures 3.10, 3.11, and 3.14 show that the
antennas are working properly and are ready to combine with the phase shifter
component.
3.4. Fabrication o f a 4-Element Steerable Array Antenna at 5.8 GHz
To demonstrate the feasibility of our concept, 4-element steerable array antennas are
designed and fabricated at 5.8 GHz for a wireless network. The array antennas are
designed to scan with a phase difference o f -105° to +105° and with an expected beam
scan angle o f -40 0 to +40°.
Figure 3.15 through 3.18 illustrate the fabricated 4x1 and 4x2 array antenna with and
without the dielectric slabs [44]-[47]. The substrate is IS620 with ssub = 3.73 and a
thickness o f 1.6 mm. The height o f the dielectric slab is 5 mm with a dielectric constant
o f sr = 10.5 which was obtained using Duroid 6010. The width o f the patch antenna is
17 mm and the height o f the patch antenna is 12.66 mm. In order to remove the grating
mode, the antenna separation is set to 22 mm which is less than X 12 while the input
impedance is 103 Q . The preset delay lines in Figure 3.15 through 3.18 have -315 0 for
element 1, -210° for element 2, and -105°, for element 3; whereas, the dielectric slabs
introduce a maximum phase shift o f -210° for element 2, -420° for element 3, and -630°
for element 4. This corresponds to the phase diagram shown in Figure 2.6. The lengths
o f the dielectric slabs on the CPW are 20.9 mm, 41.8 mm and 62.7 mm.
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40
Figure 3.19 through 3.21 show the simulated and measured H-plane radiation patterns
o f a 4x1 array antenna at 5.8 GHz. Figure 3.19 (a) is for d = 5 mm with the peak at -40°,
Figure 3.20 (a) is for d = 0.1 mm with the peak close to 0°, and Figure 3.21 (a) is for d =
0 with the peak at 30°. Figures 3.19 (b), 3.20 (b), and 3.21 (b) show the measured Hplane radiation pattern o f a 4x1 array antenna for the same three values o f d. Figure 3.22
through 3.24 show the simulated and measured H-plane radiation patterns o f a 4x2 array
antenna at 5.8 GHz. Actually, H-plane radiation patterns o f a 4x1 and a 4x2 array
antenna are almost same. Similarly, Figure 3.22 (a) is for d - 5 mm with the peak at -40°,
Figure 3.23 (a) is for d = 0.1 mm with the peak close to 0°, and Figure 3.24 (a) is for d =
0 with the peak at 30°. We obtain good agreement for the d = 0 and d = 0.1 mm cases for
a 4x1 and a 4x2 array antenna, respectively, but when the slab is on the CPW, the peak
of the beam is only at 13° in Figure 3.21(b). This discrepancy seems to be due to a small
residual gap (less than 10 pm) between the dielectric slab and conductors. This gap is
difficult to avoid when using flat bottom slabs. In order to solve this problem, we added
protruding notches to the flat bottom dielectric slabs, and tested them with a 4x2 array
antenna. Figure 3.24(b) illustrates the H-plane radiation patterns with protruding notches
at the flat bottom dielectric slabs. This modification makes the shape of slabs similar to
the one shown in Figure 3.3. With the addition o f small notches, the antenna beam can
now be scanned up to +40°.
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Figure 3.15 Photo o f a 4x1 array antenna without dielectric slabs.
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Figure 3.16 Photo o f a 4x1 array antenna with dielectric slabs.
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Figure 3.17 Photo o f a 4x2 array antenna without dielectric slabs.
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Figure 3.18 Photo o f a 4x2 array antenna with dielectric slabs.
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45
0°
-30'
■90'
90°
1 .5 d B
OdB
120'
-150'
150'
(a)
o°
-90°
90°
15dB
-10dB
+
5dB
OdB
180°
(b)
Figure 3.19 Radiation patterns of a 4x1 array antenna when d - 5 mm at 5.8 GHz. (a)
Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
46
0°
.-60'
■90'
5dB
OdB
120'
120'
-150'
150'
±180°
(a)
o°
-30‘
90°
OdB
120'
120'
150'
-150°
±180°
(b)
Figure 3.20 Radiation patterns o f a 4x1 array antenna when d = 0 at 5.8 GHz. (a)
Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern.
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47
-30'
60s
- -90'
90°
15dB
-1 M B
-5dB
OdB
120‘
150'
(a)
o°
-30‘
*60'
-90°
90°
15dB
5d B
OdB
120'
120°
■
150'
(b)
Figure 3.21 Radiation patterns o f a 4x1 array antenna when d = 0.1 mm at 5.8 GHz. (a)
Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern.
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48
0°
-30®
v-60'
90°
IS d S
/
-lO dB
.
.5dB
120'
OdB
120'
-150°
±180°
(a)
o°
30°
-90°
15dB
-IQ dB
5dB
OdB
120°
120°
-150'
150°
(b)
Figure 3.22 Radiation patterns o f a 4x2 array antenna when d = 5 mm at 5.8 GHz. (a)
Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern.
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49
0°
-
30'
v-60'
-90'
154B
lO dB
-5dB
OdB
120'
120 '
-150'
150'
(a)
o°
v-60'
60°
•90°
90°
isdB
.
-Ip d B
5dB
OdB
120°
•
(b)
Figure 3.23 Radiation patterns o f a 4x2 array antenna when d = 0.1 mm at 5.8 GHz. (a)
Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern.
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50
0°
90°
lOdB
OdB
120'
120°
150'
±180°
(a)
0°
60°
90°
lO dB
5dB
OdB
120°
150°
150°
(b)
Figure 3.24 Radiation patterns o f a 4x2 array antenna when d = 0 at 5.8 GHz. (a)
Simulated H-Plane radiation pattern (b) Measured H-Plane radiation pattern.
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51
Chapter 4
A 20 GHz Steerable Array Antenna Using 3-bit Dielectric Slab Phase
Shifters on a Coplanar Waveguide
The global satellite systems play a significant role in civilian and military
communications. Currently, several military communication systems are using the Kand Ka-band frequencies (18-40 GHz). Although the initial attempt by Teledesic was not
successful, it is expected that deployment o f high-speed satellite-based communication
links w ill be inevitable in the future. The current low-data rate satellite phones such as
Iridium w ill have a limited use as a high-speed data link. One o f the problems o f
Teledesic was the expected high cost o f a ground station which can track the low-earth
orbit (LEO) satellites. We believe that a low-cost steerable antenna is one of the missing
links for the future o f flexible wireless communication systems. The most flexible
satellite-to-ground communication system is based on electronic phased-array antenna
technology [48]-[50]. However, the cost o f a phased-array antenna is related to the
number of active elements, and thus the present systems are often too expensive for
many commercial and military applications. The antenna beam steering can also be done
by mechanically moving the reflector or lens [51]. Although the mechanically steerable
antennas can be less expensive than the electric phased-array antennas, the electro­
mechanical actuators/motors are usually bulky and prone to mechanical failure.
To minimize the reflection caused by the dielectric slab and also obtain the
desired phase shifter, the dielectric constant o f the slab has to be set to a specific value.
Figure 4.1 shows the final layout of the antenna designed at 20 GHz. Using a 3-bit phase
shifter, the antenna beam angle can be scanned from -45° to +45°. Details w ill be
discussed in the following sections.
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Microstrip
TL
Phase
Shifters
Coplanar
TL
Microstrip
TL
I
Preset Delay Lines
(Zo=106.5Q)
^ Transformer
Zo=50Q
Input
Figure 4.1 Block diagram of a 4x4 steerable array antenna.
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53
4.1. Phase Shifter Based on a Movable 3-Bit Dielectric Slab
The basic concept o f the phase shifter is illustrated in Figure 4.2. The CPW has airgaps
between the center signal line and ground lines. As the movable dielectric slab moves
closer or into the gap o f the CPW, the effective dielectric constant changes and is given
as a function o f d for the given structure. In this dissertation, we assume that the slab can
be either attached on the substrate ( d = 0 or very small) or far away from the substrate
(d = °o or c/> 5 mm in our case).
This structure can be modeled as an unmatched TL section (or a layered
structure). The transmission coefficient T of a layered structure is given as
(4.1)
where Tx is the reflection coefficient at the boundary due to a semi-infinite layer and
0 is the phase shift due to a slab. When 0 becomes m% where m is an integer, the
reflection from the slab diminishes (or |7] becomes 1). This is the impedance matching
condition which can also be stated in terms o f the effective slab length of mX/2. The slab
length must satisfy this condition to minimize reflection. In addition, when d is changed
from d = oo to d = 0 , the phase shift o f the slab section w ill be related to the effective
dielectric constant o f these two states. Therefore, if we want to create a desired phase
shift without creating undesired reflection, we need to satisfy specific conditions for
each slab.
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ir ir
Figure 4.2
Phase shifter. A 3-D schematic of a ground-signal-ground (G-S-G) CPW
with a movable dielectric slab. The width of the signal and ground traces are S (0.8 mm)
and Gw (2.5 mm), respectively. The gap between the ground and the signal is G (0.4
mm) and the substrate thickness is h (0.508 mm). The height of the dielectric material is
2.5 mm and its length is /.
^1
^2
^3
-o----------------------- o----------------------- o----------------------- o
Z"o’
o ’ " bb
ZV
1’ "w11
Z~,
n<2’ ‘2
Z~, w,3
‘ ■3’
Z o ’, nv
b
o----------------------- o----------------------- o----------------------- o-----------
Figure 4.3 TL model of a 3-bit phase shifter.
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55
4.1.1. Impedance Mismatch and Possible solutions
Suppose we want a 3-bit phase shifter given by 45°, 90° and 180° sections as
shown in Fig. 3. Then the available phase shifts w ill be eight states (0°, 45°, 90°, 135°,
180°, 225°, 270°, and 315°).
The fixed CPW section is denoted by Zo and nbwithout dielectric slabs. The
sections l\, h, and h w ill have dielectric slabs, and are given by (Z i, n{) for
the # / 4 section,
* ) for the # /2 section,
( Z 2 , 72
) for the# where Z and n are the
( Z 3 , 773
characteristic impedance and the index o f refraction o f each section. We hope to create
conditions with no reflection for all states. We also aim to obtain the lengths l\, h, and h
as well as the index o f refraction t?i, t72, and
773
in terms of n\>. This can be done by
satisfying the following conditions:
1) For # /4 section (45°, bit 0), we need to satisfy
(t7j - n h)k0lx = # /4 and nlk0ll = m xn
where 7771 is an integer. By eliminating /,, 771 is given by
4 777,
nx = nb~A —
4 t77j -1
(4.2)
I f we choose 777, = 1, then
4
77, = - 77,
3
, ,
and /, =
7 7 7 ,#
7 7 jX 0
3 1
= —•—
8 77,
(4.3)
2) For # /2 section (90°, bit 1), we need to satisfy
(t7 2
- n b)k0l2 = # /2
and n2k0l2 = m 2n
where m2 is an integer. By eliminating l2, n2 is given by
2w 2
"2 = ”b~— —
2 t772 —1
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/>.
A\
(4-4)
56
(4 .5)
3) For# section (180°, bit 2), we need to satisfy
(«3 - n A)k 0/3 = n and n3k0/3 = m^n
where m3 is an integer. By eliminating /3, «3 is given by
(4.6)
(4.7)
4.2. 4x4 Array Antenna with Phase Shifter at 20 GHz
A low-cost material and fabrication process are among our top design goals. The
20 GHz array antenna and phase shifters are fabricated on the FR-4 based IS640
substrate which is less expensive than Duroid 5880. The characteristics of IS640
are£r = 3 .3 8 , thickness = 0.508 mm (20mil), and loss tangent = 0.0045, respectively.
The 4x4 array antenna was chosen after testing the E- and H-plane radiation patterns of
4x1,4x2, and 4x4 array antennas [52]. Figures 4.4 and 4.6 show the fabrication of a 4x2
and a 4x4 array antenna for testing purpose, respectively. Figures 4.5 and 4.7 show the
E- and H-plane radiation patterns o f a 4x2 and a 4x4 array antenna. E- and H-plane
radiation patterns are in agreement and working properly for a 4x2 and 4x4 array
antenna. They are shown in Figures 4.4 and 4.6, respectively [53].
To reduce the E-plane beam width, four patch antennas are connected in series
as shown in Figure 4.7. Although the E-plane side-lobe can be reduced by tapering the
patch sizes, it was not included in our design. The antenna patch size is 5.8 mm x 3.85
mm and the input impedance is 106.5 Q. The spacing for series elements between edges
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57
1 SJTl 1r': i
1
I
Figure 4.4 Fabrication o f the 4x2 array antenna at 20 GHz.
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58
0
330
■14dfem
300
60
■16.5dBm
-2Q4Bm
270
90
240
120
210
150
180
(a)
0
330
■14dBm
300
270
90
240
120
210
150
180
(b)
Figure 4.5 4x2 Measured radiation patterns (a) H-plane radiation pattern (b) E-plane
radiation pattern.
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Figure 4.6 Fabrication o f the 4x4 array antenna at 20 GHz.
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60
o
330
■14dBm ,. a ;
60
300
90
270
120
240
150
210
180
(a)
0
330
14dBm
300
90
270
120
240
150
210
180
(b)
Figure 4.7 4x2 Measured radiation patterns (a) H-plane radiation pattern (b) E-plane
radiation pattern.
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61
is 4 mm, and the spacing for patches between center to center is 7.4 mm which is less
than 0.5 X at 20 GHz. The E-plane beam size can further be reduced by increasing the
number of series elements [54], [55].
The CPW section has a signal line width of S = 0.8 mm, a ground line width of
Gw= 2.5 mm, and a gap between the signal line and the ground line o f G = 0.4 mm. The
bottom layer o f the CPW section does not have a ground plane and via is used to
connect to the ground. Based on HFSS simulations, it is found that the effective index of
refraction without slab is n* =1.152 (£ eff = 1.329). Ideally, the dielectric slab should fill
the gap when d = 0 as shown in Figure 4.2. Unfortunately, this requires additional costly
machining. To reduce processing costs, we investigated the use o f slabs without the
bottom notches which go into the CPW gaps. The disadvantage o f this flat slab is that
the effective dielectric constant can be sensitive to a small residual air space and its
effect must be included in the simulation. For n f 4 section, we choose mi = 1 and obtain
n\ = 1.536 and l\ = 4.48 mm. The effective dielectric constant o f this section must be
£eff\ = n\ = 2.359. To create £effl= 2.359, we use a dielectric material w ith£r = 3.7 3.
Table 4.1 shows the effective dielectric constant as a function of residual air space. I f a
small airgap (10-20/ffli) exists between the dielectric slab and the conductor, the
dielectric material with er = 3.73 is suitable for 45° sections.
For 7t/2 and n sections, we choose m2 - 1 and m3 = 2 , and obtain «2 = «3 = 2«b= 2.305, h
= 3.26 mm, and I 3 = 6.52 mm. The effective dielectric constant o f these sections must be
£'ffl=n\ = 5.316. To create seffl= 5.316, we use a dielectric material withgr = 1 0 .2 . Table
4.2 shows the effective dielectric constant as a function of residual air space for the 7i/2
section. I f a small airgap (5-10/rm) exists between the dielectric slab and the conductor,
the dielectric material with er = 10.2 becomes suitable for 90° and 180° sections.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
62
TABLE 4.1
Simulation results withe,. =3.73 slab (slab length = 4.48 mm)
d (fzm)
IS21 I
IS11I
£ e ff\
20
0.976
0.073
2.33
19
0.976
0.068
2.341
18
0.976
0.067
2.358
17
0.976
0.064
2.363
16
0.976
0.06
2.375
A ir
0.987
0.006
1.329
TABLE 4.2
Simulation results withe, = 10.2 slab (slab length= 3.26 mm)
d (/im)
IS21 I
|Sn|
£ e ffl
8
0.979
0.070
4.936
5
0.982
0.011
5.308
4
0.982
0.02
5.42
3
0.981
0.04
5.57
A ir
0.987
0.006
1.329
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
Figure 4.8 Photo o f a 4x4 array antenna without dielectric slabs.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
64
M bWi
BitO
*=3.73
BitO
£,=3.73
BitO
£,=3.73
BitO
*=3.73
Bit 1
£,=10.2
Bit 1
*=10.2
Bit 1
£,=10.2
Bit 1
*=10.2
Bit 2
0.2
Bit 2
*=10.2
Bit 2
*=10.2
Bit 2
*=10.2
Figure 4.9 Dielectric slab positions on a CPW for the 3-bit phase shifter.
S l l measured data
PQ
Ie
I
-15
-20
18.5
19.5
20.5
21.5
Frequency(GHz)
Figure 4.10 Measured Sn of a 4x4 array antenna without slabs as shown in Figure 4.8
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
65
The feeding network is a simple one-to-four power divider with X74
impedance matching sections as shown in Figure 4.8. Also shown in Figure 4.9 are the
positions o f the dielectric slabs for the 3-bit phase shifter. Although twelve actuators w ill
be required to move them independently, a simple mechanism can be used if this
antenna is utilized for the satellite tracking. The input reflection Sn is measured with
VNW A and shown in Figure 4.10. It is better than -15 dB at 20 GHz.
4.3. Simulated and Measured E- and H-Plane Radiation Patterns o f a 4x4
Array Antena
The radiation patterns measured at different phase shift conditions are shown in
Figure 4.11 through Figure 4.15. Figure 4.8 shows a 4x4 array antenna without the
dielectric slabs on it. This corresponds to the no-phase shift among 4 elements, and,
therefore, we should expect a beam peak at 0°. Figure 4.11 shows the measured and
simulated H-plane radiation patterns o f this case, and Figure 4.12 shows that of E-plane.
The simulated and measured E- and H-plane radiation patterns are in agreement. The
large sidelobe in E-plane (Figure 4.12) may be due to the interference from the feeding
network. This can be reduced by placing the CPW and feeding network sections on the
bottom side.
Figure 4.13 through 4.15 show the radiation patterns with the phase shift. In all
cases, the measured data is normalized with respect to the peak value o f the 0° phase
shift (Figure 11). Figure 4.13 shows the simulated and measured H-plane radiation
patterns with dielectric slabs on a CPW. The phase shifts from left to right are -135°, 90°, -45°, and 0° which correspond to the 3-bit patterns o f (110,010,100,000). The
beam scan angle (peak) is at -15° (345°) for this configuration. Figure 4.14 shows the
simulated and measured H-plane radiation patterns with dielectric slabs on the CPW.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
66
These phase shifts from left to right are -270°, -180°, -90°, and 0° which corresponds
to the 3-bit patterns o f (011,001,010,000). The beam scan angle (peak) is at -30° (330°)
for this configuration. Figure 4.15 shows the maximum beam scan angle of +45° which
can be obtained with -135° phase shifting among elements. From left to right, the phase
shift is 0°, -135°, -270°, and -405° (45°) with the corresponding 3-bit pattern of
(000, 110,011, 100).
Figure 4.11 through 4.15 are four examples o f the eight possible states which
can be obtained from a 3-bit phase shifter. Other beam scan angles can be obtained by
changing the dielectric slab positions. With this simple setup, the main beam can be
scanned from -45° to +45° at 20 GHz. For a 3-bit phase shifter, each CPW has three
dielectric slabs. The 4-element array, therefore, requires twelve slabs as illustrated in
Figure 4.9. I f arbitrary beam scanning is needed, twelve actuators are required. As a
possible application for satellite tracking antennas, whose scan angle can be predicted
based on the satellite position, we could use the position-coded rod for each bit. In our
case, three rods with notches that press down on the dielectric slabs w ill be sufficient.
This is exemplified in Figure 4.16. The ON-OFF positions are pre-coded on the rod as
notches. By moving all 3 rods simultaneously, we can scan the beam from -45° to +45°.
This idea w ill require only one motor. I f a 2-D scan is needed for satellite tracking, the
rotational stage with another motor w ill be sufficient as illustrated in Figure 4.17. The
expected height o f this antenna w ill be relatively thin, and it w ill basically be a planar 2D scanning antenna. The layout in Figure 4.17 has CPW and feeding network on the
bottom side o f PCB. This configuration w ill reduce the interference in radiation patterns
due to mechanical items placed on the top side.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
67
0°
-30"
s-60'
:.............
-20 dB
-10dB
H-90‘
OdB
H-Plane M easured
120'
H-Plane Simulated
150'
-150'
Figure 4.11 Simulated and measured H-plane Radiation Patterns: No phase shift case.
Slab positions are (000,000,000,000).
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
68
0°
-30'
c60
-20dB
-lOdB
E -Plane Measured
120'
OdB
120
'
E -Plane Simulated
150'
-150'
Figure 4.12 Simulated and measured E-plane Radiation Patterns: No phase shift case.
Slab positions are (000,000,000,000).
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
69
0°
-1-90"
20 dB
-lOdB
OdB
H-Plane Measured
120'
H-Plane Simulated
150'
-150'
Figure 4.13 Simulated and measured H-plane Radiation Patterns. The phase shift among
elements is -45° which corresponds to the beam angle o f 15°. Slab positions are
(110,010,100,000).
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
70
0°
*60'
.......
20 dB
-J-90'
-lOdB
H -Plane M easured
120'
OdB
■120'
H -Plane Simulated
150'
-150'
Figure 4.14 Simulated and measured H-plane Radiation Patterns: The phase shift among
elements is -90° which corresponds to the beam angle o f 30°. Slab positions are
(011,001,010,000).
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
71
0°
-30'
,-60'
■;............ -1-90'
-20 dB
-10dB
H-Plane Measured
120'
OdB
120'
H-Plane Simulated
150'
-150'
Figure 4.15 Simulated and measured H-plane Radiation Patterns: The phase shift is 135° which corresponds to the beam scan angle of 45°. Slab positions are
(000,110,011,100).
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
Figure 4.16 Actuators for a 3-bit phase shifter. Each rod actuates 4 slabs with the same
phase shift and all rods are synchronized. One bit pattern on the top section shows the
( 110 ,0 1 0 , 100 ,0 0 0 ) case.
T
u
r
n
t
a
b
l
e
Figure 4.17 Conceptual view o f a 2-D satellite tracking antenna based on our tunable
design. CPW and feeding network are on the bottom side of PCB.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
73
Chapter 5
A Steerable Phased-Array Antenna Using Mechanically Controllable 4Bit Dielectric Slab Phase Shifters on a Coplanar Waveguide
Automobile collision avoidance radar (CAR) has become a recent feature in
automotive design. 24 GHz steerable antennas are currently being used in new
commercial applications such as W LAN (Wireless local area network) and CAR system
[56]- [58]. In order for automotive radar to be most effective, a combination o f mediumrange detection (in front o f the car) and short-range detection (in all directions) is
required [59], [60]. The 24 GHz collision avoidance radar in an automobile and W LAN
require a beam scanning mechanism.
The phase shifters are a critical element for electronically scanned phased-array
antennas, and typically account for a significant amount of the cost of producing an
antenna array. The reduction o f fabrication cost opens possibilities for many applications.
In this Chapter, we propose that a movable dielectric slab which is placed close to a
coplanar waveguide (CPW) with an airgap can be used as a phase shifter for the array
antenna. The proposed array antenna consists of series-fed patch antennas, phase shifters,
and the feeding network as shown in Figure 5.1. To minimize the reflection caused by
the dielectric slab and also to obtain the desired phase shifter, the dielectric constant of
the slab has to be set to a specific value. Previous studies have demonstrated that the
performance o f the CPW-based phase shifter has been tested. Figure 5.1(b) shows the
final layout of the antenna designed at 24 GHz. We have already produced the similar
schematic of Figure 5.1(a) [61]. In the actual system illustrated in Figure 5.1(a), the
CPW section unexpectedly radiated, causing the radiation pattern of the E-plane to be
slightly distorted. The large sidelobe in the E-plane (Figure 5.1(a)) may be due to the
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
74
Patch
Antenna
Microstrip
TL
Phase
Shifters
Coplanar
TL
Microstrip
TL
T _J
T
-i-* (Zo=105O)
/^Transformer
Zo—500
Input
^10mm
(a)
(b)
Figure 5.1 Block diagram o f an 8x7 steerable array antenna, (a) Front Feeding (b) Back
Feeding.
interference from the CPW and the feeding network. This can be reduced by placing
both the CPW and the feeding network section on the bottom side (Figure 5.1(b)). One
o f the advantages to this newly proposed phase shifter design is that the whole antenna
can be designed without using any solid state phase shifters or MEMS devices. By using
a 4-bit phase shifter, the antenna beam angle can be scanned from -52.5° to +52.5°.
Details w ill be further discussed in the following sections.
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75
5.1. Phase Shifter Based on a Movable 4-bit Dielectric Slab
The basic concept of the phase shifter is illustrated in Figure 5.2. The CPW has airgaps
between the center signal line and ground lines. As the movable dielectric slab moves
closer or into the gap o f the CPW, the effective dielectric constant changes and is given
as a function of d for the given structure. In this dissertation, we assume that the slab can
be
either attached on the substrate ( d = 0 or very small) or far away from the
substrate ( d =
qo
or d > 5 mm in our case).
This structure can be modeled as an unmatched TL section (or a layered
structure). The transmission coefficient T of a layered structure is given as
(5.1)
where r ( is the reflection coefficient at the boundary due to a semi-infinite layer and
6 is the phase shift due to a slab. When 6 becomes mn where m is an integer, the
reflection from the slab diminishes (or |7] becomes 1). This is the impedance matching
condition which can also be stated in terms of the effective slab length o f mX/2. The slab
length must satisfy this condition to minimize reflection. In addition, when d is changed
from d = oo to d = 0 , the phase shift of the slab section w ill be related to the effective
dielectric constant of these two states. Therefore, if we want to create a desired phase
shift without creating undesired reflection, we need to satisfy specific conditions for
each slab.
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76
Figure 5.2 Phase shifter. A 3-D schematic of a ground-signal-ground (G-S-G) CPW with
a movable dielectric slab. The width o f the signal and ground traces are S (0.8 mm) and
Gw (4.6 mm), respectively. The gap between the ground and the signal is G (0.4 mm)
and the substrate thickness is h (0.508 mm). The height of the dielectric material is 2.5
mm and its length is /.
/,
i2
/3
/4
-o ----------------------------o----------------------------o ----------------------------o------------------------------o -
Z Q, n b
Z v n,
Z 2, n 2
Z 3, «3
Z4,
n4
Z 0, nb
o--------------------- o--------------------- o--------------------- o-----------------------o-----------
Figure 5.3 TL model of a 4-bit phase shifter.
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77
Suppose we want a 4-bit phase shifter given by 22.5 °, 45 °, 90 0 and 180 0
sections as shown in Figure 5.3. Then the available phase shifts w ill be sixteen states (0°,
22.5°, 45°, 67.5°, 90°, 112.5°, 135°, 157.5°, 180°, 202.5°, 225°, 247.5°, 270°, 292.5°,
315°, and 337.5°).
The fixed CPW section is denoted by Zo and nhwithout dielectric slabs. The
sections h, 1% h and U w ill have dielectric slabs, and are given by (Z i, n\) for
the n 18 section, (Z 2, ni) for the
71/ 4
section, (Z 3, ni) for the n j 2 , and (Z 4, nt) for
the n where Z and n are the characteristic impedance and the index of refraction of each
section. We hope to create conditions with no reflection for all states. We also aim to
obtain the lengths l\, h, h and U as well as the index of refraction m, ni, «3 and «4 in
terms o f «b- This can be done by satisfying the following conditions:
1) For
71 18 (22.5
°, bit 0), we need to satisfy
( nl - n b) k 0ll =7r/S and nxk0lx = mx7z
where m\ is an integer. By eliminating lx, n\ is given by
8wi
«i = nb- —
(5. 2)
8/Wj - 1
For wi = 1, we have nx = —nh and lx= - - —
7
16 nb
2) For
71/4
section (45 °, bit 1), we need to satisfy
(n2- n b)k 0l2 =7t/4 and n2kQl2 = m2n
where m2 is an integer. By eliminating l2, «2 is given by
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
(5.3)
78
4 m2
(5 .4 )
4m2 -1
I f we choose m2 = 1, then
4
«2 = t « a
3
m2;r
34,
/ = - L - = - . J 2.
n2x0
8 wA
(5.5)
3) For ;r/2 section (90 °, bit 2), we need to satisfy
(n3 - n b)k 0l3 = n f 2
and n3k0l3 = m3n
where m3 is an integer. By eliminating l3, n%is given by
2m,
(5.6)
2 m3 - 1
For m3 = 1 : n3 - 2nb and l3 = - ^ ~
4«a
(5.7)
4) For n section (180 °, bit 3), we need to satisfy
(n 4 - n b)k 0l 4 = n and nAkalA= mAn
where m$ is an integer. By eliminating /4, «4 is given by
«4 =
«
A
m4 - 1
For m4 = 2: nA= 2nband /4 = —•—
2
(5 -8 )
(5.9)
5.2. 8x7 Tapered Array Antenna with Phase Shifter at 24 GHz
A low-cost material and fabrication process are among our top design goals. The
24 GHz array antenna and phase shifters are fabricated on the FR-4 based IS640
substrate which is less expensive than Duroid 5880. The characteristics of IS640
arQ£r = 3.38, thickness = 0.508 mm (20mil), and loss tangent = 0.0045, respectively. To
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
79
TABLE 5.1
Simulation results withsr = 2.94 slab (slab length= 4.74 mm).
d (/mi)
IS21 I
IS11I
S e ff\
120
0.983
0.007
1.684
115
0.98
0.007
1.701
110
0.987
0.003
1.726
105
0.985
0.008
1.752
Air
0.989
0.009
1.33
TABLE 5.2
Simulation results withsr = 3.73 slab (slab length = 4.06 mm).
d (/mi)
IS21 I
|Sn|
%2
21
0.982
0.01
2.34
20
0.982
0.01
2.36
19
0.983
0.02
2.38
18
0.981
0.01
2.408
A ir
0.989
0.009
1.33
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
80
TABLE 5.3
Simulation results w ith ^ = 10.2 slab (slab length=2.71 mm).
d (fan)
IS21 I
IS11I
£effT,
8
0.979
0.070
5.08
7
0.982
0.011
5.177
6
0.982
0.02
5.317
5
0.981
0.04
5.53
reduce the E-plane beam width, seven patch antennas are connected in series as shown
in Figure 5.4. In addition the E-plane side-lobe can be reduced by tapering the patch
sizes which is enough to realize aimed Chebyshev Array distribution [62]-[63]. The
antenna patch sizes are 4.8 mm x 3.2 mm, 4.395 mm x 3.2 mm, 3.332 mm x 3.2 mm and
2.61 mm x 3.2 mm, respectively. The input impedance is 105 Q . The spacing for series
elements between edges is 3.5 mm, and the spacing for patches from center to center is
6.2 mm which is less than 0.5 X at 24 GHz. The 8x7 array antenna was chosen after
testing the E- and H-plane radiation patterns o f 4x4, and 8x4 array antennas. The Eplane beam size can further be reduced by increasing the number of series elements.
The CPW section has a signal line width of S = 0.8 mm, a ground line width of
Gw= 4.6 mm, and a gap between the signal line and the ground line of G = 0.4 mm. In
order to get rid o f the interference from the CPW and the feeding network, both the
CPW and the feeding network section are designed to be placed on the bottom side. The
top layer o f the CPW section does not have a ground plane and via is used to connect to
the ground. Based on HFSS simulations, it is found that the effective index o f refraction
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
81
without slab is
=1.153
1.33). Ideally, the dielectric slab should fill the gap
when d = 0 as shown in Figure 5.2. Unfortunately, this requires additional costly
machining. To reduce processing costs, we investigated the use o f slabs without the
bottom notches which could go into the CPW gaps. The disadvantage o f this flat slab is
that the effective dielectric constant can be sensitive to a small residual air space and its
effect must be included in the simulation. For the n!% section, we choose m\ = 1, and
obtain n\ = 8/7«b = 1-32, l\ = 4.74 mm. The effective dielectric constant of these sections
must beseffl
= 1.73. To c rea te f^ = 1.73, we use a dielectric material w ithsr = 2.94.
Table 5.1 shows the effective dielectric constant as a function o f residual air space for
the 7r/% section. I f a small airgap (100-110/zm) exists between the dielectric slab and the
conductor, the dielectric material with sr = 2.94 is suitable for the 22.5 0 section. For
njA section, we choose m2 - 1 and obtain m = 1.537 and h = 4.06 mm. The effective
dielectric constant o f this section must b
= n\ = 2.36. To create seff2= 2.36, we use
a dielectric material w ith fr = 3.73. Table 5.2 shows the effective dielectric constant as a
function of residual air space. I f a small airgap (10-20//m) exists between the dielectric
slab and the conductor, the dielectric material with er =3.73 is suitable for the 45°
sections. For 7tl2 and n sections, we choose m^= 1 and m\ = 2, and obtain n^= n4=
2«b= 2.306, h = 2.71 mm, and U = 5.42 mm. The effective dielectric constant of these
sections must beseffi = n] = 5.317. To createseff3= 5.317, we use a dielectric material
with£r = 10.2 . Table 5.3 shows the effective dielectric constant as a function o f residual
air space for the n 12 section. I f a small airgap (5-10/zm) exists between the dielectric
slab and the conductor, the dielectric material with sr = 10.2 becomes suitable for the
90° and 180° sections.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
82
(a)
(b)
Figure 5.4 (a) Top view of an 8x7 array antenna without dielectric slabs, (b) Bottom
view of an 8x7 array antenna without dielectric slabs.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
83
B itO
sr=2.94
B itO
8r=2.94
B itO
Sr= 2.94
B itO
8r=2.94
B itO
sr=2.94
B itO
8r=2.94
B itO
8r=2.94
B itO
8r=2.94
B it 1
8^3.73
B it 1
8r=3.73
B it 1
8r=3.73
B it 1
8r=3.73
B it 1
8f“ 3.73
B it 1
8^3.73
B it 1
8F 3.73
B it 1
81—3.73
B it 2
8r=10.2
B it 2
8r=10.2
B it 2
8r=10.2
B it 2
8r=10.2
B it 2
8r=10.2
B it 2
8 ^ 1 0 .2
B it 2
8r=10.2
B it 2
8r=10.2
B it 3
8r=10.2
B it 3
8r=10.2
B it 3
8r=10.2
B it 3
er=10.2
B it 3
er=10.2
B it 3
8r=10.2
B it 3
8r=10.2
B it 3
8r=10.2
Figure 5.5 Dielectric slab positions on a CPW for the 4-bit phase shifter.
S l l m easured data
-10
-25
-30
22.5
23.5
24.5
25.5
Frequency(GHz)
Figure 5.6 Measured Sn o f an 8x7 array antenna without slabs as shown in Figure 5.4.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
84
The radiation patterns measured at different phase shift conditions are shown
in Figure 5.7 through 5.13. Figure 5.4 shows an 8x7 array antenna without the dielectric
slabs on it. This corresponds to the no-phase shift among 7 elements, and, therefore, we
should expect a beam peak at 0 °. Figure 5.7 shows the measured and simulated H-plane
radiation patterns of this case, and Figure 5.8 shows that of E-plane. The simulated and
measured E- and H-plane radiation patterns are in agreement.
Figure 5.9 through 5.13 show the radiation patterns with the phase shift. In all
cases, the measured data is normalized with respect to the peak value of the 0 0 phase
shift (Figure 5.7). Figure 5.9 shows the simulated and measured H-plane radiation
patterns with dielectric slabs on a CPW. The phase shifts from left to right are -157.5 °, 135 °, -112.5 °, -90 °, -67.5 °, -45 °, -22.5 °, and 0 0 which correspond to the 4-bit patterns
o f (1110, 0110,1010, 0010,1100, 0100,1000, 0000). The beam scan angle (peak) is at 7.5° (352.5°) for this configuration. Figure 5.10 shows the simulated and measured Hplane radiation patterns with dielectric slabs on the CPW. These phase shifts from left to
right are -315°, -270°, -225 °, -180°, -135°, -90°, -45°, and 0 ° which correspond to the
4-bit patterns o f (0111, 0011, 0101, 0001, 0110, 0010, 0100,0000). The beam scan angle
(peak) is at -15 0 (345°) for this configuration. Figure 5.11 shows the simulated and
measured H-plane radiation patterns with dielectric slabs on a CPW. The phase shifts
from left to right are -270° (-630°), -180° (-540°), -90° (-450°), 0 ° (-360°), -270°, 180°, -90 °, and 0 ° which correspond to the 4-bit patterns o f (0011, 0001, 0010, 0000,
0011, 0001, 0010, 0000). The beam scan angle (peak) is at -30 0 (330 °) for this
configuration. Figure 5.12 shows the simulated and measured H-plane radiation patterns
with dielectric slabs on the CPW. These phase shifts from left to right are -225 0 (-945 °),
-90° (-810°), -315° (-676°), -180° (-540°), -45° (-405°), -270°,-135°, and 0 ° which
correspond to the 4-bit patterns o f (0101, 0010, 0111, 0001, 0100, 0011, 0110, 0000).
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
85
The beam scan angle (peak) is at -45 0 (315°) for this configuration. Figure 5.13 shows
the maximum beam scan angle o f -52.50 (307.5 °) which can be obtained with -157.5 0
phase shifting among elements. From left to right, the phase shift is -22.5 0 (-1102.5 °), 225° (-945°), -67.5° (-787.5°), -270° (-630°), -1125° (-472.5°), -315.5°,-157.5°, and 0
° with the corresponding 4-bit pattern o f (1000, 0101, 1100, 0011, 1010, 0111, 1110,
0000).
Figure 5.7 through 5.13 are six examples o f the sixteen possible states which
can be obtained from a 4-bit phase shifter. Other beam scan angles can be obtained by
changing the dielectric slab positions. With this simple setup, the main beam can be
scanned from -52.5° to +52.5° at 24 GHz. For a 4-bit phase shifter, each CPW has four
dielectric slabs. The 8-element array, therefore, requires thirty two slabs as illustrated in
Figure 5.5. I f arbitrary beam scanning is needed, thirty two are required. As a possible
application for automobile collision avoidance radar, whose scan angle can be predicted,
we could use the position-coded rod for each bit. In our case, four rods with notches that
press down on the dielectric slabs w ill be sufficient. The ON-OFF positions are precoded on the rod as notches. By moving all 4 rods simultaneously, we can scan the beam
from -52.5° to +52.5°. This idea w ill require only one motor
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
86
0°
-30"
v-60'
■■7T> ...........
-2 0 dB
120’
-lO d B
i-90‘
OdB
H-Plane Measured
H-Plane Simulated
150'
-150'
Figure 5.7 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna: No phase shift case. Slab positions are (000,000,000,000, 000,000,000,000).
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87
0
°
-30'
,-60'
-2 0 dB -lO d B
E -P lane Measured
120'
OdB
120'
E -Plane Simulated
150'
-150'
Figure 5.8 Simulated and measured E-plane Radiation Patterns o f an 8x7 tapered array
antenna: No phase shift case. Slab positions are (000,000,000,000, 000,000,000,000).
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88
0°
-30“
.......
-2 0 dB -lOdB
H-Plane M easured
120'
i-90'
OdB
120'
H-Plane Simulated
150'
-150'
Figure 5.9 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna: The phase shift among elements is -22.5° which corresponds to the beam angle
o f 7.5°. Slab positions are (1110, 0110,1010, 0010, 1100, 0100,1000, 0000).
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89
0°
-30'
,-60'
- j- 9 0 '
-2 0 dB -lOdB
120'
OdB
120'
H-Plane Measured
H-Plane Simulated
150'
-150'
Figure 5.10 Simulated and measured H-plane Radiation Patterns o f an 8x7 tapered array
antenna: The phase shift among elements is -45° which corresponds to the beam angle
o f 15°. Slab positions are (0111, 0011, 0101, 0001, 0110, 0010, 0100, 0000).
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90
0°
-30“
v-60'
i
-20dB -lOdB
-|-90'
OdB
H-Plane Measured
120 '
H-Plane Simulated
150'
-150'
Figure 5.11 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna: The phase shift among elements is -90° which corresponds to the beam angle
o f 30°. Slab positions are (0011, 0001, 0010, 0000, 0011, 0001, 0010, 0000).
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91
0°
-30'
60'
t
-2 0 dB -lOdB OdB
H-Plane Measured
120 '
'
120'
H-Plane Simulated
150'
-150'
Figure 5.12 Simulated and measured H-plane Radiation Patterns of an 8x7 tapered array
antenna: The phase shift among elements is -135° which corresponds to the beam angle
of 45°. Slab positions are (0101, 0010, 0111, 0001, 0100, 0011,0110, 0000).
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92
0
°
-30'
,-60'
-20dB -lOdB
120"
H-Plane Measured
OdB
120'
H -Plane Simulated
150'
-150"
Figure 5.13 Simulated and measured H-plane Radiation Patterns o f an 8x7 tapered array
antenna: The phase shift among elements is -157.5° which corresponds to the beam
angle o f 52.5°. Slab positions are (1000, 0101,1100, 0011,1010,0111, 1110, 0000).
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93
5.3. 8x4 Array Antenna with Phase Shifter at 24 GHz
In the previous section, we produced the 8x7 tapered array antenna. Using a
tapered schematic, die side lobe of E-plane radiation pattern has been reduced. Before
fabrication o f the 8x7 tapered array antenna, we fabricated the 8x4 array antenna placing
both the CPW and the feeding network section on the bottom. We proved that 8x4 array
antennas with top feeding and bottom feeding are in agreement. The following section
illustrate the simulated and measured radiation patterns.
Figure 5.15 shows the measured and simulated H-plane radiation patterns o f this
case, and Figure 5.16 shows that o f E-plane. The simulated and measured E- and Hplane radiation patterns are in agreement. Figure 5.17 through 5.21 show the radiation
patterns with the phase shift. In all cases, the measured data is normalized with respect
to the peak value o f the 0 ° phase shift (Figure 5.15). Figure 5.17 shows the simulated
and measured H-plane radiation patterns with dielectric slabs on a CPW. The phase
shifts from left to right are -157.5°, -135°, -112.5°, -90°, -67.5°, -45°, -22.5°, and 0°
which correspond to the 4-bit patterns o f (1110, 0110, 1010, 0010, 1100, 0100, 1000,
0000). The beam scan angle (peak) is at -7.5 0 (352.5 °) for this configuration. Figure
5.18 shows the simulated and measured H-plane radiation patterns with dielectric slabs
on the CPW. These phase shifts from left to right are -315°, -270°, -225°, -180°, -135°,
-90°, -45°, and 0 ° which correspond to the 4-bit patterns o f (0111, 0011, 0101, 0001,
0110, 0010, 0100, 0000). The beam scan angle (peak) is at -15 ° (345°) for this
configuration. Figure 5.19 shows the simulated and measured H-plane radiation patterns
with dielectric slabs on a CPW. The phase shifts from left to right are -270 ° (-630 °), 180° (-540°), -90° (-450°), 0 ° (-360°), -270°, -180°, -90°, and 0° which correspond to
the 4-bit patterns of (0011, 0001, 0010, 0000, 0011, 0001, 0010, 0000). The beam scan
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94
angle (peak) is at -30° (330°) for this configuration. Figure 5.20 shows the simulated
and measured H-plane radiation patterns with dielectric slabs on the CPW. These phase
shifts from left to right are -225 ° (-945 °), -90° (-810°), -315°(-676°), -180° (-540°), 45 0 (-405 °), -270°,-135 °, and 0 ° which correspond to the 4-bit patterns of (0101, 0010,
0111, 0001, 0100, 0011, 0110, 0000). The beam scan angle (peak) is at -45° (315°) for
this configuration. Figure 5.21 shows the maximum beam scan angle o f -52.5 0(307.5 °)
which can be obtained with -157.5 0 phase shifting among elements. From left to right,
the phase shift is -22.5° (-1102.5°), -225° (-945°), -67.5° (-787.5°), -270° (-630°), 1125° (-472.5°), -315.5°,-157.5°, and 0 ° with the corresponding 4-bit pattern o f (1000,
0101, 1100, 0011, 1010, 0111, 1110, 0000).
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(b)
Figure 5.14 (a) Top view o f a 8x4 array antenna without dielectric slabs, (b) Bottom
view o f a 8x4 array antenna without dielectric slabs.
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96
0°
-30‘
-90'
-2 0 dB
-lOdB
H-Plane Measured
120'
OdB
120*
H-Plane Simulated
150'
-150"
±
180°
Figure 5.15 Simulated and measured H-plane Radiation Patterns o f a 8x4 array antenna:
No phase shift case. Slab positions are (0000,0000,0000,0000,0000,0000,0000,0000).
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97
0°
-30'
^
..............
-2 0 dB -lOdB
E -Plane Measured
120'
la r
OdB
120'
E -Plane Simulated
150'
-150'
Figure 5.16 Simulated and measured E-plane Radiation Patterns of a 8x4 array antenna:
No phase shift case. Slab positions are (0000,0000,0000,0000, 0000,0000,0000,0000).
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98
0°
-30'
>-60'
■■■■*............ :.............-j-90'
-20dB
120'
-lOdB
OdB
■120‘
H-Plane M easured
H -Plane Simulated
-150'
±180°
Figure 5.17 Simulated and measured H-plane Radiation Patterns o f a 8x4 array antenna:
The phase shift among elements is -22.5° which corresponds to the beam angle o f 7.5°.
Slab positions are (1110, 0110,1010, 0010, 1100, 0100,1000, 0000).
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99
0°
-30'
v-601
-90'
-2 0 dB -lOdB OdB
H -Plane M easured
120"
120
'
H-Plane Simulated
150'
-150'
Figure 5.18 Simulated and measured H-plane Radiation Patterns o f a 8x4 array antenna:
The phase shift among elements is -45° which corresponds to the beam angle of 15°.
Slab positions are (0111, 0011, 0101,0001, 0110, 0010,0100, 0000).
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100
0
°
-30'
........
i-90'
-2 0 dB -lO d B
120'
OdB
120'
H-Plane M easured
H-Plane Simulated
150'
-150'
Figure 5.19 Simulated and measured H-plane Radiation Patterns o f a 8x4 array antenna:
The phase shift among elements is -90° which corresponds to the beam angle of 30°.
Slab positions are (0011, 0001, 0010, 0000, 0011, 0001, 0010, 0000).
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101
0
°
— ■!...............:.......... i-90'
-2 0 dB
-lO d B
H-Plane Measured
120'
OdB
120'
H-Plane Simulated
150'
-150'
Figure 5.20 Simulated and measured H-plane Radiation Patterns o f a 8x4 array antenna:
The phase shift among elements is -135° which corresponds to the beam angle o f 45°.
Slab positions are (0101, 0010, 0111, 0001, 0100, 0011, 0110, 0000).
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102
0°
-30"
,-60'
1-90'
-20dB -lOdB
OdB
H -Plane Measured
120'
H -Plane Simulated
150'
-150'
Figure 5.21 Simulated and measured H-plane Radiation Patterns of a 8x4 array antenna:
The phase shift among elements is -157.5° which corresponds to the beam angle o f
52.5°. Slab positions are (1000, 0101, 1100, 0011, 1010, 0111, 1110, 0000).
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103
5.4. A Mechanically Steerable Array Antenna Using Controllable
Dielectric Phase Shifters fo r 77 GHz Automotive Radar System.
A mechanically steerable antenna using an adjustable dielectric phase shifter is
designed and developed for automotive radar applications [64]-[66]. The required
dielectric constant of sr = 5.6 can be designed using a ceramic composite material.
Numerical simulations using Ansoft HFSS and Designer are conducted at 77GHz [67].
The microstrip transmission line to a WR-12 waveguide transition as the initial feeding
point has also been investigated [68].
Automobile collision avoidance radar has become a recent feature in automotive
design. In order for automotive radar to be most effective, a combination of mediumrange detection (in front of the car) and short-range detection (in all directions) is
required [69], The 77 GHz collision avoidance radar in an automobile requires a beam
scanning mechanism. One o f the advantages o f this newly proposed phase shifter design
is that the whole antenna can be designed without using solid state phase shifters or
MEMS devices. In this Chapter, we propose that a movable dielectric slab added to a
coplanar waveguide (CPW) can be used as a phase shifter. The radiation patterns of 3x8,
5x8, and 7x8 steerable array antennas with dielectric phase shifter were introduced at 77
GHz [70]. The preset delay was added to a scan beam from -20° to +20°. We have not
fabricated the actual antenna therefore there are no measured results on this dissertation.
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104
Patch
Antenna
Phase
Shifter
Preset
Delay Lines
(a)
(b)
Figure 5.22 (a) Block diagram o f a 3x8 transmitting array antenna and (b) a 7x8
receiving array antenna.
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105
Figure 5.22 shows the transmitter antenna and receiver antenna, respectively. The
antenna as a whole is composed o f a patch antenna, a CPW (with or without a phase
shifter), and preset delay lines. The element separations of the transmitter antenna and
receiver antenna were set to 2.92 mm and 5.84 mm which corresponded to 0.75 A and
1.5A, while the input impedance was 100Q . The signal trace width o f CPW was 0.5
mm and the ground width o f CPW was 1.5 mm. The gap between the ground and the
signal is 0.25 mm. The substrate has esuh = 2.2 (Duroid 5880) and substrate thickness is
0.254 mm.
The required dielectric constant of the dielectric material is sr =5.6. A
ceramic composite material such as Duroid 6006 (er = 6 .l) is suitable as the dielectric
material with some adjustment. This is because when a dielectric slab is added to a CPW,
it may produce a small residual gap (o f less than 1 0 ^ ) between the dielectric slabs and
conductors.
To minimize the reflection occurring when the dielectric material is
added to the CPW, we set the length o f the modified section to be At 2 (or mAt 2 where
m is an integer).
The length o f the dielectric material onto the CPW is 1.82 mm which
corresponded to 180°. Similarly, the lengths o f the dielectric materials on the CPW are
3.65 mm, 7.3 mm, and 10.95 mm which correspond to 360°, 720°, and 1080°,
respectively. In addition, preset delay lines are added to minimize the number of
dielectric slabs. The preset delay lines at Figure 5.22 (a) are set to 0°, -90°, and 0°. The
preset delay lines at Figure 5.22(b) are set to 0°,-180°,-360°,-540°,-360°,-180°, and 0°.
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106
i
t
i
t
r¥irTirTTT¥TinriinrTrrnT^rTirTTW i^
Substrate (
X
er = 2.2
)
t
DMMHaalab
I
Jll llll II 111 II lU HI innrrTTTrrrrrTinnn
Substrate ( $ =22 )
t
X
Substrate ( $=22 )
Figure 5.23 Utilization o f dielectric slabs on a CPW.
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107
Hill
P
E
E
o
10mm
(b)
(a)
10mm
(c)
Figure 5.24 Layout o f the whole antenna. (a)3x8 transmitting array antenna (b)5x8
receiving array antenna (c)7x8 receiving array antenna.
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108
Figure 5.23 illustrates how to utilize the dielectric slabs on a CPW. In order to scan
the beam pattern, the dielectric slabs are inserted into the CPW. The effective dielectric
constant and the characteristic impedance are calculated as a function o f the movable
dielectric slab position on the CPW transmission line. Figure 5.23 shows the left and
right dielectric slabs, based on the center transmission line. They alternate and are
repeatedly inserted into the CPW. Depending on the way the dielectric slab is facing, the
beam angle is scanned from -20 degrees to +20 degrees. Figure 5.24 illustrates the
layout o f the whole antenna. The width o f the patch was 1.155mm and the height was
1.27 mm. The width o f the transmission is 0.5 mm and the spacing between the patches
is 1.4 mm. The bottom layer o f the CPW section does not have a ground plane and via is
used for connecting to the ground. In order to distribute power equally, 3-way, 5-way,
and 7-way power dividers are optimized. In addition, for the initial feeding position, a
microstrip line to the WR-12 waveguide transition is used. Figure 5.25 through 5.27
illustrate the radiation patterns when the dielectric slabs alternate and are repeatedly
inserted into the CPW. The expected beam scan angle is from -20° to +20°. Figure 5.28
shows the radiation patterns o f both the transmitted and the received combined response.
The large side lobe is a result o f the grating lobe in the receiving antenna.
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109
•180
(a)
•180
(b)
0
180
(C)
Figure 5.25 Radiation patterns of a 3x8 array antenna, (a) Radiation patterns occurring
when there are no delays lines and no phase shifters, (b) Radiation patterns occurring
when the dielectric slab (180°) is only inserted into the left CPW. (c) Radiation patterns
occurring when the dielectric slab (180°) is only inserted into the right CPW.
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110
-180
(a)
o
(b)
(c)
Figure 5.26 Radiation patterns of a 5x8 array antenna, (a) Radiation patterns occurring
when there are no delays lines and no phase shifters, (b) Radiation patterns occurring
when the dielectric slab (360°and 720°) is only inserted into the left CPW. (c) Radiation
patterns occurring when the dielectric slab (360° and 720°) is only inserted into the right
CPW.
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Ill
•ISO
(a)
•sol
-tso
(b)
0
-ISO
(C)
Figure 5.27 Radiation patterns of a 7x8 array antenna, (a) Radiation patterns occurring
when there are no delays lines and no phase shifters, (b) Radiation patterns occurring
when the dielectric slab (360°, 720° and 1080°) is only inserted into the left CPW (c)
Radiation patterns occurring when the dielectric slab (360°, 720° and 1080°) is only
inserted into the right CPW.
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112
cr
.-30*
30*
30*
-3 0 *
60*
90*
+90*
■90* 9 0 *
20dB
20dB -lOdB OdB
190*
-1 5 0 *
lOdB OdB
-150*
15 0*
± 180*
(a)
o*
(b)
o*
-3 0 *
.-3 0 *
60*.
90*
.............
20dB
-lOdB
20dB -lOdB
OdB
120*
150*
-ISO*
(C )
+90*
OdB
120*
ISO*
-ISO*
(d)
Figure 5.28 Combined radiation patterns, (a) Combined radiation patterns with a 3x8
transmitting array antenna (Figure 5.25(b)) and a 5x8 receiving array antenna (Figure
5.26(b)). (b) Combined radiation patterns with a 3x8 transmitting array antenna (Figure
5.25(c)) and a 5x8 receiving antenna (Figure 5.26 (c)). (c) Combined radiation patterns
with a 3x8 transmitting array antenna (Figure 5.25(b)) and a 7x8 receiving array antenna
(Figure 5.27(b)). (d) Combined radiation patterns with a 3x8 transmitting array antenna
(Figure 5.25(c)) and a 7x8 receiving array antenna (Figure 5.27(c)).
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113
Chapter 6
Conclusions and Future Work
We provided a novel design of a steerable array antenna using movable
dielectric slabs on a CPW. The phase shifter was based on a movable dielectric slab
placed close to a CPW. The effective dielectric constant and the characteristic
impedance can be calculated as a function of slab height. Using the proposed phase
shifter, the antenna beam angle can be maximally scanned from -52.5° to +52.5°.
In Chapter 2, we proposed a new concept o f microwave dielectric slab phase
shifters on a CPW. Using this phase shifter, we can minimize the reflection with or
without dielectric material (O n-O ff positions) at the designed frequency. The measured
phase o f S21 with the vector network analyzer (VNW A) at seven phase shift positions
created by the 3-bit dielectric slab combinations has been illustrated. The measured
phase was close to that o f the eight phase conditions (0, -45°, -90°, -135°, -180°, -225°, 270°, -315°). The reflection in all cases was small at 6 GHz.
A design o f a steerable array antenna at 5.8 GHz for a wireless network has
been demonstrated in Chapter 3. The phase shifter was based on a movable dielectric
slab placed close to a CPW with airgaps. The dielectric slabs on a CPW continuously
moved from d = 0 to d = 5 mm. The impedance mismatch can be avoided by choosing
the slab dielectric constant and length for the two extreme positions, namely at d = 0 and
d > 2 mm. It may be possible to further reduce |Sn| by optimizing the impedance
mismatching position at d > 0 instead o f d = 0. We also showed that the use of preset
delay lines can reduce the number o f phase shifters. Although the test antenna was
designed for 5.8 GHz, the technique can be applied at a much higher frequency. The
measured H-plane radiation patterns verified that the beam angle can be scanned from
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114
-30° to + 300 . This antenna is suitable for applicable in a cordless phone or W LAN
( Wireless Local Area Network).
The purpose o f Chapter 4 was to show the feasibility o f designing a simple
steerable antenna at 20 GHz. Using extensive numerical simulations, we obtained the
desired structure and material to create a 3-bit phase sifter fabricated on an IS640
substrate. The proposed antenna was suited for scanning in one direction such as in an
indoor/outdoor wireless system. For a satellite tracking antenna which requires a 2-D
scan, the rotational motion must be provided by a motor-driven turntable. Although
further design work is needed to make a practical system, we have demonstrated that our
idea is feasible, and we should be able design a low-cost beam steerable antenna at
M M W frequencies. We believe the proposed method is particularly suited for high
M M W frequency array antennas for which the solid state-based or MEMS-based phase
shifter is difficult and expensive to design [71], [72]. This 20 GHz antenna can be
applicable for military satellite uplinks.
We designed and fabricated beam steering antennas based on mechanically
controllable microwave phase shifters at 24 GHz in Chapter 5. Using the extensive
numerical simulations and experience of the previous section, we obtained the desired
structure and material to create a 4-bit phase sifter fabricated on an IS640 substrate. In
Chapter 4, we tested that the performance o f a new antenna with a phase shifter.
However, the CPW section unexpectedly radiated, causing the radiation pattern o f the Eplane to be slightly distorted. The large sidelobe in the E-plane may be due to the
interference from the CPW and the feeding network. In order to reduce the large
sidelobe in the E-plane, we placed both the CPW and the feeding network section on the
bottom side. The measured radiation pattern illustrated that the sidelobe o f the E-plane
was much smaller than that of the E-plane in Chapter 4. One of the advantages to this
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115
newly proposed phase shifter design was that the whole antenna can be designed
without using any solid state phase shifters or M EM S devices. By using a 4-bit phase
shifter, the antenna beam angle can be scanned from -52.5° to +52.5°. This 24 GHz
antenna can be used in new commercial applications such as W LAN and automotive
collision avoidance radar systems. In addition, we proposed an array antenna with phase
shifters that can be suited for applications in an automotive radar system at 77 GHz. As
the dielectric slab was inserted into the CPW in an alternating and repetitive fashion, the
beam angle was able to be scanned from -20° to +20°. Future steps involve the design
and testing o f a microstrip line for a waveguide transition. The waveguide transition
would be on a single layered dielectric substrate, as the initial position.
We investigated the steerable array antenna with movable dielectric slabs on a
CPW. Using this technique, we can design and fabricate much higher frequency range
antennas such as 60 GHz, 77 GHz, 95 GHz, etc. Recently, 7 GHz o f unlicensed
bandwidth around 60 GHz opened allowing for a variety of applications. Some of these
include Gb/s point-to-point links, wireless local area networks with extraordinary
capacity, and vehicular radar at nearby frequencies. Presently, the exploitation of this
band is minimal because o f the high cost o f the compound semi-conductor technology
needed to process the mm-wave signals. For these reasons, there is much potential for
my research for investigating operations at 60 GHz. We w ill continue to conduct
research to fabricate the antenna and to make comparisons with simulations and
experiment data.
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116
Bibliography
[1] R. C. Hansen, Phased Array Antennas, W iley Interscience, 1998
[2] R. J. Mailoux, Phased Array Antenna Hanbook, Artech House, 1994
[3] R. C. Johnson and H. Jasik, Antenna Engineering Handbook, 1984
[4] Y. T. Lo and S. W. Lee, Antenna Handbook: Theory, Applications, and Design,
1988
[5] R. S. Elliott, Antenna Theory and Design, W iley Interscience, 2003
[6] G. Cortes-Medellin and P. F. Goldsmith, “Analysis o f active surface reflector
antenna for a large millimeter wave radio telescope,” IEEE Trans. Antennas Propag.,
vol. 42, no. 2, pp. 176-183, Feb. 1994
[7] T. K, Sakar, M . C. Wicks, M . Salazar-Palma, and R. J. Bonneau, Smart Antennas,
Wiley Interscience, 2003.
[8] D. M . Pozar, Microwave and RF Design o f Wireless Systems, Wiley, 2001.
[9] Y. Kuga, J. Cha, J. A. Ritcey, and T. Kajiya, “Mechanically steerable antennas using
dielectric phase shifters,” in IEEE Antennas Propag. Soc. Int Symp. Dig., vol. 1.
Monterey, CA, June 20-24,2004, pp. 161-164.
[10]
E. Perret, H. Aubert, and H. Legay, “Scale-changing technique for the
electromagnetic modeling o f MEMS-controlled planar phase shifter,” IEEE Trans.
Microw. Theory Tech., vol. 54, no. 9, pp. 3594- 3601, Sept. 2006.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
117
[11]
F. D. Flaviss, N. G. Alexopoulos, and O. M . Stafsudd, “ Planar microwave
integrated phase-shifter design with high purity ferroelectric material,” IEEE Trans.
Microw. Theory Tech., vol. 45, no. 6, pp. 963-969, June. 1997.
[12]
J. S. Hayden and G. M . Rebeiz, “Very low-loss distributed X-band and Ka-band
MEMS phase shifters using metal-air-metal capacitors,” IEEE Trans. Microw.
Theory Tech., vol. 51, no. 1, pp. 309-314, Jan. 2003.
[13]
B. Acikel, T. R. Raylor, P. J. Hansen, J. S. Speck, and R. A. York, “A new high
performance phase shifter using BaSrTiOs thin films,” IEEE Microw. Wireless
Compon. Lett., vol. 12, no. 7, pp. 237- 239, July 2002.
[14]
S. Barker and G. M . Rebeiz, “Distributed MEMS true-time delay phase shifters
and wide-band switches,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 11, pp.
1881 - 1889, Nov. 1998.
[15]
J. B. L. Rao, D. P. Patel, and V . Krichevsky, “ Voltage-controlled ferroelectric
lens phased arrays,” IEEE Trans. Antennas Propag., vol. 47, no. 3, pp. 458 -468,
Mar. 1999.
[16]
T. Y. Yun and K. Chang, “A low-loss time-delay phase shifter controlled by
piezoelectric transducer to perturb microstrip line,” IEEE Microw. Guided Wave
Lett., vol. 10, no. 3, pp. 96-98, Mar. 2000.
[17]
S. G. Kim, T. Y. Yun, and K. Chang, “Time-delay phase shifter controlled by
piezoelectric transducer on coplanar waveguide,” IEEE Microw. Wireless Compon.
Lett., vol. 13, no. 1, pp. 19-20, January 2003.
[18]
P. Cheung, D. P. Neikirk, T. Itoh, “Optically controlled coplanar waveguide
phase shifters,” IEEE Trans. Microw. Theory Tech., vol. 38, no. 5, pp. 586-595, May
1990.
[19]
A. M . Vaucher, C. D. Striffler, and C. H. Lee, “Theory o f optically controlled
millimeter-waveguides,” IEEE J. Quantum Electron., vol. QE-16, pp. 277-288, Mar.
1980
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
118
[20]
J. Cha, Y. Kuga, and S. Lee,” A continuously steerable array antenna using
movable dielectric slabs on a coplanar waveguide,” Microw. Opt. Technol. Lett., vol.
48, no. 11, pp. 2222-2227, November 2006.
[21]
J. Cha, and Y. Kuga, “A phased-array antenna with mechanically controllable
microwave phase shifter,” URSI 2005 University of Colorado, Jan. 5-8 2005,
Boulder, CO.
[22]
J. Baker-Jarvis and E. J. Vanzura, “Improved technique for determining
complex permittivity with the transmission/reflection method,” IEEE Trans. Microw.
Theory Tech., vol. 38, no. 8, pp. 1096-1101, Aug. 1990.
[23]
Y. Kuga, S. W. Lee, M . Taya, A. Almajid, S. Lee, J. F. Li, and R. Watanabe,
“Experimental and numerical studies o f dielectric properties o f BaTiOs-platinum
composites at microwave frequencies,” IEEE Trans. Dielectr. Electr. Insul., vol. 12,
no. 4, pp. 827-834, Aug. 2005,
[24]
J. Cha, Y. Kuga, and T. Kajiya, “Mechanically steerable antennas with
controllable microwave phase shifters at 20 Ghz,” IEEE Antennas Propag. Soc. Int
Symp. Dig., vol. 1, Washington D.C., July 03-06, 2005, pp. 691-694.
[25]
C. P. Wen, “ Coplanar Waveguide: A Surface Strip Transmission Line Suitable
for Non-Reciprocal Gyromagnetic Device Application,” IEEE Trans., vol. MTT-17,
pp. 1087-1090,1969.
[26]
J. Helszajn, Microwave planar passive circuits and filters, Wiley, 1994
[27]
S. G. Pintzos, “Full-wave spectral domain analysis o f coplanar strips,” IEEE
Trans., vol. MTT-39, pp. 239-246,1991.
[28]
Y. Liu, K. Cha, T. Itoh, “Non-leaky coplanar (NLC) waveguides with conductor
backing,” IEEE Trans. Microw. Theory Tech., vol. 43, no. 5, pp. 1067-1072, May.
1995.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
119
[29]
K. C. Gupta, R. Garg, I. Bahl, and P. Bhartia, Microstrip Lines and Slotlines,
Artech House, 1996.
[30]
A. K. Bhattacharyya, Electromagnetic fields in multilayered structures theory
and applications, Artech House, 1994.
[31]
G. Ghione and C. U. Naldi, “Coplanar waveguide for M M IC applications:
effect of upper shielding, conductor backing, finite-extent ground planess, and lineto-line coupling”, IEEE Trans. Microw. Theory Tech., vol. 35, no. 3, pp. 260-267,
March 1987.
[32]
C. Veyres and V. F. Hanna, “Extension o f the application of conformal mapping
techniques to coplanar lines with finite dimensions,” Int. J. Electron., vol. 48, pp. 4756,1980
[33]
S. S. Bedair and I. Wolff, “Fast, accurate and simple approximate analytic
formulas for calculating the parameters o f supported coplanar waveguides for
(M )M IC ’s,” IEEE Trans. Microw. Theory Tech., vol. 40, no. 1, pp. 41-48, Jan. 1992.
[34]
S. Lee and Y. Kuga, “ Complex dielectric constant measurements and model
comparison of lossy dielectric materials,” 2000 IEEE AP-S International Symposium,
July 16-21 2000, Salt Lake City, UT.
[35]
L. Tsang, J. Cha, and J. Thomas, “Electric fields of spatial Green’s functions of
microstrip structures and applications to the calculations o f impedance matrix
elements, Microw. Opt. Technol. Lett., vol. 20, no. 2, pp. 90-97, Jan. 1999.
[36]
D. H. Schaubert, D. M . Pozar, and A. Adrian, “ Effect of microstrip antenna
substrate thickness and permittivity: comparison of theories with expreriment,”
IEEE Trans. Antennas Propag., vol. 37, no. 6, pp. 677-682, Jun. 1989.
[37]
L. Tsang, J. Cha, C. C. Huang, and C. Chan, “Surface electric fields o f multi­
layered medium green’s functions and the calculation of impedance matrix elements
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
120
of microstrip structures,” IEE Proc.-Microw. Antennas Propa., vol. 147, no. 3. pp.
179-186, Jun. 2000.
[38]
H. F. Lee and W . Chen, Advances in microstrip and printed antennas, W iley
Interscience, 1997.
[39]
W . L. Stutzman and G. A. Thiele, Antenna Theory and Design, John Wiley &
Sons, 1998.
[40]
K. R. Carver and J. W. Mink, “Microstrip Antenna Technology,” IEEE Trans.
Antennas Propag., vol. AP-29, pp. 2-24, Jan. 1981.
[41]
F. Gardiol, Microstrip circuits, W iley Interscience, 1994
[42]
C. Kizitltas, D. Psyshoudakis, J. L. Volakis, and N. Kikuchi,” Topology Design
Optimization o f Dielectric Substrates for Bandwidth Improvement o f a Patch
Antenna”, IEEE Trans. Antennas and Propagations, vol. 51, no. 10, pp. 2732-2743,
Oct. 2003.
[43]
E. Levine, G. Malamud, S. Shtrikman, and D. Treves, “A study of microtrip
array antenna with the feed network,” IEEE Trans. Antennas Propag., vol. 4, no. 4,
pp. 426 -434, Apr. 1989.
[44]
C. F. Wang, F. Ling, and J. M . Jin, “A fast full-wave analysis o f scattering and
radiation from large finite arrays o f microstrip antennas,” IEEE Trans. Antennas
Propag., vol. 46, no. 10, pp. 1467 -1474, Oct. 1998.
[45]
N. Yuan, T. S. Yeo, X . C. Nie, and L. W. Li, “A fast analysis o f scattering and
radiation of large microstrip antenna arrays,” IEEE
Trans. Antennas and
Propagations, vol. 51, no. 9, pp. 2218-2226, Sept. 2003.
[46]
K. L. Wu, M . Spenuk, J. Litva, and D. G. Fang, “ Theoretical and experimental
study o f feed network effects on the radiation pattern of series-fed microstrip
antenna arrays, IEE Proceedings-H 138, no. 3,1991
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
121
[47]
B. B. Jones, F. Y. M . Chow, and A. W . Seeto, “The synthesis o f shaped patterns
with series-fed microstrip patch arrays,” IEEE Trans. Antennas Propag., vol. 30, no.
6, pp. 1206-1212, Nov. 1982.
[48]
G. Maral and M . Bousquet, Satellite Communications Systems, W iley, 2002.
[49]
T. Pratt, C. Bostiam, and J. Allnutt, Satellite Communications, Wiley, 2003.
[50]
B. R. Elbert, Satellite Communication Applications Handbook, Artech House,
2004.
[51]
B. A. Munk, Finite Antenna Arrays and FSS, John W iley & Sons, Inc, 2003.
[52]
T. Metzler, “Microstrip series arrays,” IEEE Trans. Antennas Propag., vol. 29,
no. 1, pp. 174 -178, Jan. 1981.
[53]
J. Cha and Y . Kuga, “A mechanically steerable phased-array antenna with
controllable microwave phase shifter,” IEEE NETC, May 17-19, 2005, Bellevue,
WA
[54]
R. R. James and P. S. Hall, Handbook of Microstrip Antennas, IEE-Series, Peter
Peregrinus Ltd., 1989.
[55]
C. A. Balanis, Antenna Theory, Wiley, 1997.
[56]
J. Cha and Y. Kuag, “ A steerable array antenna using mechanically
controllable 4-bit dielectric slab phase shifter on a coplanar waveguide at 24 GHz,”
submitted to IEEE Microw. Wireless Compon. Lett., 2006.
[57]
J. Schoebel, T. Buck, M . Reimann, M . Ulm, M . Schneider, A. Jourdain, G. J.
Carchon, and H. A. C. Tilmans, “ Design considerations and technology assessment
of phased-array antenna systems with RF MEMS for automotive radar applications,”
IEEE Trans. Microw. Theory Tech., vol. 53, no. 6, pp. 1968-1975, June 2005.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
122
[58]
H. Iizuka, K. Sakakibara, T. Watanabe, K. Sato, K. Nishikawa, “Millimeter-
wave microstrip array antenna with high efficiency for automotive radar sytems,”
R&D Review o f Toyota CRDL vol. 37. no.2 pp. 7-12.
[59]
A. Kawakubo, S. Tokoro, Y. Yamada, K. Kuroda, and T. Kawasaki,
“Electronically scanning millimeter-wave radar for forward objects detection,” Soc.
Automotive Eng., Warrendale, PA, SAE Tech. Paper 2004-01-1122,2004.
[60]
J. Freese, H. L. Blocher, J. Wenger, and R. Jakoby, “Microstrip patch arrays for
a millimeter-wave near range radar sensor,” in Proc. German Radar Symp., Berlin,
Germany, Oct. 11-12,2000, pp. 149-153
[61]
J. Cha, Y. Kuga, A. Ishimaru, and S. Lee, “ A 20 GHz steerable array antenna
using 3-bit dielectric slab phase shifters on a coplanar waveguide,” accepted for
publication in IEEE Trans. Antennas Propag., 2006.
[62]
A. Safaai-Jazi, “A new formulation for the design o f chebyshev arrays,” IEEE
Trans. Antennas Propag., vol. 42, no. 3, pp. 439 -443, Mar. 1994.
[63]
A. T. Villenevue, “Taylor patterns for discrete arrays,” IEEE Trans. Antennas
Propag., vol. 32, no. 10, pp. 1089 -1093, Oct. 1984.
[64]
A. G. Demeryd, “Linearly polarized microstrip antennas,” IEEE Trans.
Antennas Propag., vol. 24, pp. 846 -851, Nov. 1976.
[65]
R. A. Sainati, “CAD o f microstrip antennas for wireless applications,” Artech
House, Boston, 1996
[66]
J. T. Aberle, D. M . Pozar, and C. R. Birtcher, “Evaluation o f input impedance
and radar cross section of probe-fed microstrip patch elements using an accurate
feed model, IEEE Trans. Antennas Propag., vol. 39, pp. 1691 -1696, Dec. 1991.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
123
[67]
J. Cha and Y. Kuga, “A mechanically steerable array antenna using
controllable dielectric phase shifters for 77 GHz automotive radar sytems,” 2006
IEEE AP-S international symposium, July 9-14, Albuquerque, N M .
[68]
H. Iizuka, T. Watanabe, K. Sato, and K. Nishikawa, “Millimeter-wave
microstrip line to waveguide transition fabricated on a single layer dielectric
substrate,” R&D Review o f Toyota CRDL vol. 37. no.2 pp. 13-18.
[69]
T. Yuan, J. Y. Li, L. Zhang, and M . S. Leong, “A novel series-fed taper antenna
array design and analysis,” APMC2005 Proceedings,
[70]
J. Cha and Y . Kuga, “Steerable array antenna with dielectric phase shifters for
77 GHz automotive radar applications,” URSI 2006 University o f Colorado, 4-7 Jan.
2006, Boulder, CO.
[71]
G. M . Rebeiz and J. B. Muldavin, “RF M EM S switches and switch circuits,”
IEEE Microw. Mag., pp. 59-71, Dec. 2001.
[72]
A. Jourdain, X . Rottemberg, G. Carchon, and H. A. C. Tilmans, “ Optimization
of 0-level packaging for RF-MEMS devices,” in 12th Int. SolidState Sensors,
Actuators, and Microsystems Conf, Boston, M A , Jun. 9-12, 2003, pp. 1915-1918.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
Appendix A: Measurement Processing
Physical Design
Guideline
CPW simulation
Array Antenna Simulation
Layout Design
Negative Image Etching
Measurements
Figure AppAl Block diagram o f a measurement processing.
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125
Figure AppAl shows the block diagram of measurement processing. Initially, the
physical design should be considered. We derived the dielectric constant of the substrate
and the operating frequency, based on the applications. After choosing the substrate and
frequency, we investigated the CPW. Using Ansoft HFSS simulation, we found the
width o f the signal line, the gap size between signal and the ground lines, and the input
impedance. The array antenna size, spacing between edges, and the radiation patterns
were calculated using Ansoft Designer. The antenna layout was designed on Visio. With
the Visio file, we could then fabricate an antenna using the negative imaging etching
method. This job was completed in the Physics shop in the Department o f Physics. The
fabricated antenna was needed to measure the return loss using a VNW A (Vector
Newtork Analyzer) as well as far-field radiation patterns in an anechoic chamber.
Fig AppA2 illustrates the antenna setup for the far-field radiation patterns. In order to
perform the measurement, place the A U T on a pedestal then secure it with tape. Make
sure the antenna boresight is facing the source antenna (initial 0 deg position). The A UT
should not move when the stage is rotated. Connect the A UT to the coaxial cable which
is connected to an SA (Spectrum Analyzer). Turn ON the signal generator and spectrum
analyzer. Set the signal source to the desired frequency. Set the spectrum analyzer center
frequency to the desired frequency and span to 100 M Hz. Find the signal. This must be
at least 20 dB above the noise level. Turn ON the stepping motor controller and PC. If
you need to adjust the antenna position (rotational angle), you can do so using a CW and
CCW switch on the motor controller. Finally, run the measurement program on PC.
Figure AppA3 shows the photo of the anechoic chamber.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
Signal
Generator
Source Ant.
AUT
SA
HPIB
PC
DAS
Pedestal
SM
controller
Rotational Stage
Figure AppA2 Far-field radiation pattern measurements setup.
Figure AppA3 Photo of the anechoic chamber
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127
Appendix B: HFSS and Designer Simulations
Figure AppBl 3D model setup of a CPW with the dielectric slab in HFSS
Figure AppBl illustrates the HFSS simulation setup. The port size is very integral to
obtaining the exact solution. Mostly, the height o f the port size must be at least 5 times
higher tan the thickness o f the substrate. In addition, the width size of the port should be
much wider than 5 times the width o f the signal line. Ansoft HFSS (High Frequency
Structure Simulator) is a 3D electromagnetic-Field simulation for high-performance
electronic design. HFSS is the industry-standard software for S-parameter and fullwave
SPICE extraction and for the electromagnetic simulation of high-frequency and high­
speed components. HFSS is widely used for the design o f con-chip embedded passives,
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
PCB interconnects, antennas, RF/microwave components, and high-frequency IC
packages. HFSS is based on the finite element method (FEM).
J
Figure AppB2 2D model setup of a Designer with 4x4 array antenna.
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129
Figure AppB2 shows the 2D model setup o f an Ansoft Designer. This layout
includes the array antenna part and feed network section. Ansoft Designer provides an
integrated schematic and front-end design management for complex analog, RF and
mixed-signal applications. By leveraging advanced electromagnetic field simulators
dynamically, they can be linked to powerful circuit and system simulations. Ansoft
Designer enables engineers to design, optimize, and validate component, circuit, and
system performance long before building a prototype in hardware. Ansoft Designer is
based on Method o f Moment (M O M ) and is suitable for 20D structure simulations.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
130
VITA
Junho Cha was bom July 18, 1969, in Seoul, Korea. He received his B.S. degree in
Electronics Engineering from Kwangwoon University, Seoul, Korea in 1996. He
completed both his M.S. and Ph. D. degree in Electrical Engineering from University o f
Washington, Seattle, WA, USA in 1998 and 2006, respectively. The title o f his Master’s
thesis was “Radiation from Microstrip Lines”. While pursuing his Ph. D, Junho was a
research assistant at the Electromagnetics and Remote Sensing Laboratory at the
University o f Washington and completed his doctoral research under the guidance of
Professor Yasuo Kuga. Additionally, from 2000-01, he completed an internship at the
Intel Corporation. Junho continues to work as a Research Associate in the Department of
Electrical Engineering at the University o f Washington. His research interests include of
microwave and millimeter-wave array antennas, high frequency devices and materials,
and numerical and experimental electromagnetics.
List of publications
Journal papers
• Junho Cha. and Yasuo Kuga, “A Steerable Phased-Array Antenna Using
Mechanically Controllable 4-bit Dielectric Slab Phase Shifter on a Coplanar
Waveguide at 24 GHz,” submitted to IEEE Microw. Wireless Compon. Lett., 2006.
• Junho Cha. Yasuo Kuga, Akira Ishimaru, and Sangil Lee “A 20 GHz Steerable
Array Antenna Using 3-bit Dielectric Slab Phase Shifter on a Coplanar Waveguide,”
accepted for publication in IEEE Trans. Antennas Propag., 2006.
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
131
•
Junho Cha. Yasuo Kuga, and Sangil Lee, “A Steerable Array Antenna Using
Movable Dielectric Slabs on a Coplanar Waveguide,” Microwave and Optical
Technology Letter, vol. 48, n o .ll, pp. 2222-2227,2006.
•
L. Tsang, Junho Cha. and C. C. Huang, “Surface Electric Fields o f Multi-layered
Medium Green’s Functions and Calculation o f Impedance Matrix Elements of
Microstrip Structure”, IE E Proc. Microwave, Antennas and Propagation, vol. 147,
no. 3, June 2000, p. 179-186.
•
L. Tsang, Junho Cha. and J. R. Thomas, “Electric fields o f spatial Green’s functions
of Microstrip structures and applications to the calculations o f impedance matrix
elements, ” Microwave and Optical Technology Letters, vol. 20, no. 2, pp. 90-97,
1999.
•
Junho Cha. and N . Y. Kim, “Semiconductor Magnetic Field Sensors”, Journal of
the Korean Institute o f Electrical and Electronic Material Engineers, vol. 9, No. 5,
pp. 512-517, June 1996
Conference papers
•
Junho Cha. and Yasuo Kuga, “A Mechanically Steerable Array Antenna Using
Controllable Dielectric Phase Shifters for 77 GHz Automotive Radar Systems,”
2006 IEEE AP-S international Sysposium, July 9-14, Albuquerque, N M .
•
Junho Cha. and Yasuo Kuga, “Steerable Array Antenna with Dielectric Phase
shifters for 77 GHz Automotive Radar Applications,” URSI 2006 University o f
Colorado, 4-7 Jan. 2006, Boulder, CO.
•
Junho Cha. Yasuo Kuga, and J. T. Kajiya, “A Mechanically Steerable Array
Antenna with Controllable Microwave Phase shifters at 20 GHz, ” 2005 IEEE A P S
international Sysposium, pp. 691-694, July 3-8, Washington DC.
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132
•
Junho Cha and Yasuo Kuga, “A Mechanically Steerable Phased-Array Antenna
with Controllable Microwave Phase Shifter”, IEEE NETC, May 17-19, 2005,
Bellevue, WA.
•
Junho Cha. and Yasuo Kuga, “A Phased-Array Antenna with Mechanically
Controllable Microwave Phase Shifters,” URSI 2005 University o f Colorado, 5-8
Jan. 2005, Boulder, CO.
•
Yasuo Kuga, Junho Cha. James A. Ritcey, and James T. Kajiya, “Mechanically
Steerable Antennas Using Dielectric Phase Shifters,” 2004 IEEE AP-S International
Symposium, pp. 161-164, June 20-26, Monterey, CA.
•
L. Tsang, Junho Cha. and J. Thomas, “Surface Electric Fields o f Spatial Greens’s
Functions o f Microstrip Structures and applications to the Calculations of Impedance
Matrix Elements”, Proceedings o f the Progress in Electromagnetics Research
Symposium, Taipei, Taiwan, March 1999
Reproduced with permission of the copyright owner. Further reproduction prohibited without permission.
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