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JPS5045601

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DESCRIPTION JPS5045601
, Sansui-Electric Co., Ltd. '-■ Japan Patent Office 7 ° published special patent publication ■ JP
5 Q, 45601 1 13462 359,-Specification 1, Name of f 閑 I. fj Ill @ conversion equipment of 閑 2 許
The signals of a plurality of channels obtained from the 1 @ song 111 original sound field of the
talk are converted into 2 channels. The “channel” signal of 2 channels is applied, and a
predetermined addition / subtraction is performed between the 2 channels. A variable matrix
circuit that outputs signals of four channels including two channels in front and two channels in
back, and a phase relationship between the signals of the two channels are detected, and in
accordance with the phase relationship, in a direction in which the polarities mutually expire. A
phase discriminator that outputs two changing signals, and matrix coefficients of the matrix
circuit according to each output signal of the phase discriminator in each channel. The field
effect transistor according to claim 11, wherein the unnecessary variable 11 does not occur, and
the variable resistance element and the variable resistance element have a variable resistance,
and the variable resistance) uses a field effect transistor as a ladder. Using any four of the field
effect transistors formed in the same array as a plurality, the source field of another field effect
transistor in the array is bias voltage of the field effect transistor. A sound 4I signal conversion
device, which is processed to apply an average value signal obtained by combining the output
signals of the phase discriminator to the gate of the source-follower-connected field effect
transistor. (2) The two-channel No. 41 formed by converting signals of a plurality of channels
obtained from the original sound field into two channels is applied, and additions and
subtractions of the measurements are performed between the two-channel signals to obtain two
front and two rear. A variable matrix circuit that outputs four channels of channels and the front
'V, two channels of signal levels are compared between the signals, and the directions of KN are
made different from each other depending on the level difference. 2 Of the comparators that
output the 11th No. and the outputs of the comparators, variable control so that unnecessary
EndPage: 1 signal is not generated in each matrix coefficient of the matrix circuit according to
each signal of the comparators In the sound 4111 signal converter using a variable resistance
transistor, the field effect transistor is Using any one of a plurality of field-effect transistors
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formed in a plurality of meters in the array, the bias voltage of the field-effect transistors is
provided by testing the other field-effect transistors in the play to the source follower A soundtimes 41x recombination device, wherein the output signal of the comparator is combined with
the gate of a source-follower-connected turtle field effect transistor to obtain an average value
signal.
8. Detailed description of the invention In the present invention, t and t are signals of 4 channels
obtained from the original sound field are converted into 2 channels, and 2 channels of signals
are converted to almost original 4 channels of signals. It relates to the improvement of the 5 * 1 *
@ converter. Recently, 9 Toe was obtained from the original sound field 7'l! 4チ、′ヤンネル。
Convert to E t-2 channel signal and convert to ref-4 and 岱 2-1, then rtt- pick up and restore
almost original 4 channel / channel signal 1, and use each speaker as a speaker Destiny Matrix-to
4 channels-Various stereo systems have been developed and used for laughs. A representative
example of such a four channel stereo system is shown in FIG. In the original sound field 1,
microphones MLν, MRν, ML 璽, MRm are disposed at the four corners of the original sound
field 1 and sound is collected, and four channel signals LP, Rp. Each of the signals Ly # My # and
L "* R- is sent to the encoder 2 one by one. In this encoder 2, for example, as shown in FIG. 2, the
resistance matrix circuit 11 has a phase shift characteristic of the phase shift circuit 12.13 with
respect to each of the resistance transformation circuits 12.13 for the special transformation of i!
! It is taken from the phase shift circuit 11615 which has phase characteristics. · Therefore, in the
above encoder 2, “4 channel signals Lv and RνeLm * Rm ′ are encoded in a predetermined
phase relationship to be a 2 channel signal, converted to R, and the converted 2 channel signal ,
R are transmitted by the transmission systems 3 and 4, and are related to the recording medium
5 such as a record board or a tape. Then, a signal of 2 channels and R are picked up from the
recording medium 5 and transmitted to a predetermined htm by the transmission system 6.17. Is
this transmission system 6.7, for example, from broadcast-栂 or distribution medium? It is a
reproduction system etc. The signals L and Rf of 2 channels transmitted by the above-mentioned
transmission system 7 are led to a signal conversion device or decoder 8. For example, as shown
in FIG. 8, this decoder 8 comprises: 1 resistance matrix circuit 16, cherry phase circuits J7 and
18 having the same phase shift characteristic, phase shift circuit 19 having this hole
transformation 1 path JFtJ & the above phase shift circuit 17. Phase shift circuit 18 having 18.
Phase inversion circuit 21.22 ′ ′: like: aLJ ′ ′ I,. Signal LPI 會 RFI HLP 11 mRBIVL after
adjustment to No. 1B No. Ly * RyeLmtK, I.
And, these four channels-signal 1. Loudspeakers SLy, 5) (F), which are respectively arranged in
411ii 1 by rounding Fl, Ryl, LsleRlI with each microphone MLylMRFeMLI * IVIRI of the original
sound field 1 in 1 Nicouto 9. In addition to 8 LB '* SRI, three-dimensional reproduction of a
sound image is performed. In such a four-channel fist stereo system, the functions of the encoder
2 and the decoder 8 'are further detailed in detail in a-1-41! ! I think as shown in l, l. Nahi, in the
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middle of the sea, oh left)), RetD, right 'IIJ, FoB8 + 7 way, B. Shows a sword. マ イ ク ロ フ ォ ン
Earth ° field such as ° 9 右側, right R 6 シ counterclockwise microphone any microphone d φ
、 E φ 0∼ E 0, 0 ~ 0-LE of the mouth U direction The encoder 2 outputs “L”, “Ra”,
“EndPage: 23”. つ と き つ つ つ π つ 2π) Back force, R becomes . After that, the total
(gold) output L-Ra% of the encoder 2 However, F (81): sound source at the angle φ forward
sound source 'B (Sφ): the rear sound source at the angle φ Become. The signal thus encoded, ie,
the output of the decoder 2, is passed through the decoder B. In the direction 5 of an arbitrary
angle θ, as in the case of the above encoding, the output F (8φ) of the decoder 8 is used in the
case of decoding the sound in front of ∼0 to 1t in the same manner as in the encoding. 1. , F,
(8φ), 1 = L sinfl / 2 + Reos # / 2 * e ** e 141 pa 'JP JP 5O-45601 (3), and when π to θ to 2π,
恢 9 rDii Ode: i -'- Ft, 6 sensors, 8 meters, 8 degrees 6 B (8 phi) 11 d, B (S phi) 1. =-J L sin mA J
Reos mA @ · · · · · · (51. Therefore, the total (totally) output F (Sφ) 1 of the decoder 8 and B (8φ)
1 are obtained. Now, in the encoder 2, the sound in the direction of the angle φ of soba is
encoded in the scratch of the second channel, No. 2, H, and this signal is decoded by the decoder
8 in the direction of any angle 角If the characteristic of the case is considered, the output Eθ °
light of the decoder 8, E6 =, Lgin # /! RR cos 峠 '= Eφ sin isin′′04 + Eφ cosf ′ ′ trcoo 14 /
TA = E □. --? 11 '@ 1111 "@ @ @ @;' @ @, (/, 12 degrees, and so on.
If this is shown graphically, as in FIG. 5, the crosstalk in the case where the signal coded in the
direction of φ is decoded in the direction of θ can be clearly seen. In this way, the crosstalk in
the direction of 90 channels 11 to 9 channels is approximately -8 dB, so that the input of
encoder 2 is also made symmetrical (square) with 4 channels and the matrix configuration of de
cog 8 is also made. Symmetric (square. In this case, the output of the corg 2 is R (the matrix
configuration of the encoder) where 0.414 in Δ, and the decoder outputs Ly1 ', * Rv1, Lll. Rla,
(matrices of decoder, configuration), where .DELTA. When the decoder matrix configuration
according to the 9th equation is shown as ク ト ktor 1, as shown in FIG. The left signal indicates
that only ν is input. This cross)-The output of the decoder 8 in the same direction as the input to
the radar 2 is QdB, and the output of the decoder 8 with a 90 ° difference from the input of the
encoder 2 is -8dB 9, reverse direction q It becomes -COdB. However, in the conventional matrix
four-channel stereo system as described above, there is a large gap between the dorm channel
and only about 3 db of separation. That is, as shown in FIG. 8 (al, (b), when the sound source is
present only on the microphone MLν side on the left front side in the example of the original
sound field 1, the reproduction hot water 9 is originally only on the speaker gLν on the front
side of the roof I think that the playback sound should appear, the theory, that the contact
channel 哀:; SRF, SL-also has a sound of Nikko. These extra signals are the main signals of the 70
St. Iku No. Mi 必 s favorite signal. The ratio of the main signal ↑ to the end page: 3 of the と and
End crossnik = issue is 1 to 0.7 and the difference is about 8 dbO. This occurs for each channel,
and cross talk occurs with each other in the direction of the arrow as shown in Xie. The presence
of these micro-stoke subdivisions makes the sound field unnatural and not clean, and sensitizes
the separation Yoshi to create a circle of information about the direction. Therefore, in order to
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falsify such four sets, an 11 r 1 l r book signal converter as shown in FIG. The details will be
described below. In FIG. 1θ, the input terminals 31L and 31R to which the signals of the two
channels and R are applied are respectively connected to the input terminals of the variable
recovery matrix circuit 32.
This Kut-matrix circuit 32 is a 2-channel signal. Signal of the first matrix circuit 33.2 channel
which synthesizes R to obtain two sum polarizations (L ′ ′, n) w − signals of R channel, and R
to obtain a difference signal (L−R) 2 matrix circuit 34, this second matrix variable gain amplifier
35.2 channel 1-No. Eighth matrix circuit 36 for combining R to obtain two difference signals −
(L−a) and (I, −R) having different polarities from each other, z-channel signal, combining R and
its sum signal ( A fourth matrix circuit 37 for obtaining L + R-), a second variable gain multiplier
38 for inverting the output (L + R) of the fourth matrix circuit 37, an output of the first matrix
circuit 33 and the above The front left signal L22 and the front right signal RP2 are obtained by
combining with the output of the first variable gain amplifier 35, and the both signals Lt2. . 5th
mad 2 nox times% s9. A combination of the output of the matrix circuit 36 of the easy 8 and the
output of the second seismic gain increase IJ gate J8 is t to obtain a left signal 2 left and a rear
right signal R12, and a nine picture signal Lm2, The sixth matrix-path 40.2 channel signal which
attenuates aRm2 by 1 / v and the signal of the 8th variable gain amplifier 41 ° which increases
the right signal R of one side of H and the signal of 2 channels, The fourth variable gain amplifier
42 for amplifying the other left signal of H, the output of the eighth variable gain amplifier 41
and the signal of the two channels, and the other left signal of H are combined to obtain the front
left signal Ly5 and The seventh matrix circuit 43 for obtaining the rear left signal 7) 5, the output
of the fourth variable gain amplifier 42 and the two-channel signal, and the R -1 right signal RA
are synthesized to obtain the front right signal Rv5 and the rear right Eighth matrix circuit 44 to
obtain signal 3 Corresponding each output Lv2sRv2sLm2eRs2 of the fifth and sixth → tricks
circuit 39 ° 40 and each crystal power 'LF3 # Lms # Rys # R Fuji 3 of the seventh and eighth
matrix circuit 43.44 respectively No. 9 to synthesize one, m.sup. Twelfth matrix (ro) path 45. . It
consists of 46.41.48. The first variable gain amplifier 35 has a variable gain coefficient of f and
jj1 and f1 J according to the first output Ef of the first 1p phase discriminator 49 described later.
So it changes from 8.414-0. The second variable gain amplifier 38 has a variable gain coefficient
b, which is dependent on the second output Eb of the first phase discriminator 49 described
later. Changes from θ to 8.414 as shown in FIG.
Said 8th variable variable gain booster 4 is! This gain mantissa J changes from 8.414 to 0, for
example, as shown in FIG. 11 according to the first output U of the phase discriminator 56 of the
fringe 2 described later. The fourth variable power amplifier 42 has a variable gain coefficient r,
and the gain coefficient r is indicated, for example, at 116 in accordance with the second output
Er of the second phase discriminator 56 described later. Varies from 0 to 8.414. Thus, the first
phase discriminator 49 of the input terminals 31L, 31'R'IC is connected. This f? ! 10 phase
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discriminator 49h, phase discrimination between signal 'L, R of 2' channel is performed to detect
the phase relationship existing between the signals, and the first corresponding to this, this 141%
second Output Ef, ..., Eb. EndPage: 45-0 ··· This is supplied to the eleventh second variable gain
amplifier ss, ss, and has, for example, the characteristic (change of ± 2.414) as shown in FIG. .
That is, when the phase relationship between signals of two channels and R is in phase (θ °),
the first output Ef is a positive voltage, and the second output Eb is a negative voltage. In
addition, when the phase relationship becomes opposite phase (180 °), the first output Ef
becomes a negative voltage and the second output Hb becomes a positive voltage. Furthermore,
7 m! When the above-mentioned phase relationship is 90 °. −1. 'Second output Ef. Ebt will
be the same as a turtle. このように! Phase valve discriminator 49 of No. 1 of 1. The second
outputs Ef, Eb are signals of two channels, and the output terminals of the first phase
discriminator 4; and the 'first and second variable gain amplifiers 35 according to the phase
change between R' and R '. : During IB, the correction circuit 50,. 5 and J are set up. The
correction time, path 50.51.perp., Characteristic i of the first phase discriminator 49, for
example, correction as shown in FIG. 18, so that control in the positive direction and control in
the negative direction are different, The gain coefficients f and b of the eleventh and second
variable gain amplifiers 35 and 38 are changed so as to be 1 at the orchid (center) between the
front and the back as shown in FIG. That is, with respect to the center l, control in the positive
direction is +8.414, and 11iI in the negative direction is 0. Here, each 1F, J * coefficient of il # the
11th second variable input gain multiplier 35m 3 B by the first phase discriminator 49. ', A state
V (see FIG. 11). In other words, since the encoder negative Madosuk's base is as shown in the
penalty distribution (8), it is possible for the signal of the i channel, the one left shift -kjL of R,
and Therefore, the phase changes continuously toward 0 ° 〃1 et al. + 90 °, and similarly, the
reciprocal stone color number R changes systematically toward 06 or 1p-90 °.
Thus, the phase difference of the two-channel signal, R 同, changes continuously from the front
to the back from θ ° to 180 ′, and the upper gain covert f, b is shown in the turtle 11 figure.
Oh. Change. Further, the input terminals 3JL and 3JhKFi phase shift pictures @sz and ss are
connected respectively. The phase shift circuit 63 produces a phase shift characteristic delayed
by Fi.sub.45.degree. From the phase shift characteristic of the phase shift circuit 52. FIG. Each
output of the phase shift circuit 52.53 is supplied to a matrix circuit 54; 6'5. Matrix of matrix
[glMb 4 f 2, the phase shift circuit 52 ° 6-3. The j channel signal after being phase shifted by.
And a circuit 55 for combining the signals L% R-45 of the two channels. And a circuit for
obtaining the difference signal 1L-(-R-45)). Thus, the matrix circuit 54 is supplied. This second
phase discriminator 65 is a phase discriminator between each output (L + (R-45.degree.)) And (L(11L.sup.-, 45.degree.)) Of the above-mentioned I.sup. To detect the phase relationship between
the signal rkiK and the presence l, and the first output El and the second output 1? To output the
output EJ # Er of the eleventh axi 2 to the first 1e eighth and fourth variable gain multipliers /
senders 41.42, respectively, the first phase discriminator 149 and the first phase discriminator
149. For example, a colleague has the characteristic as shown in FIG. A correction circuit 57.58 is
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provided on each of the output terminals of the second phase discriminator 56 and one of 1rIJItl:
eighth and fourth variable gain amplifiers 41 and 42, respectively. This correction circuit 57.58
corrects the characteristics of the second phase discriminator 66 as shown in FIG. 18 in the same
manner as the “correction circuits 50” and 51, as shown in FIG. Variable side of 4, gain
coefficient of gain amplifier 41 ° 42! , R, as shown in FIG. 11, is changed so as to become l
between the left side and the right side (center). Here, the eighth and fourth variable gain lateral
multipliers 41. . The manner in which each of the 42 gains 4ik * I3 and r are extracted as shown
in FIG. 11 will be described. That is, since the phase difference between the output (L + (R−45
°)) of the above-mentioned EndPage: 5 circuit 64 and the output (L− (R−45 °)) of the matrix
circuit 56 is detected. The tonality moves to the right K so the phase difference between the
output (L + (R-4-5 °)) and (L-(R-45 °)) is from left to right 0 towards.
It changes continuously to ~ 180 °. Therefore, the gain coefficient 43−, rtd changes as shown in
FIG. -The outputs of the ninth to twelfth matrix circuits 45 to 48 are respectively supplied to the
output terminals 63, 64 and 65. 66 through the phase shift circuits 59.degree. The abovementioned left terminal LF4 from the output terminal 63, the above-mentioned output terminal
64 or the left signal L14, and the right-hand signal kmaYr + from the above-mentioned output
terminal 66 are set to 2. “The phase shift circuit 59, 6.0 has equal phase shift characteristics,
and the phase shift circuit 61 is 90 ° behind the“ phase shift characteristic of the phase shift
circuit 59.60 ” -J) Use one having a phase shift characteristic, and use one having a phase shift
characteristic that leads the above-mentioned phase shift circuit 62.degree. . Next, a circuit that
embodies the s + IWc0J fast return frame matrix circuit 32 will be described by 114-. Resistor
group 71 forms a matrix circuit 34 of 謁 2, resistor group 72 forms the fourth matrix IJ circuit
37, and resistor group 73 p 74 # 75. 76 is the eleventh nest 8, respectively. The fifth to twelfth
matrix circuits 33, 3 □ 6 * 39-40 * 4B to 48 are configured. Base or input terminal JJLK divided
nfchq N type transistor 77 rLz signal and divide it into 1 '= @ L and -L different in polarity
mutually, and this each signal, L gate -L to each above matrix circuit It is a transistor for
supplying. The transistors are transistors connected to the input contacts 31B for supplying the
respective signals R and -R to the respective matrix circuits. 35 is the first t, the first variable gain
field + s can, and the output signal of the second matrix circuit 34 is increased. The npn-type
transistor 29 is used to divide the output signal of the transistor 79 into signals different in
inertia from one another and to supply the respective signals to the fifth matrix-path 39. The
gain is controlled by the first output 21 of the phase discriminator 49 of 181, and the main body
is a junction field effect transistor (hereinafter simply referred to as FE'I) 8, 1 mainly controlled.
It is composed of 38 is the second variable gain 瑠 II! In the J unit, the output signal of the fourth
matrix circuit 37 is amplified and supplied to the sixth matrix [Bk 114.0]. Is mainly composed of
a gain control FET 8'3 that controls the first six phase discriminator 49 according to the second
output EfK of the first six phase discriminator 49.
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4 × ld The eighth variable-rate ranger 84, FET II5 for gain control which controls the
amplification gain of the transistor 84 in accordance with the output E7 of ml of the second
position discriminator 56 is extended to the living body. Reference numeral 42 denotes the 48th
variable gain amplifier, and the eighth signal! NPN transistor 86 for boosting I 供給 す る
supplying to the tricks circuit 44, boosting-of this transistor 86-JJgf: FE 'for gain damping that
performs tll J in response to the second output Er of the second phase discriminator 56 l'87 the
main 0 body 5i! It is The ffi resistors 88 to 91 in the figure are FETs 81.83685. , And lj7 for
adjusting each bias Kameda, and variable resistors, and the units 92 to 95 are for adjusting each
control +4 of the FETs 81, 83, 85, 87. Here, the 纂 1-fourth '5' variable gain amplifier 35.3 g, 41.
. Since the operation of 4B is respectively equivalent, the variable gain amplifier 35 of 1i 41 will
be described as a representative, the output If 'of the first phase discriminator 49Qml is the gain
of the correction circuit 50 and the variable resistance When it is applied to the gate of the IIET
81 through 92f, the FET, s1 勢 makes the EndPage: 6 in response to the “signal Ef”, and its
internal resistance value (resistance value between source and drain) changes. That is, the FET
81 operates as an edible resistor) whose resistance value changes in accordance with the signal
Ef. When this happens, the resistance value between the ground of the transistor 29 and the
ground of the transistor 29 is converted to an i with respect to the ac signal because it is
connected in parallel with the emitter resistance H, m of the FET 81tI'i). The amplification gain of
the transistor 9 is controlled. That is, as described above, the gain coefficient f of the first
variable gain amplifier 36 is produced according to the first output Ett of the first phase
discriminator 42. The operation of FIG. 1 is described in the table with 15 or more
configurations. For example, input terminal 31L now. When the signal of the 2 channels is
applied to 31 R and R is applied, it is demodulated and the fifth to fifth. The respective outputs Lv
2 '' obtained from the sixth matrix circuit 39. 40; 2 ′ ′ t, 4-: (′ knee + 26 ((′ ′ l + f) L + (□
−13, Lm 2 −L / y 4 <(L−R) × b (R to Xi)) = L /) 4 ( (1 + b) L-(1 '-b) R) R 2 / for ((L-n) + b (L +
R)) = letter' 1! 1 ((1 + b) R-(1-b) L) and the seventh and eighth!
Trickles circuit 4J, 44Rm! i 表 Express as follows. Lデ!
7L+JRLss=L−4R−Ry5mR+r’I;R■5xlij−、yI。 The final fourchannel signal LF4ef obtained from the output terminals 63 to 66 is synthesized by the matrix
circuits −45 to 48 of the respective value numbers and the 9th to ill doubled in this way. y4 #
L14 # R pus 4 L as follows. L system 4 = L 'system 2 + L v 3-' = L '\' (, (1 + f-h 4) L + (1-f break)
R), "冨 1 / I! ((I-h / ri L + R mot t (1;-R)-1 v 4 JR) "'-(10) R v 4 e-R 2 -axis 3" VB VB ((1 + L-1) meter
(1- f ~ 1 r,) L) = Shi 'y' l !! ((L 峠 V)) R モ j R j (R-L) + 冨)) ... (11) * 4 =-J (L 塾 2 + t, m 5)--J L /) 15!
((1 + b−h / 1) L− (1−b + 46) R) = − JVv′Q ((1 + s / n) L−R + b (U + R) uR) ′ ′ ′ ′ · (1g)
Rm4 = J (R · 2 + RIB ) = Jl / l ((1 + b → y'1) R-(1-b + 100 r) L). =Jし
J((lhl)R−L+b(R+t、)v’! The operation of the path 32 will be described in
detail below. Only the front left channel and channel will be described as a representative. Here,
since the equation (10) of the final left signal LF 4 demodulated is the whole 4 and the equation
(10) is simply attached, Ly 4 =, when the equation 4 is omitted. (1 + f) L + (17 f) R (distress R) =
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(l-1)-1-1-f) L + CI-f'W 'to 8) R ..... (1) Therefore, if it is in the case where the setter 9 is not applied
now or all channels are equally input), each gain coefficient teb * JarB becomes l in this state.・ It
is set. Therefore, if 1 is substituted for f and b in the above equation (14), then Ly4- (1 + y'2 + l) L
+ (1-1 + 11−15) R4,414L + 1.414R = 8.414 (L + 0.414R), The vector directions of the
demodulation matrix in this case are as shown in FIG. Let the above-mentioned 8.414 t-basis
level at this time be Qdb. -The remaining eight channels can also be expressed with the above
thought. Therefore, in the set state ν, the vector directions of the respective re-matrices are
respectively directed to symmetry (square) as shown in FIG.
In this state, it is now assumed that the phase identifier 49 of silk l is in phase (0 °) operation,
and the second phase discrimination @ 56 is It is the opposite phase (180 ') operation. Therefore,
each gain factor L, and so on. EndPage: 7Li4 = (l is also 8.414) L + (1-8, 414) R-5, 828L-2, 414R
= 5. 828, which substitutes the number into this equation (14). It becomes (L-0, 4141-+).
Therefore, the vector direction of the forward right signal RP being inputted is changed from the
front right signal Rv to the reverse direction in which crosstalk does not occur or to the rear left
channel LmO side. Next, assuming that the rear left signal Lm is input to the encoder, in this case,
the w, l's digit #I 9 P separater 49 becomes the reverse phase (180 °) IEi 1, and the second digit
discriminator 56 , Are in-phase (θ °) operation. Thus, each cell 11 coefficient is f = 0, b = 8.414,
-6 = 8.414, r = Q. Substituting this coefficient of 8 into the above equation (14), Li4 = (1-1 v / L +
(1 ×, x8.414) R ′, = 2.414 L +, 828 R = 5.828 (0, 414 L + R) − It becomes'. When the rear left
signal Lm is input, as shown in FIG. 17, the vector direction of the rear-matrix corresponding to
the front left signal L14 'which is the 1 @ contact channel is four stoke from the rear left signal
Lm. Change to the reverse direction Tsukishi square channel R 0 0 @ where does not occur. Next,
assuming that the front right signal Lv is input to the encoder, in this case, the first v second
phase discriminators 49 and 56 respectively generate in-phase (θ °). Therefore, each gain
coefficient is f = 8.414, b-0, -4a = 8.414, r = 0. Substituting the obtained coefficient into the
above equation (14), L12 = (1- + A + 8.414) L (1-8, 41m1 × 8.414) R ′ ′ 5.828L + 2.414R,
x5asis <L + 0.414R Therefore, when the front right signal LP itself is inputted, the vector
direction of Li2 does not change, and as shown in FIG. 18, it corresponds to both channels lt4 #
L4 which are in contact with each other. The vector direction of the l-matrix is the opposite
direction from the front right signal L1 where no p-stoke occurs or the rear right channel 1m.
Slow down to jealousy. Next, forward right signal R? Assuming that the middle right central
signal Re between the rear right signal Rm and the rear right signal Rm is input, in this case, the
first phase discriminator 49 ′ operates with a phase difference 90 ′, and!
The eight phase discriminators 56 are in reverse phase (180 °). Once each gain factor is f = l-b,
Fl-II = Oar is 8-414. Substituting this gain coefficient into the equation (14), L12 = ('l-h / rPr-1) L
+ (1-1 +' b) R = 8.414L. Therefore, when the right center signal Re is input, as shown in FIG. 19,
the front left signal. The vector direction ° of the return 14w corresponding to L12 changes in
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the right central signal R · · · · ト ー ク ト ー ク 妙 1 逆 逆 逆 中央 left central channel Le (LIIIK).
Also, at this time, right around Central British Signal R @? In order to reproduce 4 'signal-(sound)
in stereo (front and rear stereo), the matrix corresponding to the signal RF4' 11m4 of both
adjacent channels is opened, and the matrix corresponding to the front left signal Lv4 Lower
one's gain. (In this case-7 db lower) Next, assuming that the left center signal Le between the
front left signal LP and the rear left signal Lm is input to the encoder, in this case, the first phase
discriminator 49 With 90 ° operation, the second phase separation 656 is in phase (0 °)
operation. Therefore, each gain coefficient is f ′ ′, b = 1−J−8, 414, r = 0. Substituting this
gain coefficient into the equation (14), Li4 = (1 + s /! +1)L+(1”l”h’I! X8.
'414) R = 8.414 L + 4. 828 R 4.8 g 8 (0, 707 L +' R) = 4. 828 X 1.3; 26 (Lmbe 85.2 ° 十 Re * s
85.2 °) = 5.914 (Lsim 85.2) When the left central signal Le is input EndPage: 8 as shown in FIG.
20, a signal (sound) near the left central signal La is obtained. Is now generated in stereo (round
after stereo), @ with the demodulation matrix corresponding to the signals Lv4 and 5m4 of both
adjacent channels open, and the level of the signal Lv4, L1'4 due to a change in the demodulation
matrix Later on to increase the gain of the matrix so as to prevent it from falling. Next, assuming
that the middle rear center signal CI between the rear left signal Lm and the rear right signal RB
is input to the encoder, in this case, the first phase discriminator 49 reverse phase (180 °)
operation and 懲The second phase discriminator 56 operates with a phase difference of 90 °,
but f = O 2 b = 8.414. J”I’mr、]1となる。 Substituting this gain ゝ coefficient into the
above equation (14), Lv4 = (1 + 1 ′; i) t + (i + 4) k−2,414 CI, 10 B) = 8.414 (L sin 4 r 5 ° R cos
45 °).
After that, when the rear central signal C1 is input, j-, as shown in FIG. 21, the vector direction of
the demodulation matrix corresponding to the front left signal [, P4 is from the rear central
signal Cm. In the opposite direction where crosstalk does not occur, tr forward center channel
CyO changes to itself. At this time, at this time, in order to reproduce the signal (sound) near the
rear center 1g No. C1 in stereo (mostly left and right stereo), the later theory matrix
corresponding to the signals L14 and R14 of both adjacent channels @ is opened. In order to
prevent the level of the signals Lm4 and Rm4 from being lowered, the gain of the @ chest matrix
is 9 at the top and the gain of the rim) corresponding to the front left signal Lv4 is the lower
edge i. Next, assuming that the middle front center signal CF between the front left signal L1 and
the front left signal RF is input to the encoder and it is nine, the i th phase discriminator 49
operates in phase, (0 °). The second phase discriminator "unit 661" @ 90 ° difference
operation. したがって。 Each gain coefficient 紘 · · · t 5 = 5.5g 5 (t-R) = a, 52s (L-o, 17iaa) = 5.
828 × 1.0 15 '(Laos 9.7 °-Rsln 9.7 °)' = 5 9 ° 14 (Lgos 9.7 °-Rsim 9.7 °). In the case where
the front center signal C 九 is inputted, as shown in FIG. 1128, the adjacent picture is reproduced
in order to reproduce the signal (sound) in the vicinity of the front center signal Cr in stereo (#t,
left and right stereo). When the ceramic matrix corresponding to the signal Lv'aRy4 of the
channel is opened, the demodulation of the signal Lv'eKy4 is prevented so that the level of the
signal Lv'eKy4 is reduced due to the change of the frequency. Ori gets up in the matrix. The
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above-mentioned explanation was made about only Bod-Mingsha forward left channel only, but
other? The same is true for the remaining channels, so the explanation #J is omitted. このように
し。 Of the variable recovery vI4 matrix circuit 32 according to the 2 channel signal and the
phase relationship of R @! The trix coefficient is variably controlled. That is, it changes as shown
in FIG. It is to be noted that the directions of the sound source input as shown in FIG. 15 to FIG.
28 are indicated. As described above, the decoder shown in FIG. 1θ supplies a 2-channel signal
to the variable matrix circuit, and performs a predetermined addition 11 cm between the 2channel signals in this matrix circuit before. Signal of 4 channels consisting of 2 channels and 1
rear channel are taken out ', and the phase relationship between the signals of the 2 channels is
detected by a phase discriminator, and the matrix circuit of By controlling the matrix e number
so as not to generate an unnecessary signal in each channel individually, it is possible to improve
the separation of each channel −) without including necessary information included in each
channel The
In the case of the 10'1 iip decoder described above, the matrix circuit j2! When the Trix
coefficient is freely adjusted EndPage: 9j, as is apparent from the above description, the
resistance value between the source of the control FWT and the drain is changed according to
the output signal of the phase discriminator. By changing the resistance value between the power
supply transistor and the ground of the amplification transistor and thereby controlling the gain
factor of the countable gain amplifier, the matrix factor of the matrix factor 32 is variably
controlled. There is. However, a decoder using an FET as an adjustable resistor f which variably
controls the matrix coefficient like this has been opened as follows. In other words, generally,
paT has different operating points depending on the PET used in order to cling the pinch off with
pressure (hereinafter simply referred to as Vp). For this reason, it is necessary to stitch Vp for
each FET to be used. For example, in FIG. 14, with the variable resistors 88 to 91, -FETs 81.8B,
'85. Each of Ij 7 (Ias 篭 pressure f: adjusting, each FET o, V p ′ ′ is 114 M ”. In this way, it is
very bothersome and inconvenient because it is not possible to make ill & l's regret for that kind
of FIT. Also, the control width of the FET must be set so that the gain coefficient of the variable
gain amplifier is 1 illjll from 0 to 8.414 around 1 as shown in al1. For example, in FIG. 14, FETs
81, 83, 85 by changing one of the variable resistors 02-95. Each control width of 81 is set.
However, when the variable resistors 92 to 95 are eaten, FET #J. 8: 1.85. Each Vp of lJf / is
shifted again. Because it is necessary to re-adjust Vp of each FET, it is very troublesome.
Furthermore, when the center point [0] between the output signals of the phase discriminator is
changed, the output signals of the phase discriminator do not change symmetrically as shown in
FIG. When this happens, the control center point of the control FE and T fluctuates accordingly,
and the control range fluctuates. This rounding, the variable matrix coefficient 11j can not be
accurately controlled, and the separation is deteriorated. I also had a problem. 'This invention
was made in accordance with the above eight articles and has its purpose. The place is the matrix
coefficient. Funny 11! It is an object of the present invention to provide a pre-write signal
conversion device which can measure no adjustment of the threshold pressure vth (or Vp) of the
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control FIT for lJa, and can prevent the fluctuation of the control width of the FET.
The present invention leads to the following considerations. That is, if Vp of the control FET can
be made non-aligned by managing Vp of the FET, it is theoretically impossible at present with
conventional junction FETs. However, it seems to be possible in 'MOS tv' pg 'r, and the threshold
pressure of the MO8 FET is simply referred to as vth. (This corresponds to the Vp of the junction
FET)], for example, when the distribution of 295 (in the case of the insulator of a) is investigated,
the deviation of the network ± 8. , Vth 4.9V ~ 5.5V) シ ′, の ま ま ′ 嬬 嬬 junction type FET よ
夛 夛 Although the deviation is small, no adjustment of vth. However, when the distribution of vth
of MOB type FETs in the same array was examined, it was about ± 1.8% (± o in vth) with
standard deviation ± 8σ (6 □ sov in σ). , Osv). This is said to be sufficient-accuracy. Therefore,
it is possible to eliminate 1 and vth by using the voltage obtained by the FET of 01 in play (based
on the vth of the DMO 8 type FET) as the bias voltage of PET. From the above, the present
invention uses, for example, any of the MO8 type FETs in which multiple layers are formed in the
same array as a control FET, and the bias voltage of the FET is set to the other MO8 in the above
array. The output FET of the phase discriminator is obtained by synthesizing each output signal
of the phase discriminator by compensating the v'th of the control FET by providing the FET of
the source follower connection 1 and compensating the v'th of the control FET 1 An average
value signal is added to compensate for the variation of the control center point of the control
FET due to the variation of the center point between the force signals by the outputs of the sea
phase discriminator. 7. Refer to the drawing for 1j & f + of the main text, attach the same
reference numerals to the same part as the reference EndPage: 10, and omit the static power of
the static axis. As shown in FIG. 24, the present BAh, even as shown in FIG. 24, is a pair of cans,
FETs Jl1, #J, # 5, J, 7 formed in the same array. And four other MOg type FETs are used
respectively, and these FETs # 1,... The FET 96 is a FET for vth compensation, and the FET 9g of
the FET 96 is source-connected, and the ring pressure (source voltage) obtained from the FE'l ''
96 connected with the racetrack is F'LTa1 # 8B, It is given as each of the 85 and 87 bias
voltages.
Also, the first phase discriminator-a! e;> sl, ms output force signal Ef, Eb, 7 n resistance sensitive
resistance R f, average resistance signal obtained by combining Rh (one cabinet signal E f,
resistance 箪The y-equivalent value 16 (central point of No. 1b No. 1b 'EJ + Er 藺) obtained by
synthesizing with equivalent resistances RJ and RrK is added to the gate of the above-mentioned
compensation FET 96, respectively. The vth of the t * d% hardened phase FET 96 in this 4111 is
the reference, and the voltage obtained from this FET 96 becomes the bias voltage of each of the
control FETs 81, #J, 85, and # 7. ) Costs FgT8J, 8J. Each bias voltage of 8JaJ7 is equal to the vth
of the compensation FET 56. Therefore, even if it is open, FEi '81 m 8 B, h 6, δ 2 よ g [柑 FE ′ l
′ ′ 9617) Vth is Y-identical to each other n! fl, regardless of how much it is lmm-each vtb is
compensated dynamically, and it is necessary to adjust vth each time as in the prior art <, Vth
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non-composition is measured,-extremely 1 #! It is Oi, and it is a wig of the KX attachment. Also,
the output signal f f of the first phase discriminator 49. A central point between Eh and an output
signal E-6 of the second phase discriminator 56. Even if the central point between Er fluctuates.
It is possible to prevent the fluctuation of each control width of the control FETs 81.88 and 85 °
87 due to the fluctuation. That is, for example, it is assumed that the central point between the
output signals Ef and Eb of the 19th μ μ discriminator 49 fluctuates. In this case, since the
average value signal if of the output signals Ef and Eb of the first phase discriminator 49 is
applied to the gate of the compensation FE 796, the gate voltage of the F and ET 96 changes t
according to the fluctuation. The source voltage, that is, the control FET 81.88. Each 85.87 bias
voltage changes. As a result, the control center point of the pg'r'st'sg, a, 85.87 changes in
response to the change. That is, each control center point of the FETs 81. 88 ', 8j, 87 follows the
fluctuation of the center point L12. The control FETs 81, 8B, 85, 870 are always automatically
corrected toward the central point between the output signals of the one phase discriminator. ま
た。 The same applies to the case where the central point between the output signals EleEr of the
second phase discriminator 56 changes. Therefore, even if the center point between the output
signals of the phase discriminator fluctuates, each control width of the control FE 781. Variable
control of the scan factor can be performed.
In the above embodiment, although a case where a plurality of MO8 type FET'e formed in the
same array is used is described, the present invention is not limited to this, for example, a
plurality of formed in the same array A junction type FET may be used. In this case, however, the
deviation of Vp of each FET must be equal to or less than that of one of the MO8 FETs--Also, in
the above embodiment, the signal between two channels and the phase between R The phase
difference is detected by the first phase discriminator 49, and according to its output, the first
one. The phase relationship between the second variable gain, control, and 2-channel signal, H
phase deviation EndPage: No. 11 and the difference signal is detected by the second phase
discriminators 5, 6, Accordingly, the case of controlling each variable gain coefficient -4ar of the
j-th and fourth variable gain amplifiers 41.41 has been described. However, the present invention
is not limited to nine as shown in FIG. In place of the discriminator 49, a first ratio softener -111
is provided, K, a signal of 2 channels at the input end of the first comparator 1110, and a is
synthesized to obtain a sum signal (L + R) The IJ circuit 112 and the matrix circuit 118 for
obtaining the difference signal (L-R) are provided, and the outputs (L + R) and ('L-) of the IJ
circuit 112.degree. = R), and the level difference between the signals According to the 1st 0
magazine 0 variable gain increase! Control each variable gain, coefficients f and b of the
comparator 35.3-8, and an alternative second comparator 114 'of the + distribution 21 phase
discriminator 56, a second comparator of the amplifier In the same manner as in the above
embodiment, the case where the signals of R2 and R are compared by 114 and the eighth and
fourth variable gain amplifiers 41.degree. According to the level difference between the signals
can be implemented. The specific circuit example of the variable demodulation matrix circuit 3'2
in this case is shown in FIG. 26. The system of the present invention is not limited to the above
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embodiment, and the main part of the present invention is not changed. Of course, various
modifications can be made within the scope. As described above in detail, according to the
present invention, the adjustment of vth (or Vp) of the control FIT which variably controls the dJ,
Ix coefficient is measured, thereby greatly simplifying the adjustment operation of the apparatus,
and It is possible to provide an acoustic signal converter capable of preventing the fluctuation of
the control width of the FET due to the fluctuation even if the central point between the output
signals of the phase discriminator (or the comparator) fluctuates.
4. Brief description of the drawings, 'FIG. 1 is a probe diagram showing an antecedent 4-channel
stereo system, FIG. 3 and FIG. 4 to 9 are diagrams for explaining FIG. 1 respectively, FIG. 10 is a
block line showing an acoustic signal conversion device according to the prior art, and FIG. 11 is
a diagram for FIG. The figure which shows the change state of each gain coefficient of a 4th
variable gain amplifier, the figure which shows each characteristic of the 1-1st 21 phase
discriminator in FIG. 12 company FIG. 10, and FIG. 18 said above. Diagram showing
characteristics of first and second phase discrimination-after correction-Fig. 14 is a block
diagram of the variable matrix circuit in Fig. 1O, and Figs. 16 to 28 show the operation of Fig. 1O
respectively. FIG. 24 is a circuit diagram showing one embodiment of the present invention, FIG.
24 and FIG. FIG. 6 is a block diagram showing another embodiment of the present invention and
a circuit diagram of the main part, respectively. 32 ··· Variable matrix circuit, 49.56 ··· Phase
discriminator, 81, 88, 85.81 ··· Variable resistance element for mad 5 lix factor variable control
FIT for control, 9 ts · Vtb compensation Fl'l 151. 114 ・ ・ ・ Comparator, "Applicant agent"
Patent attorney, Takehiko Suzue EndPage: 12-piece 2 @ juice 3 figure 4 figure O off 5 figure -8, 3
dB-5 JOB piece 6 @ piece 8 figure (a) ( b) 'piece 9 figure EndPage: 13 pieces 11 figure' 0-----1-1-3
A 14 piece 1 poor figure yF 13 @ End Page: 14-y Pts mouth piece 16 figures piece 180, piece r1
@ off the coast 9 figures 20 figures EndPage: 15aX-list of attached documents (3) drawings (4)
Requests for duplicates ('Iga Inventors, patent applicants or agents other than the above address
Shiba Nishikubo Sakuragawacho Minato-ku, Tokyo 2 Street number 17 Mori Pino (. Name (5743)
Patent Attorney Takeo Miki, '= 22: 3! = 1: 1, 4 * 22 'n Name (6881) Patent attorney Atsushi
Tsuboi Address same place special side Shodan-4511 (1B) Showa Date Patent Office Secretary
Hideo Saito 1 Showcase of the Case Japanese Patent Application No. 48 -95 908 No. 2, name of
the invention 4, agent "Address: Tokyo Metropolitan Ward, Shiba-ku Kubo, Sakuragawacho 2,
17th No. 17 Mori Building 〒 105 Telephone 03 (502)-3181 (Large Representative), ', Name
(5847) Patent attorney Suzue-Takehiko 4-5, voluntary correction? Details of the correction (1)
Description is given in FIG. The following description tube is inserted between "" and "other" of
the 6th line. "The above is the explanation for the case of using FBTt as a matrix variable
resistance element in the 4f channel matrix decoder of QS system, but-it is possible to switch to
another matrix system bird cage-8Q system if necessary. In the same manner as described above,
the same circuit is used for the FFfT in the same manner as described above in the case where
the electronic level is made variable using FIT in order to obtain a bias power source compatible
with each system in the 4 channel decoder FIG. 27 of another FET in a circle shows a bias power
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supply switching path of a four-channel matrix decoder switchable to different types of Setrix
systems as above (ie, FIG. As shown in the figure, it is used as Q8-1 (for m11, for Qe, for SQ,
FIT150, 151 ° 1521-compensation FIT The same FBTs 15B and 154 as the control FBT shown
in FIG. 24 as the voltage (source voltage ratio) t bias voltage obtained from the FIT1i sourcefollower-connected and the source-follower-connected FITs (corresponding to front and back in
this circuit) Q5-Hall).
Q8.8 Q selector switches 155, 156 and 157t are provided via. Therefore, if the switching pin
156. 157 is in the switching position corresponding to <8Q, even if the values of the control FF
1T 15 B, 1.54 and 8 Q, and F 11 iT 152 differ, if automatically shown! rvth bites compensation.
Changeover switch 155.156 is another! Even when one is in the replacement device, the vth of
each PET is compensated as described above. It is a discontinuous diagram of [circuit of main
part-EndPage: 18 Warning: Page] on page 46, line 181 of the specification. Is a circuit diagram of
the main part, and FIG. 27 is a circuit diagram showing a main part in another further alternative
embodiment of the present invention. And correct. (8) Attached with illll Figure 27 is added. Oki
27 @ EndPage: 19
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